AM to PM correction system for polar modulator

- RF Micro Devices, Inc.

A transmitter includes a polar modulator that creates phase and amplitude signals which in turn drive a power amplifier. To compensate for AM to PM conversion of the amplitude signal into the amplified signal, a compensation signal is generated from the amplitude signal and combined with the phase signal such that when amplified, the compensation signal cancels the AM to PM conversion. The compensation signal may have an offset term, a linear term, a quadratic term, and a cubic term. A second embodiment comprises a technique by which AM to AM conversion may concurrently be addressed using a second compensation signal.

Skip to: Description  ·  Claims  ·  References Cited  · Patent History  ·  Patent History
Description
RELATED APPLICATIONS

The present application is related to concurrently filed, commonly assigned, commonly invented U.S. patent application Ser. No. 10/147,579, entitled “AM TO AM CORRECTION SYSTEM FOR POLAR MODULATOR.”

FIELD OF THE INVENTION

The present invention relates to controlling a power amplifier, and more particularly to controlling the power amplifier in a manner to correct the output Error Vector Magnitude (EVM) and spectrum of the power amplifier.

BACKGROUND OF THE INVENTION

Transmitters form one half of most communication circuits. As such, they assume a position of prominence in design concerns. With the proliferation of mobile terminals, transmitter design has progressed in leaps and bounds as designers try to minimize components and reduce size, battery consumption, and the like. Likewise, modulation schemes are continuously updated to reflect new approaches to maximize information transfers in limited bandwidths. Changes in standards or standards based on newly available spectrum may also cause designers to approach modulating transmitters with different techniques.

Many different standards and modulation schemes exist, but one of the most prevalently used in the world of mobile terminals is the Global System for Mobile Communications (GSM). GSM comes in many flavors, not the least of which is General Packet Radio Service (GPRS). GPRS is a new non-voice value-added service that allows information to be sent and received across a mobile telephone network. It supplements today's Circuit Switched Data and Short Message Service. GSM allows many different types of mobile terminals, such as cellular phones, pagers, wireless modem adapted laptops, and the like, to communicate wirelessly through the Public Land Mobile Network (PLMN) to the Public Switched Telephone Network (PSTN).

One relatively recent change has been the advent of the Enhanced Data for GSM Evolution (EDGE) scheme in GSM systems. This system contains amplitude modulation components, and, as a result, the power amplifier must be linear, never operating in saturation if classical modulation techniques are employed. Such a system lacks the efficiency of one that operates the power amplifier in saturation.

If a polar modulation system is used instead of a classical modulation system, then the power amplifier may operate in saturation and efficiency would be greatly improved. In addition, if the polar signals are generated by a digital method, such a system does not require the use of a high current drain quadrature modulator. Quadrature modulators are undesirable from a design standpoint in that they draw large amounts of current, and hence, drain batteries comparatively fast.

Analog components cause design problems for polar modulators in that the phase and amplitude signals must be aligned so that they arrive at the power amplifier at the desired time. Because of path variations with variable time delay analog components, this time aligning is difficult to achieve. Any solution to controlling the power amplifier should be able to eliminate or reduce reliance on a quadrature modulator and provide digital components such that time alignment is comparatively easy to do.

Unfortunately, further complicating matters, the amplitude signal that controls the power amplifier will cause unwanted phase components to be created in the output of the power amplifier due to the non-linearities of the power amplifier. This is sometimes called AM to PM conversion, and it degrades the spectral purity of the system and the Error Vector Magnitude. Thus, a need also exists to be able to counteract or eliminate the unwanted AM to PM conversion signal from the transmitted phase signal.

An additional concern is that the power amplifier may have a non-linear gain with varying output power. This may create what is called AM to AM conversion. The AM to AM conversion may have both phase and amplitude distortion components, and to create a better control system, these should be reduced or eliminated as well.

SUMMARY OF THE INVENTION

The present invention addresses the problems of the AM to PM conversion by introducing a correction term into the phase path to counteract the non-linearity of the power amplifier. In particular, the present invention uses a polar modulator to generate a phase signal and an amplitude signal. The amplitude signal is split with one part being processed conventionally to control the power amplifier. The other part of the amplitude signal is used to create a correction term that is formed by a sum of polynomials. In particular, the amplitude signal may be acted upon to create a linear term, a quadratic term and/or cubic term, and then summed with an offset term. The summed result is added to the phase signal. The phase signal is then converted to a frequency signal for processing by a phase locked loop.

In an exemplary embodiment, a number of coefficients needed to create the sum of polynomials are stored in a look up table. Further, the look up table has different coefficients based on different power output levels. The terms may be created with physical elements.

In another embodiment, the present invention corrects for both AM to PM conversion and AM to AM conversion using similar methodologies. However, the AM to AM conversion is addressed by introducing a compensation term in the amplitude signal prior to being used to control the power amplifier's input supply voltage.

Those skilled in the art will appreciate the scope of the present invention and realize additional aspects thereof after reading the following detailed description of the preferred embodiments in association with the accompanying drawing figures.

BRIEF DESCRIPTION OF THE DRAWING FIGURES

The accompanying drawing figures incorporated in and forming a part of this specification illustrate several aspects of the invention, and together with the description serve to explain the principles of the invention.

FIG. 1 illustrates an exemplary mobile terminal such as may use the present invention;

FIG. 2 illustrates a block diagram of a prior art modulation scheme;

FIG. 3 illustrates a block diagram of an exemplary embodiment of the present invention;

FIG. 4 illustrates a transmitter chain embodying the present invention;

FIG. 5 illustrates a block diagram of a phase to frequency conversion process;

FIG. 6 illustrates a block diagram of a second exemplary embodiment of the invention;

FIG. 7 illustrates a transmitter chain incorporating the second embodiment; and

FIG. 8 illustrates a block diagram of a third exemplary embodiment of the present invention.

 DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the invention and illustrate the best mode of practicing the invention. Upon reading the following description in light of the accompanying drawing figures, those skilled in the art will understand the concepts of the invention and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims.

While the present invention is particularly well-suited for use in a mobile terminal, and particularly a mobile terminal that is operating in an Enhanced Data for GSM Evolution (EDGE) scheme in a GSM system, it should be appreciated that the present invention may be used in other transmitters, either wireless or wirebased, as needed or desired.

The present invention is preferably incorporated in a mobile terminal 10, such as a mobile telephone, personal digital assistant, or the like. The basic architecture of a mobile terminal 10 is represented in FIG. 1, and may include a receiver front end 12, a radio frequency transmitter section 14, an antenna 16, a duplexer or switch 18, a baseband processor 20, a control system 22, memory 24, a frequency synthesizer 26, and an interface 28. The receiver front end 12 receives information bearing radio frequency signals from one or more remote transmitters provided by a base station (not shown). A low noise amplifier 30 amplifies the signal. A filter circuit 32 minimizes broadband interference in the received signal, while a downconverter 34 downconverts the filtered, received signal to an intermediate or baseband frequency signal, which is then digitized into one or more digital streams. The receiver front end 12 typically uses one or more mixing frequencies generated by the frequency synthesizer 26.

The baseband processor 20 processes the digitized, received signal to extract the information or data bits conveyed in the received signal. This processing typically comprises demodulation, decoding, and error correction operations. As such, the baseband processor 20 is generally implemented in one or more digital signal processors (DSPs).

On the transmit side, the baseband processor 20 receives digitized data from the control system 22, which it encodes for transmission. The encoded data is output to the radio frequency transmitter section 14, where it is used by a modulator 36 to modulate a carrier signal that is at a desired transmit frequency. The modulator 36 may have an optional memory unit 38 associated therewith. Power amplifier 40 amplifies the modulated carrier signal to a level appropriate for transmission from the antenna 16.

As described in further detail below, the power amplifier 40 provides gain for the signal to be transmitted under control of the power control circuitry 42, which is preferably controlled by the control system 22. Memory 24 may contain software that allows many of these functions to be run. Alternatively, these may be a function of sequential logic structures as is well understood.

A user may interact with the mobile terminal 10 via the interface 28, which may include interface circuitry 44 associated with a microphone 46, a speaker 48, a keypad 50, and a display 52. The interface circuitry 44 typically includes analog-to-digital converters, digital-to-analog converters, amplifiers, and the like. Additionally, it may include a voice encoder/decoder, in which case it may communicate directly with the baseband processor 20.

The microphone 46 will typically convert audio input, such as the user's voice, into an electrical signal, which is then digitized and passed directly or indirectly to the baseband processor 20. Audio information encoded in the received signal is recovered by the baseband processor 20, and converted into an analog signal suitable for driving speaker 48 by the interface circuitry 44. The keypad 50 and display 52 enable the user to interact with the mobile terminal 10, input numbers to be dialed and address book information, or the like, as well as monitor call progress information.

While the present invention is well-suited for incorporation into a mobile terminal, such as the mobile terminal 10 just described, the present invention is also well-suited for use in wireless transmitters associated with wireless LANs and the like. As such, the present invention is not limited to a particular apparatus.

In the past, as illustrated in FIG. 2, the modulator 36 directs two signals to the power amplifier 40. In an exemplary prior art embodiment, a polar modulator 36 directed an amplitude signal (r) and a phase signal (φ) to the power amplifier 40. The amplitude signal (r) controlled the power supply voltage of the power amplifier 40, potentially replacing or including the power control circuitry 42, while the phase signal (φ) was amplified by the power amplifier 40 to create ADESIRED∠φDESIRED. The output (AO∠φO) of the power amplifier 40 was corrupted by AM to PM conversion within the non-linear power amplifier 40, represented by φ(r), and AM to AM conversion, represented by A(r), resulting in an output signal of ADESIRED*A(r)∠φDESIRED+φ(r).

The present invention corrects the AM to PM conversion within the output by preliminarily distorting the phase signal such that when it is converted to a frequency signal and amplified by the power amplifier 40, the predistortion element cancels the AM to PM conversion distortion element introduced by the amplitude signal (r). This is illustrated in a simplified format in FIG. 3, wherein the amplitude signal (r) is split by a polar converter 54 within the modulator 36 and directed to a compensator 56, where a predistortion signal φ′(r) is generated and then negatively added to the phase signal by an adder 58. The combined signal (φ-φ′(r)), expressed as an equivalent frequency signal, is amplified by the power amplifier 40. The power amplifier 40 imposes a transfer function upon the combined signal and generates φDESIRED-φ(r) from the combined signal. The distortion of the amplitude signal is added, and the φ(r) terms cancel out, leaving ADESIRED*A(r)∠φDESIRED.

More specifically, the present invention may be situated in the radio frequency transmitter section 14, as better illustrated in FIG. 4. Specifically, the radio frequency transmitter section 14, and particularly the modulator 36, may include several components, including a serial interface 60, a mapping module 62, first and second filters 64, 66, and the aforementioned polar converter 54. Other components of the modulator 36 will be discussed below.

The serial interface 60 receives Non-Return to Zero (NRZ) serial data from the baseband processor 20 at the bit rate of the system. NRZ data may be a 1B1B code with one line bit for each associated binary bit. In an exemplary embodiment, the modulation scheme for the modulator 36 uses an EDGE modulation scheme, and thus, the bit rate is 812.5 kbps. This data is passed to the mapping module 62, where the data is grouped into symbols of three consecutive data bits, Grey coded, and rotated by 3π/8 on each symbol as per European Telecommunications Standards Institute (ETSI) specifications. The resulting symbol is mapped to one of sixteen points in an I, Q constellation.

Both the I and the Q components for each point are then filtered by the first and second filters 64, 66 respectively. In an exemplary embodiment, the first and second filters 64, 66 are EDGE finite impulse response (FIR) filters. This, as dictated by the ETSI specifications, shapes the response between symbol times.

After filtering, both the I and Q components are sent to the polar converter 54. The polar converter 54 uses a classical CORDIC (coordinate rotation digital computer) algorithm or like rectangular to polar conversion technique. Thus, the polar converter 54 generates phase (φ) and amplitude (r) equivalent signals. Further information about CORDIC algorithms may be found in Proceedings of the 1998 ACM/SIGDA Sixth International Symposium On Field Programmable Gate Arrays by Ray Andraka, February 22-24, pp. 191-200 and “The CORDIC Trigonometric Computing Technique” by Jack E. Volder IRE Trans on Elect. Computers, p. 330, 1959, both of which are hereby incorporated by reference in their entirety.

The amplitude signal (r) is split and directed to the compensator 56. The compensator 56 introduces a compensation term to the phase signal that, after further processing, counteracts the distortion introduced by the AM to PM conversion in the power amplifier 40.

The compensator 56 acts to create a sum of polynomials along the lines of the following equation:

∑ i = 0 N ⁢ C i ⁡ ( r ⁡ ( t ) ) i
In this particular case, N=3 and the equation expands to the following:
φ′(r)=C0+C1r(t)+C2(r(t))2+C3(r(t))3
φ′(r) is termed herein the compensation signal. It is readily apparent that φ′(r) has an offset term, a linear term, a quadratic term, and a cubic term selectable by the hardware implementation.

In an exemplary embodiment of the present invention, the coefficients Ci are associated with the control system 22, and particularly in non-volatile memory 24 associated therewith. Alternatively, the coefficients may be stored in memory 38 if such is present. In an exemplary embodiment, the coefficients may be stored as a look up table. It is further possible that the coefficients are stored as a function of sequential steps performed by hardware. The coefficients are determined through a best fit analysis of a function that substantially matches the unamplified inverse of φ(r). In a more preferred embodiment, a piecewise function is created with each piece being determined by a given power level. This is done to improve the fit between the functions. For example, if only one set of coefficients were used, φ′(r) might not fit well at the ends or perhaps in the middle of the relevant range of values. By implementing a piecewise function, a good fit between the equations may be achieved throughout the curve of relevant values. In an exemplary embodiment, a set of coefficients is created for each 2 dBm power step. This corresponds to the power steps defined in the ETSI standards. To calculate the coefficients, a program such as MATHCAD may be used to derive a match to an empirical power amplifier curve. The coefficients may be tested through an ADS simulation or the like.

The output of the compensator 56 is subtracted from the phase signal (φ) by the adder 58 to create a combined signal. The adder 58 is also termed herein a combiner. The output of the adder 58 (the combined signal) is directed to a phase to frequency converter 68 where the output is converted to a frequency signal (f). More detail on the phase to frequency converter 68 is provided below with reference to FIG. 5. After conversion to the frequency signal (f), magnitude adjusters 70, 72 then adjust the magnitude of the r and f signals to a level expected by the time aligner 74, such that they comply with the appropriate standard. Next, a relative time delay is applied as necessary to the signals for best Error Vector Magnitude (EVM) and spectrum by the time aligner 74. Because these are preferably digital components, concerns about variations in analog components and the corresponding variation in time delays downstream are minimized.

At this point, the r and f signals separate and proceed by different paths, an amplitude signal processing path and a frequency signal processing path, to the power amplifier 40. With respect to the amplitude signal processing path, the amplitude signal is converted to an analog signal by D/A converter 76. While not shown, a ramping function may be combined with the amplitude signal prior to digital-to-analog conversion. The output of the D/A converter 76 is used to set the collector voltage on the power amplifier 40 through a collector regulator 78. As the amplitude signal changes, the voltage at the power amplifier 40 collector changes, and the output power will vary as V2/Rout (Rout is not shown, but is effectively the load on the power amplifier 40). This is sometimes known as “plate modulation”.

The frequency signal is directed to a digital filter 80, a digital predistortion filter 82, and a phase locked loop (PLL) 84, as is described in commonly invented, commonly owned U.S. patent application Ser. No. 10/139,560, filed May 06, 2002, entitled DIRECT DIGITAL POLAR MODULATOR, which is hereby incorporated by reference in its entirety. The PLL 84 generates an output at the desired radio frequency. In an exemplary embodiment, the frequency signal is applied to a single port on a fractional N divider within the PLL 84.

In general, the PLL 84 comprises a reference source that is fed to a phase comparator. The phase comparator compares the edges of the reference source to the output of the fractional N divider and produces a correction signal. The correction signal is low pass filtered and input to a voltage controlled oscillator (VCO). The VCO outputs a frequency modulated signal at the RF carrier, which in turn is fed back to the fractional N divider. The divisor of the fractional N divider is modulated by the frequency signal. Further information on fractional N PLLs, how to modulate a signal by varying the fractional N divider, and the like may be found in U.S. Pat. Nos. 6,359,950; 6,236,703; 6,211,747; 5,079,522; 5,055,802; and 4,609,881 which are hereby incorporated by reference in their entireties.

The phase to frequency converter 68 is explicated with reference to FIG. 5. The phase signal arrives and is split into a delay path and a normal path. A clock 86 controls a delay element 88 in the delay path. The output of the delay element 88 is subtracted from the normal path by adder 90. The output of the adder 90 is multiplied by 2π in a multiplier 92, and a frequency signal (f) is output. This structure takes advantage of the relationship

f = 2 ⁢ π ⁢ ⅆ ( phase ) ⅆ t
in a digital sense. The delay and the subtraction approximates the derivative as (phase(N)−phase(N−1))/T, where T is the period for the clock 86. Other phase to frequency conversions could also be used if needed or desired.

It is interesting to note that this derivative function causes the offset term of the sum of polynomials to be a zero value. However, this does cause an impulse function at the point where the constant is introduced. That is, when the constant changes from one constant to another, such as at the boundary of a piece of the piecewise function, there is effectively a square wave transition. At that point, there would be an instantaneous frequency change. This may create a desired phase offset at the output of the power amplifier 40. This is particularly useful in a General Packet Radio Service (GPRS) system to avoid phase discontinuities when power levels are switched.

In a somewhat related embodiment, illustrated in FIGS. 6 and 7, instead of introducing the compensation signal into the phase signal to correct AM to PM conversion, the compensation signal may be introduced into the amplitude signal to correct AM to AM conversion. As noted above, AM to AM conversion (A(r)) is caused by the power amplifier 40 having non-linear gain with varying output power. FIG. 6 illustrates, in a very simplified form, the concept of this embodiment of the present invention. The amplitude signal is split and sent to a compensator 94. The compensator 94 generates a compensation signal A′(r) which is then added back to the amplitude signal by adder 96. When the amplitude signal (r) and the compensation signal A′(r) are introduced at the power supply input of the power amplifier 40, AM to AM conversion caused by the non-linearities gain of the power amplifier 40 is canceled, and the desired output signal ADESIRED∠φDESIRED is generated, albeit still corrupted by the AM to PM conversion φ(r).

FIG. 7 represents a more detailed view of how the compensator 94 fits within the modulator 36. In essence, the majority of the circuit functions like the circuit of FIG. 4. Compensator 94 acts to create a sum of polynomials along the lines of the following equation:

A ′ ⁡ ( r ) = ∑ i = 0 N ⁢ C i ⁡ ( r ⁡ ( t ) ) i
In this particular case, N=3 and the equation expands to the following:
A′(r)=C0+C1r(t)+C2(r(t))2+C3(r(t))3
A′(r) is termed herein a compensation signal and r(t) is the amplitude of the modulation from the polar modulator 36. It is readily apparent that A′(r) has an offset term, a linear term, a squared term, and a cubic term. In an exemplary embodiment, the offset term C0 and the coefficient for the linear term C1 are zero. An offset term would act the same as increasing or decreasing the output power level. As the collector regulator 78 already addresses this, it is not necessary to repeat the control here. Likewise, a linear term would only change the fundamental amplitude and not change the shape of the curve, so a linear term for this compensation signal makes little sense.

When the exemplary embodiment A′(r) is combined with r(t) in the adder 96, the combined signal is:
r′(t)=r(t)+C2(r(t))2+C3(r(t))3
which converts easily to the following:
r′(t)=r(t)*[1+C2(r(t))+C3(r(t))2]
Thus, even though the adder 96 is an adder, the effect is to multiply the term r(t) by a correction factor that deviates from unity by A′(r). This signal then passes through the power amplifier 40 with AM to AM distortion. This distortion, as previously noted, is A(r). The goal is thus to make the term [1+C2(r(t))+C3(r(t))2] the inverse of the AM to AM distortion such that A(r)* [1+C2(r(t))+C3(r(t))2]=1. When this condition is true, the AM to AM distortion has been canceled.

Alternatively, if the adder 96 were instead a multiplier, then the correction terms could have a linear term and an offset term. From a design standpoint, this removes a multiplier from the compensator 94 and inserts a multiplier in place of the adder 96. The concept of canceling the AM to AM conversion with its inverse remains the same.

In an exemplary embodiment of the present invention, the coefficients Ci are associated with control system 22, and particularly in non-volatile memory 24 associated therewith. Alternatively, the coefficients may be stored in memory 38 if such is present. In an exemplary embodiment, the coefficients may be stored in a look up table or the like. It is further possible that the coefficients are created as a function of hardware. The coefficients are determined through a best fit analysis of a function that matches the expected AM to AM distortion A(r). In a more preferred embodiment, a piecewise function is created with each piece being determined by a given power level. This is done to improve the fit between the functions. For example, if only one set of coefficients were used, A′(r) might not fit well at the ends or perhaps in the middle of the relevant range of values. By implementing a piecewise function, a good fit between the equations may be achieved throughout the curve of relevant values. In an exemplary embodiment, a set of coefficients is created for each 2 dBm power step. This corresponds to the power steps defined in the ETSI standards.

To calculate the coefficients, a program such as MATHCAD may be used to derive a match to an empirical power amplifier curve. The coefficients may be tested through an ADS simulation or the like.

In yet another embodiment, illustrated in FIG. 8, both compensation schemes may be used simultaneously, although it should be noted that the functions now provide some interaction therebetween so that together both the AM to PM conversion and the AM to AM conversion is canceled.

Those skilled in the art will recognize improvements and modifications to the preferred embodiments of the present invention. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.

Claims

1. A transmitter comprising:

a polar converter adapted to convert a signal to a phase signal and an amplitude signal;
a compensator adapted to receive the amplitude signal and distort the amplitude signal such that a compensation signal is generated;
a combiner adapted to combine the compensation signal with the phase signal and produce a combined signal;
a converter adapted to convert the combined signal to a frequency signal; and
an amplifier adapted to receive the amplitude signal and a signal based on the frequency signal and generate an output signal, wherein distortion of the output signal caused by the amplitude signal is canceled by the compensation signal; and
said compensation signal generated with a piecewise function associated with a plurality of different power levels within an overall operating range of the amplifier, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels, and transitioning between two pieces of the piecewise function causes a frequency jump of the frequency signal to at least partially compensate for a phase shift in the power amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function.

2. The transmitter of claim 1 wherein said combiner comprises an adder.

3. The transmitter of claim 2 wherein said adder subtracts the compensation signal from the phase signal.

4. The transmitter of claim 1 wherein the piecewise function comprises a linear term and a quadratic term.

5. The transmitter of claim 4 wherein the piecewise function further comprises a cubic term.

6. The transmitter of claim 4 wherein the piecewise function comprises an offset term.

7. The transmitter of claim 1 wherein each corresponding set of at least three coefficients is stored in a look up table.

8. The transmitter of claim 1 wherein the piecewise function includes at least one of an offset term, a linear term, a quadratic term, and a cubic term.

9. The transmitter of claim 1 wherein the piecewise function comprises a quadratic term and a cubic term.

10. The transmitter of claim 1 wherein the piecewise function comprises an offset term, a liner term, a quadratic term, and a cubic term.

11. The transmitter of claim 1 wherein each of the plurality of different power levels is associated with about a two decibel milliwatt (dBm) power step.

12. A method of controlling a transmitter, comprising:

generating a phase signal and an amplitude signal;
determining a power level of a power amplifier;
generating a compensation signal with a piecewise function from the amplitude signal, wherein the piecewise function is associated with a plurality of different power levels within an overall operating range of the power amplifier, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels;
combining the compensation signal with the phase signal to create a combined signal;
converting the combined signal to a frequency signal;
transitioning between two pieces of the piecewise function to cause a frequency jump of the frequency signal to at least partially compensate for a phase shift in the power amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function; and
amplifying a signal derived from the combined signal with the power amplifier such that an amplified compensation signal and a non-linearity induced phase shift from the power amplifier due to the amplitude signal cancel one another.

13. The method of claim 12 wherein generating the phase signal and the amplitude signal comprises generating the phase and amplitude signals with a polar converter.

14. The method of claim 12 wherein combining the compensation signal with the phase signal comprises subtracting the compensation signal from the phase signal.

15. The method of claim 12 wherein generating the compensation signal comprises selecting the corresponding set of at least three coefficients from a look up table based on the power level of the power amplifier.

16. The method of claim 12 further comprising controlling a power supply input of the power amplifier with the amplitude signal.

17. The method of claim 12 wherein generating the compensation signal comprises generating an offset term, a linear term, and a quadratic term; wherein the offset term, the linear term, and the quadratic term compose the piecewise function.

18. The method of claim 12 wherein generating the compensation signal comprises generating a linear term, a quadratic term, and a cubic term, wherein the linear term, the quadratic term, and the cubic term compose the piecewise function.

19. The method of claim 12 wherein the piecewise function includes at least one of an offset term, a linear term, a quadratic term, and a cubic term.

20. The method of claim 12 wherein the piecewise function comprises a linear term and a quadratic term.

21. The method of claim 12 wherein the piecewise function comprises a quadratic term and a cubic term.

22. The method of claim 12 wherein the piecewise function comprises an offset term, a linear term, a quadratic term, and a cubic term.

23. A transmitter comprising:

a power amplifier comprising an output, a frequency input, and a power supply input wherein signals presented to said power supply input produce a distortion signal at said output;
a polar modulator adapted to produce a phase signal and an amplitude signal;
an amplitude processing path adapted to receive said amplitude signal and present said amplitude signal to said power supply input;
a phase processing path comprising a compensator, a combiner and a phase to frequency converter; and
said compensator adapted to receive said amplitude signal and generate a compensation signal with a piecewise function associated with a plurality of different power levels within an overall operating range of the power amplifier, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels, and transitioning between two pieces of the piecewise function causes a frequency jump of a frequency signal to at least partially compensate for a phase shift in the power amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function;
said combiner combining said phase signal and said compensation signal to generate a combined signal and passing the combined signal to the phase to frequency converter;
said phase to frequency converter converting said combined signal to the frequency signal; and
said compensation signal acting to cancel the distortion signal at the output.

24. The transmitter of claim 23 wherein the piecewise function comprises an offset term, a linear term, and a quadratic term.

25. The transmitter of claim 23 wherein the piecewise function comprises a linear term, a quadratic term, and a cubic term.

26. The transmitter of claim 23 wherein each corresponding set of at least three coefficients is stored in a look up table.

27. The transmitter of claim 23 wherein the piecewise function includes at least one of an offset term, a linear term, a quadratic term, and a cubic term.

28. The transmitter of claim 23 wherein the piecewise function comprises a linear term and a quadratic term.

29. The transmitter of claim 23 wherein the piecewise function comprises a quadratic term and a cubic term.

30. The transmitter of claim 23 wherein the piecewise function comprises an offset term, a linear term, a quadratic term, and a cubic term.

31. A transmitter comprising:

a polar converter adapted to convert a signal to a phase signal and an amplitude signal;
a compensator adapted to receive the amplitude signal and distort the amplitude signal such that a compensation signal is generated, said compensator implementing a piecewise compensation function associated with a plurality of different power levels within an overall operating range of an amplifier, such that each piece of the piecewise compensation function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels, and transitioning between two pieces of the piecewise compensation function causes a frequency jump of a frequency signal to at least partially compensate for a phase shift in the amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise compensation function;
a combiner adapted to combine the compensation signal with the phase signal to create a combined signal;
a phase to frequency converter adapted to convert the combined signal to the frequency signal; and
the amplifier adapted to receive a signal derived from the frequency signal and generate an output signal, wherein distortion of the output signal caused by the amplitude signal is canceled by the compensation signal.

32. A method of controlling a transmitter, comprising:

converting a signal to a phase signal and an amplitude signal;
generating a compensation signal from the amplitude signal, wherein the compensation signal is generated with a piecewise function associated with a plurality of different power levels, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels;
creating a combined signal from the compensation signal with the phase signal;
converting the combined signal to a frequency signal;
transitioning between two pieces of the piecewise function to cause a frequency jump of the frequency signal to at least partially compensate for a phase shift in an amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function; and
amplifying a signal derived from the frequency signal such that the compensation signal cancels distortion of an output signal caused by the amplitude signal.

33. A method of controlling a transmitter, comprising:

generating a phase signal and an amplitude signal;
generating a first compensation signal with a piecewise function associated with a plurality of different power levels within an overall operating range of a power amplifier, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels;
generating a second compensation signal with a piecewise function comprising at least one term having a coefficient set for each of a plurality of different power levels, wherein the at least one term is not greater than a cubic term;
combining the first compensation signal with the phase signal to create a first combined signal;
combining the second compensation signal with the amplitude signal to create a power supply input signal;
transitioning between two pieces of the piecewise function to cause a frequency jump of a frequency signal to at least partially compensate for a phase shift in the power amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function; and
amplifying a signal derived from the first combined signal with the power amplifier such that an amplified first compensation signal combines with a non-linear induced distortion from the power supply input signal such that substantially only a desired frequency signal is presented at an output of the power amplifier.

34. A transmitter comprising:

a polar converter adapted to convert a signal to a phase signal and an amplitude signal;
a first compensator adapted to receive the amplitude signal and distort the amplitude signal such that a first compensation signal is generated with a piecewise function associated with a plurality of different power levels within an overall operating range of an amplifier, such that each piece of the piecewise function is based on a corresponding set of at least three coefficients, which are based on a function that substantially matches an inverse of the phase signal over a corresponding one of the plurality of different power levels, and transitioning between two pieces of the piecewise function causes a frequency jump of a frequency signal to at least partially compensate for a phase shift in the amplifier resulting from transitioning between two of the plurality of different power levels corresponding to the two pieces of the piecewise function;
a first combiner adapted to combine the first compensation signal with the phase signal to create a combined signal;
a second compensator adapted to receive the amplitude signal and distort the amplitude signal such that a second compensation signal is generated with a piecewise function comprising at least one term having a coefficient set for each of a plurality of different power levels, wherein the at least one term is not greater than a cubic term;
a second combiner adapted to combine the second compensation signal with the amplitude signal to create a power supply input signal; and
the amplifier adapted to receive the power supply input signal and a signal derived from the combined signal and generate an output signal, wherein distortion of the output signal caused by the amplitude signal is canceled by the first compensation signal and the second compensation signal.
Referenced Cited
U.S. Patent Documents
3900823 August 1975 Sokal et al.
4389618 June 21, 1983 Bauman
4609881 September 2, 1986 Wells
4837786 June 6, 1989 Gurantz et al.
4968908 November 6, 1990 Walls
5055802 October 8, 1991 Hietala et al.
5079522 January 7, 1992 Owen et al.
5313411 May 17, 1994 Tsujimoto
5430416 July 4, 1995 Black et al.
5444415 August 22, 1995 Dent et al.
5524286 June 4, 1996 Chiesa et al.
5598436 January 28, 1997 Brajal et al.
5608353 March 4, 1997 Pratt
5617450 April 1, 1997 Kakuishi et al.
5629648 May 13, 1997 Pratt
5822011 October 13, 1998 Rumreich
5900778 May 4, 1999 Stonick et al.
5952895 September 14, 1999 McCune, Jr. et al.
6008703 December 28, 1999 Perrott et al.
6101224 August 8, 2000 Lindoff et al.
6115684 September 5, 2000 Kawahara et al.
6130579 October 10, 2000 Iyer et al.
6191656 February 20, 2001 Nadler
6211747 April 3, 2001 Trichet et al.
6229395 May 8, 2001 Kay
6236687 May 22, 2001 Caso et al.
6236703 May 22, 2001 Riley
6236837 May 22, 2001 Midya
6240278 May 29, 2001 Midya et al.
6246286 June 12, 2001 Persson
6271727 August 7, 2001 Schmukler
6275685 August 14, 2001 Wessel et al.
6285239 September 4, 2001 Iyer et al.
6295442 September 25, 2001 Camp, Jr. et al.
RE37407 October 16, 2001 Eisenberg et al.
6307364 October 23, 2001 Augustine
6329809 December 11, 2001 Dening et al.
6335767 January 1, 2002 Twitchell et al.
6356150 March 12, 2002 Spears et al.
6359950 March 19, 2002 Gossmann et al.
6366177 April 2, 2002 McCune et al.
6377784 April 23, 2002 McCune
6392487 May 21, 2002 Alexanian
6462617 October 8, 2002 Kim
6504885 January 7, 2003 Chen
6522121 February 18, 2003 Coumou
6581082 June 17, 2003 Opsahl
6642786 November 4, 2003 Jin et al.
6693468 February 17, 2004 Humphreys et al.
6700929 March 2, 2004 Shan et al.
6701134 March 2, 2004 Epperson
6701138 March 2, 2004 Epperson et al.
6720831 April 13, 2004 Dening et al.
6724252 April 20, 2004 Ngo et al.
6724265 April 20, 2004 Humphreys
6724831 April 20, 2004 Hasegawa et al.
6728324 April 27, 2004 Shan et al.
6731145 May 4, 2004 Humphreys et al.
6735419 May 11, 2004 Mitzlaff
6748204 June 8, 2004 Razavi et al.
6782244 August 24, 2004 Steel et al.
6798843 September 28, 2004 Wright et al.
6801086 October 5, 2004 Chandrasekaran
6807406 October 19, 2004 Razavi et al.
6816718 November 9, 2004 Yan et al.
6819914 November 16, 2004 Yan et al.
6819941 November 16, 2004 Dening et al.
6831506 December 14, 2004 Moffat et al.
6834084 December 21, 2004 Hietala
6836517 December 28, 2004 Nagatani et al.
6901039 May 31, 2005 Sugie et al.
6901514 May 31, 2005 Iu et al.
6903604 June 7, 2005 Kim
6914943 July 5, 2005 Shimizu
6975688 December 13, 2005 Rexberg et al.
H2143 February 7, 2006 Prockup
7010276 March 7, 2006 Sander et al.
7010280 March 7, 2006 Wilson
7012969 March 14, 2006 Ode et al.
7054385 May 30, 2006 Booth et al.
7109791 September 19, 2006 Epperson et al.
7113036 September 26, 2006 Moffatt et al.
7113551 September 26, 2006 Sills et al.
7158494 January 2, 2007 Sander et al.
7349490 March 25, 2008 Hunton
7457586 November 25, 2008 Hietala et al.
7529523 May 5, 2009 Young et al.
7542520 June 2, 2009 Estrada
20010022532 September 20, 2001 Dolman
20020021764 February 21, 2002 Posti
20020041210 April 11, 2002 Booth et al.
20020044014 April 18, 2002 Wright et al.
20020060606 May 23, 2002 Andre
20020093378 July 18, 2002 Nielsen et al.
20020113905 August 22, 2002 Lee
20020160821 October 31, 2002 Kaikati et al.
20020167923 November 14, 2002 Sendonaris et al.
20030020538 January 30, 2003 Kim
20030087617 May 8, 2003 Shohara
20030133518 July 17, 2003 Koomullil et al.
20030161487 August 28, 2003 Husted et al.
20030179830 September 25, 2003 Eidson et al.
20030197558 October 23, 2003 Bauder et al.
20030197559 October 23, 2003 Ghannouchi et al.
20030215025 November 20, 2003 Hietala
20030215026 November 20, 2003 Hietala
20030227342 December 11, 2003 Liu
20040072597 April 15, 2004 Epperson et al.
20040121741 June 24, 2004 Rashev et al.
20040131129 July 8, 2004 Harron et al.
20040183511 September 23, 2004 Dening
20040198414 October 7, 2004 Hunton
20040208157 October 21, 2004 Sander et al.
20050002470 January 6, 2005 Saed et al.
20050018765 January 27, 2005 Endres et al.
20050195919 September 8, 2005 Cova
20060071711 April 6, 2006 Persson et al.
20060203899 September 14, 2006 Gee
20060280502 December 14, 2006 Sekine et al.
20070110199 May 17, 2007 Momtaz et al.
20070190952 August 16, 2007 Waheed et al.
20080219332 September 11, 2008 Brehler
20080310617 December 18, 2008 Daecke et al.
20090252255 October 8, 2009 Lee et al.
Other references
  • Volder, Jack E., “The CORDIC Trigonometric Computing Technique,” IRE Trans. On Elect. Computers p. 330, Sep. 1959.
  • Andraka, Ray, “A Survey of Cordic Algorithms for FPGA Based Computers,” Association for Computing Machinery, 0-89791-978-5, 1998.
  • Pinto et al., “Phase Distortion and Error Vector Magnitude for 8-PSK Systems, ” London Communications Symposium, Sep. 14-15, 2000, University College London, London, England.
  • Johnson, Jackie, “Power Amplifier Design for Open Loop Edge Large Signal Polar Modulation Systems,” RFDesign, Jun. 2006, pp. 42-50.
  • Cusinato, Paolo, “Gain/Bandwidth Programmable PA Control Loop for GSM/GPRS Qud-Band Cellular Handsets,” IEEE Journal of Solid-State Circuits, Jun. 2004, pp. 960-966, vol. 39, No. 6, IEEE.
Patent History
Patent number: 7813446
Type: Grant
Filed: May 16, 2002
Date of Patent: Oct 12, 2010
Patent Publication Number: 20030215025
Assignee: RF Micro Devices, Inc. (Greensboro, NC)
Inventor: Alexander Wayne Hietala (Phoenix, AZ)
Primary Examiner: David C. Payne
Assistant Examiner: Linda Wong
Attorney: Withrow & Terranova, P.L.L.C.
Application Number: 10/147,569