Adaptive optical equalization for chromatic and/or polarization mode dispersion compensation

Abstract

An adaptive optical parallel equalizer architecture is based on a controllable optical modulator device to realize an optical FIR (finite-impulse-response) filter including a plurality of coefficient taps in order to have independent control of each optical FIR filter coefficient. A unique adaptive opto-electronic LMS (least mean squares) process is utilized to generate an electronic error signal utilized to control the plurality of parallel tap coefficients of the optical parallel equalizer. The electronic error signal is used as the optimization criterion because the electronic signal after photo-detection is needed to achieve any measurable performance in terms of bit error rate (BER). In a specific embodiment, the controllable optical parallel FIR filter is realized by employing an optical vector modulator. The optical vector modulator is realized by splitting a supplied input optical signal into a plurality of parallel similar optical signals, controllably adjusting the phase and/or amplitude of each of the plurality of optical signals and delaying the resulting optical signals in a prescribed manner relative to one another. Then, the “delayed” signals are combined to yield the optical signal comprising the vector modulated input optical signal to be transmitted as an output. In one particular embodiment, both the phase and amplitude is adjusted of each of the plurality of parallel optical signals, and the error control signals for effecting the adjustments are generated in response to the optical modulator optical output signal utilizing the unique Opto-Electronic LMS process.

Description

TECHNICAL FIELD

This invention relates to optical transmission systems and, more particularly, to optical equalization.

BACKGROUND OF THE INVENTION

Intersymbol interference (ISI) is a problem commonly encountered in high-speed fiber-optic communication systems. This ISI problem can introduce bit errors and thus degrade the system performance and reliability. It is typically caused by two major impairment sources: chromatic dispersion (sometimes called group velocity dispersion or GVD) and polarization mode dispersion (PMD). Another source of transmission impairments is optical noise.

In a fiber-optic link, a number of optical amplifiers are employed to strengthen the optical signal, but at the same time add in incoherent amplified spontaneous emission (ASE) noise (commonly called optical noise).

Because of the frequency-dependent propagation constant in optical fibers, different spectral components of a pulse travel at slightly different velocities, resulting in pulse broadening in the optical domain. Two parameters are commonly used to characterize first-order and second-order chromatic dispersion (GVD) of a fiber: a dispersion parameter, in ps/km/nm, and a dispersion slope parameter, in ps/km/nm². GVD of any order is linear in the optical domain but becomes nonlinear after square-law photo-detection. Usually chromatic dispersion is static and can be effectively compensated by a dispersion compensation module (DCM) comprised of specialty fibers and other passive components. However, a DCM is usually expensive and may add unwanted latency in the optical link that causes a drop in the network quality of service (QoS). It is also possible that residual chromatic dispersion remains even after employing a DCM in the optical ink, and is desirably compensated for by an equalizer. Therefore, for the purpose of evaluating the performance of an adaptive equalizer, the first-order chromatic dispersion is specified in terms of ps/nm without explicitly specifying the fiber type and transmission distance.

Polarization mode dispersion (PMD) is caused by different travelling speeds of two orthogonal polarization modes due to fiber birefringence. Fiber birefringence originates from non-circularity of the fiber core and can also be induced by stress, bending, vibration, and so on. Thus, PMD is dynamic in nature and drifts slowly over time. PMD can be modeled as dispersion along randomly concatenated birefringent fiber segments through mode coupling between neighboring sections. Differential group delay (DGD) is the parameter used to characterize the PMD-induced pulse broadening and follows a Maxwellian distribution. As a result of this variability, the PMD of a fiber is usually characterized by the mean DGD parameter in terms of ps/sqrt(km). In addition, PMD is frequency-dependent. First-order PMD is the frequency-independent component of this frequency-dependent PMD. Second-order (or higher-order) PMD is frequency-dependent and has an effect similar to chromatic dispersion on pulse broadening.

To evaluate the performance of an equalizer, the instantaneous DGD is used instead to describe the delay between the fast and slow orthogonal polarization modes (in particular, the principal states of polarization or PSPs of a fiber). In the worst-case scenario, the input power is split equally between these two orthogonal polarization modes, i.e., the power-splitting ratio=0.5. The performance against the first-order instantaneous DGD (frequency-independent dispersion component) in ps is essential in evaluating the effectiveness of a dispersion compensator. Since these two polarization modes are orthogonal to each other, the photo-current I(t) at the photo-detector is proportional to the summation of the optical power in each polarization. Thus, first-order PMD creates linear ISI at the output of the photo-detector.

Optical equalizers have been used in attempts at compensating for these impairments. The most common form of these equalizers is a cascaded structure, which tends to have less flexibility in control of filter parameters.

In controlling these optical equalizers, usually non-adaptive equalization approaches-are used, but it has been shown that adaptive control algorithms provide good performance improvement. One such adaptive scheme is to monitor the frequency component(s) of the electronic signal. Other adaptive approaches involve nonlinear least squares optimization of criteria such as minimum mean-square-error (MSE), minimization of ISI, or maximization of eye openings. This requires the use of the modified Gauss-Newton or Levenberg-Marquardt methods, which are iterative and incapable of tracking fast change of channel condition.

SUMMARY OF THE INVENTION

These and other problems and limitations of prior known optical equalization arrangements are overcome in applicants' unique invention by employing a parallel adaptive equalizer architecture based on a controllable optical modulator device to realize an optical FIR (finite-impulse-response) filter including a plurality of parallel coefficient taps in order to have independent control of each optical filter coefficient.

Additionally, a unique adaptive opto-electronic LMS (least mean squares) process is utilized to generate an electronic error signal utilized to control the plurality of parallel tap coefficients of the parallel optical equalizer. The electronic error signal is used as the optimization criterion to generate control signals to adapt the adaptive optical equalizer because the electronic signal after photo-detection is needed to achieve any measurable performance in terms of bit error rate (BER).

In a specific embodiment of the invention, the controllable optical parallel FIR filter is realized by employing an optical vector modulator. The optical vector modulator is realized by splitting a supplied input optical signal into a plurality of similar parallel optical signals, controllably adjusting the phase and/amplitude of each of the plurality of optical signals and delaying the resulting optical signals in a prescribed manner relative to one another. Then, the “delayed” signals are combined to yield the signal comprising the vector modulated input optical signal to be transmitted as an output. Wherein a signal can be “delayed” by a zero (0) delay interval.

In one particular embodiment, both the phase and amplitude is adjusted of each of the plurality of parallel optical signals, and the error control signals for effecting the adjustments are generated in response to the optical modulator output signal utilizing the unique Opto-Electronic LMS process.

BRIEF DESCRIPTION OF THE DRAWING

FIG. 1 shows, in simplified block diagram form, one embodiment of the invention;

FIG. 2 shows, in simplified block diagram form, details of a controllable optical modulator that may be employed in the practice of the invention of the invention; and

FIG. 3 illustrates, in simplified block diagram form, details of another embodiment of the invention.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

FIG. 1 shows, in simplified block diagram form, one embodiment of the invention. Specifically, shown is optical light input terminal to which an optical input signal from an optical channel is supplied. Exemplary optical carrier signals to be processed have optical frequencies of about 2.3×10¹⁴ Hertz to about 1.8×10¹⁴ Hertz, i.e., a wavelength of about 1.3 microns to about 1.7 microns. In one example, an optical carrier signal having a wavelengti of approximately 1.55 micronns, i.e., a frequency of 1.93×10¹⁴ Hertz is supplied via input terminal 101 to controllable optical modulator 102. Also supplied to controllable optical modulator 102, via circuit path 112, is error signal e(k), which is used to phase and/or amplitude modulated, i.e., vector modulate the supplied optical signal from input terminal 101 to generate the desire transport signal at output terminal 103. As indicated above, controllable optical modulator 102 is essentially a controllable optical FIR filter. One embodiment of an optical FIR filter that may be advantageously employed as controllable optical modulator 102 in the embodiment of the invention of FIG. 1 is shown in FIG. 2 and described below. As indicated above, other embodiments for optical modulator 102 may also be equally employed in practicing the invention. One such embodiment is an array of optical waveguide gratings.

For a received optical signal E(t) supplied to controllable modulator 102 via input terminal 101 the output optical signal E_o (t) from controllable modulator 102 at output terminal 103 is $\begin{array}{cc}{E}_O\left(t\right)=\sum _{i=1}^n{\alpha }_i{e}^{j{\theta }_i}E\left(t-{\tau }_i\right)=\sum _{i=1}^n{c}_{i}E\left(t-{\tau }_i\right),& \left(1\right)\end{array}$
where n is the number of taps for the optical equalizer, α_i is amplitude parameter, θ_i and c_i=α_ie^jθ, is the i^th filter coefficient. In one embodiment, for a tap delay of 1/ƒ_s, τ_i=(i−1)/ƒ_s for i=1, . . . , n. The optical output signal E_o(t) from controllable modulator 102 is transported to an optical receiver and therein to photodiode 104. As is well known, photodiode 104 is a square-law detector and generates a current |q(k)|² in response to detection of E_o(t). Transimpedance amplifier 105 converts the current from photodiode 104 to a voltage signal, in well known fashion. The electronic voltage signal from transimpedance amplifier 105 is supplied to slicer unit 106 and to a negative input of algebraic combiner, i.e., algebraic adder 108. An automatic threshold control signal is also supplied to slicer unit 106. The threshold control is such as to slice the voltage signal from transimpedance amplifier 105 in such a manner to realize a desired output level from slicer 106. The output from slicer 106 is the desired compensated received data signal {circumflex over (d)} (k) and is supplied as an output from the receiver and to a positive input to algebraic adder 108. The error signal output from alegbraic combiner 108 is supplied to WUD ({overscore (α)}, {overscore (θ)}) unit 109, where the electronic control signal amplitude ({overscore (α)}) and phase ({overscore (θ)}) values are generated, in accordance with an aspect of the invention, utilizing the unique opto-electronic LMS process. The amplitude ({overscore (α)}) values and phase ({overscore (θ)}) values are supplied via circuit path 110 to adjust the tap coefficients in controllable modulator 102. Note that although a single circuit path 110 is shown, it will be understood that as many circuit paths are included equal to the number of controllable taps or legs included in controllable optical modulator 102. In this example, there may be N such circuit paths. Again, the values of ({overscore (α)}) and ({overscore (θ)}), in this embodiment of the invention, are generated in accordance with the unique opto-electronic LMS process. It is further noted that when only the amplitude of the received optical signal is modulated only the amplitude adjustment values ({overscore (α)}) are supplied from unit 109 to controllable optical modulator 102. Similarly, when only the phase of the received optical signal is being modulated only the phase adjustment values ({overscore (θ)}) are supplied from unit 109 to controllable optical modulator 102. Finally, when both the amplitude and phase of the received optical signal are being modulated both the amplitude adjustment values ({overscore (α)}) and the phase adjustment values ({overscore (θ)}) are supplied from unit 109 to controllable optical modulator 102.

Not shown in the above embodiment is the typical clock data recovery circuitry (CDR). Just before the CDR, an uncompensated detected signal may contain a certain amount of ISI induced by optical impairments along the optical path, such as GVD and PMD. To remove the ISI present in the electronic signal before recovering the bit stream, a coefficient-updating process is employed, in accordance with the invention, to control controllable optical modulator 102. Operating in the optical domain, this process, however, minimizes the electronic error between the compensated signal and the desired signal in the mean square sense in a similar fashion to the least-mean-square (LMS) algorithm for pure electronic equalization.

Thus, the ISI elimination process in this invention utilizes a unique opto-electronic LMS process.

FIG. 2 shows, in simplified block diagram form, details of one optical vector modulator that may be utilized as controllable optical modulator 102 employed in FIG. 1 in the embodiment of the invention. The optical vector modulator 102 is based on the summing of multiple optical tapped delay lines. The principle of operation is as follows: The to be phase-shifted and/or to be amplitude-modulated input optical signal E(t) is an optical carrier. Input optical signal E(t) is supplied to optical vector modulator 102 via input terminal 101 where it is split via input multimode interference (MMI) coupler 201 into a plurality of similar branches. Input MMI 102 is essentially a power splitter. Each of the plurality of branches is equipped with an amplitude and/or phase modulator 202-1 through 202-N to adjust the amplitude and/or phase of the input optical carrier E(t). In this example, not to be construed as limiting the scope of the invention, both the amplitude and phase is adjusted in each branch of the optical vector modulator. Each of the amplitude and phase modulators 202-1 through 202-N is followed by an optical delay line, namely, delay units 203-1 through 203-N, respectively. The delays in each of the modulator branches including phase modulators 202-1 through 202-N are generated by delay units 203-1 through 203-N respectively. Each of these delay lines in delay units 203-1 through 203-N changes the phase of the sub-carrier of the optical signal from amplitude and phase modulators 201-1 through 201-N, respectively, by a fixed amount. For example, the delay line in unit 203-1 provides a delay of τ, delay unit 203-2 provides a delay of 2 τ and delay unit 203-N provides a delay of N τ. Typically, a delay τ of 1/(N* carrier frequency) is required. In an embodiment in, which delay unit 203-1 supplies a zero (0) delay interval, delay unit 203-2 supplies a delay of τ and so on until delay unit 203-N supplies a delay of τ(N−1). Thus, if the carrier frequency is 40 GHz, the delay range should be 0, . . . , 25 picoseconds (ps). Delay τ can be equal to one (1) bit period, i.e., T=25 ps for the instance of 40 Gbps. Therefore, the delay range is 0, . . . , τ(N−1). Alternatively, delay τ can be a fraction of a bit period, for example, T/2=12.5 ps. for 40 Gbps. Thus, for the example that τ=/2=12.5 ps., the delay range is 0, . . . , (N−1)* 12.5 ps. Another MMI 204 coupler, which is for example a power combiner, combines all of the amplitude and phase adjusted, and delayed optical signals from all branches to produce a modulated output optical signal at output 106, which will interfere constructively or destructively depending on the summing optical phases from all tributary branches. Therefore, by interfering signals with different carrier phase, the phase and the amplitude of the carrier of the summing signal can be set to an arbitrary state. These interfered optical carriers will produce microwave phasors with prescribed amplitude and phase at the remote optical detector, namely, photodiode 104 of FIGS. 1 and 3.

The amplitude and phase modulator 202 of each branch can be fabricated, for example, in a material system with linear electro-optic effect, as InP, GaAs or LiNbO₃. The effective refractive index of an optical waveguide changes in proportion to the applied electrical field perpendicular to this waveguide. A high frequency distributed electrical waveguide is engineered to co-propagate with the optical wave with matched propagating velocity to deliver the local electrical field with high modulation bandwidth. The different branches will delay the optical signal by a different length of time. This results in different sub-carrier phases at the outputs of these delay lines in units 203. In the combiner 204, these different output signals that interfere constructively have a different carrier phase due to the different time delays these signals experienced. The carrier of the signal after the MMI coupler, i.e., power combiner 204, is the sum of all carriers of the signals that interfere constructively.

FIG. 3 shows, in simplified form, details of another embodiment of the invention. The embodiment of the invention illustrated in FIG. 3 is similar to that shown in FIG. 1 except it specifically employed the optical vector modulator shown in FIG. 2 for controllable optical modulator 102 of FIG. 1., and that it employs interferometer 113 (FIG. 3) for generating a signal employed in the opto-electronic LMS process. Thus, elements similar to those shown in FIG. 1 have been similarly numbered and will not be described again in detail.

In the embodiment of FIG. 3 an optical interferometer 113 is supplied via optical path 111 with the optical signal supplied via input 101 to optical vector modulator 102, and via optical path 112 with the output optical signal at output 103 of optical vector modulator 102. As is well known, optical interferometer 113 in response to the supplied optical signals develops optical output signals, which are representative of the sum and difference of the supplied optical signals from controllable optical modulator 102. These sum and difference signals are supplied to photodiodes 114 and 115. Photodiodes 114 and 115 generate electronic signals which are supplied to differential amplifier 116, which generates a correlated signal of the optical vector modulator 102, i.e., the optical FIR filter, input signal and output q*(k)r(k+i) signal, as described below in relation to Equation (5) which is supplied to WUD ({overscore (α)}, {overscore (θ)}) unit 109. The “*” denotes the complex conjugate.

Operation of this embodiment of the invention, is described for an incoming optical signal E(t) of a single polarization is sampled at a sampling rate ƒ_s=1T_s equal to or being a multiple of the bit rate ƒ_b. When ƒ_s=ƒ_b, controllable optical modulator 102 (which is essentially a FIR filter having a plurality of parallel legs) is synchronous (SYN). On the other hand, when ƒ_s is a multiple of the bit rate ƒ_b, controllable optical modulator 102 is said to be fractionally spaced (FS). Denote the sampled data vector as {right arrow over (r)}(k)=[r(k+L) . . . r(k−L)]^T, where r(k)=E(kT_s) and the superscript T denote a transpose function. The controllable optical modulator 102 coefficient vector of a length N=2L+1 is denoted as {right arrow over (c)}(k)=[c_−L(k), . . . , c_i(k), . .. ,C_L(k)]^T, where the coefficient indices are rearranged to i=−L, . . . ,L to center the middle tap of controllable optical modulator 102 for the sake of “easy” mathematical manipulation. It should be noted that {right arrow over (c)}(k) is complex in general. The output of controllable optical modulator 102 is then q(k){right arrow over (=)}{right arrow over (c)}^H(k){right arrow over (r)}^H(k)=Σ_i=−L^Lc_i*(k)r(k−i). Here the superscript H implies conjugate transpose and H=T* implies complex conjugate transpose Then, photodetector 104 (FIG. 1, FIG. 3) converts the optical output signal q(k) from controllable optical modulator 102 to an electronic signal, namely, |q(k)|²=q(k)q*(k)={right arrow over (c)}_H(k)R(k){right arrow over (c)}^H(k), where R(k)={right arrow over (r)}(k){right arrow over (r)}^H(k). It can be shown that R(k) is a Hermitian matrix and, therefore, can be diagonalized by a unitary matrix.

Error signal e(k) is generated in conjunction with the output from TIA 105 |q(k)|² and the output from slicer 106 {circumflex over (d)}(k) being supplied to the negative and positive inputs, respectively, of algebraic combiner, i.e., adder, 108 (FIG. 1, FIG. 3), namely, e(k)={circumflex over (d)}(k)−|q(k)|². It is noted that {circumflex over (d)}(k) is generated during normal operation of the invention and is the desired output. It is further noted that although not specifically shown here, it is well known that a training sequence can be employed to train controllable optical modulator 102 of FIG. 1 and optical vector modulator 102 of FIG. 3 or any other arrangement that may be employed to realize the desired FIR filter function.

The unique opto-electronic LMS process tends to minimize deterministically the cost function defined here as J(k)=|e(k)|². Therefore, taking a step in the negative gradient direction for minimizing the cost function, the opto-electronic LMS process determines the optimized {right arrow over (c)} recursively as follows: $\begin{array}{cc}\overrightarrow{c}\left(k+1\right)=\overrightarrow{c}\left(k\right)-\frac{\beta }{4}\nabla c\left\{\left|e\left(k\right)\right|\right\},& \left(2\right)\end{array}$
where β is a preset step size and ∇c{[e(k)]²} is the gradient of the cost function. In this example, ∇c{[e(k)]²}=2e(k)∇c{e(k)}=−2e(k)∇c{{right arrow over (c)}^H(k)R(k){right arrow over (c)}(k)}. Since it can be shown that ∇c{{right arrow over (c)}^H(k)R(k){right arrow over (c)}(k)}=2R(k){right arrow over (c)}(k), the opto-electronic LMS process updates the FIR coefficients in the manner that follows: $\begin{array}{cc}\begin{array}{c}\overrightarrow{c}\left(k+1\right)=\overrightarrow{c}\left(k\right)+\beta e\left(k\right)R\left(k\right)\overrightarrow{c}\left(k\right)\\ =\overrightarrow{c}\left(k\right)+\beta e\left(k\right)q^{*}\left(k\right)\overrightarrow{r}\left(k\right).\end{array}& \begin{array}{c}\left(3\right)\\ \left(4\right)\end{array}\end{array}$
Thus, the i^th FIR filter coefficient is updated as follows:
c_i(k+1)=c_i(k)+βe(k)q*(k)r(k+i).  (5)
The additional product term q* (k) results directly from the square-law detection via photodetector 104 converting the optical signal output from controllable modulator (optical vector modulator) 102 to an electronic signal. In other words, the inner product q*(k)r(k−i) between the un-equalized and equalized signals is used for the adjustment of the coefficients of controllable optical modulator 102. Alternatively, in equation (3), the sole information required for optical equalization is the optical input correlation matrix R, since the FIR filter coefficients {right arrow over (c)} are already known. To obtain the correlated signal of q(k) and r(k−i), interferometer 113 (FIG. 3) is employed. To this end, the optical input signal E(t) to and the optical output signal E₀(t) from controllable optical modulator 102 (optical vector modulator (FIG.3)) are supplied to first and second inputs, respectively, of optical interferometer 1 13. In known fashion, optical interferometer 113 generates optical signals at its outputs, which are representative of the sum and difference of the supplied optical signals from optical vector modulator 102. These optical sum and difference signals are supplied to photodiodes 114 and 115, respectively. Photodetectors 114 and 115, which are photodiodes, convert the optical output from optical interferometer 113 to electronic signals. These electronic signals are supplied to differential amplifier 116 that generates a difference signal, which is supplied to WUD({overscore (α)},{overscore (θ)}) 109 for use in generating the amplitude and phase control signals {overscore (α)},{overscore (θ)}, respectively, for each leg, i.e., tap, of optical vector modulator 102.

The above discussion assumes a polarized incoming optical signal E(t) and, thus, leads to a single-polarization opto-electronic LMS process, which can effectively mitigate GVD-induced ISI. However, for the instance of first-order PMD, two orthogonal polarizations and involved, namely, E_V(t) and E_H(t) representing the optical signals of vertical and horizontal polarizations, respectively. In consideration of both the vertical and horizontal polarizations, the electronic output from photodiode 104 is |q(k)|²=|q_V(k)|₂+|q_H(k)|², where q_V(k)={right arrow over (c)}^H(k){right arrow over (r)}_V(k) and q_H(k)={right arrow over (c)}^H(k){right arrow over (r)}_H(k) under the assumption of the optical FIR filter, i.e., optical vector modulator 102, of FIG. 3, being insensitive to polarization, i.e., {right arrow over (c)}_V={right arrow over (c)}_H={right arrow over (c)}. Hence, q(k)={right arrow over (c)}^H(k)[R_V(k)+R_H(k)]{right arrow over (c)}(k) and ∇c{[e(k)]²}=2e(k)∇c{e(k)}=−4e(k)[R_V(k)+R_H(k)]{right arrow over (c)}(k). Thus, the opto-electronic LMS process tap weight-date procedure becomes: $\begin{array}{cc}\begin{array}{c}\overrightarrow{c}\left(k+1\right)=\overrightarrow{c}\left(k\right)+\beta e\left(k\right)\left[R_V\left(k\right)+R_H\left(k\right)\right]\overrightarrow{c}\left(k\right)\\ =\overrightarrow{c}\left(k\right)+\beta e\left(k\right)\left[q_V^{*}\left(k\right){\overrightarrow{r}}_V\left(k\right)+q_H^{*}\left(k\right){\overrightarrow{r}}_H\left(k\right)\right]\end{array}& \begin{array}{c}\left(6\right)\\ \left(7\right)\end{array}\end{array}$
In scalar form, the i^th FIR filter tap coefficient is updated as follows:
c_i(k+1)=c_i(k)+βe(k)[q*_V(k)r_V(k−i)+q*_H(k)r_H(k−i)}.  (8)
If we denote
{right arrow over (q)}(k)={q_V(k),q_H(k)}^T,{right arrow over (u)}(k−i)=[r_V(k−i),r_H(k−i)}^T,
then,
c_i(k+1)=c_i(k)+βe(k){right arrow over (q)}^H(k){right arrow over (u)}(k−i).  (9
Here
{right arrow over (q)}^H(k){right arrow over (u)}(k−i)=∥{right arrow over (q)}(k)∥∥(k−i)∥∥cos (θ_q,u),
where ∥{right arrow over (q)}∥ is the Euclidean norm of {right arrow over (q)} and θ_q,n is the angle between {right arrow over (q)} and {right arrow over (u)}. In both equations (5) and (9), the knowledge of the inner product of the input {right arrow over (u)} and the equalized {right arrow over (q)} is required for the optimization of the optical FIR filter coefficients. Note that once the values for all c_i are known, the corresponding values for {right arrow over (α)}_i and {right arrow over (θ)}_i are readily generated, since c_i=α_ie^jθ_i, as shown in Equation (1) above.

The above-described embodiments are, of course, merely illustrative of the principles of the invention. Indeed, numerous other methods or apparatus may be devised by those skilled in the art without departing from the spirit and scope of the invention. Specifically, other arrangements may be equally employed for realizing the controllable optical FIR filter.

Claims

1. Apparatus for use in an adaptive optical equalizer comprising:

a controllable optical modulator having an input and an output, and being coupled to receive an incoming optical signal and configured to generate an output optical signal by phase modulation and/or amplitude modulation of the received optical signal, said controllable optical modulator including a plurality of similar optical signals in a corresponding plurality of optical paths, each of said parallel optical paths including an opto-electronic controller responsive to electronic control signals for effecting said phase modulation and/or amplitude modulation of said optical signal being transported in said optical path; and

a control signal generator responsive to an optical output signal from said output of said controllable modulator for generating said electronic control signals in accordance with predetermined criteria.

2. The apparatus as defined in claim 1 wherein said controllable optical modulator comprises arrayed waveguide gratings.

3. The apparatus as defined in claim 1 wherein said control signal generator is configured to update said adjustable control signals at a predetermined sampling rate.

4. The apparatus as defined in claim 1 wherein said controllable optical modulator is configured to operate as a controllable optical finite impulse response (FIR) filter.

5. The apparatus as defined in claim 4 wherein said FIR filter includes a plurality of parallel adjustable taps.

6. The apparatus as defined in claim 1 wherein said controllable optical modulator is configured to operate as a controllable optical vector modulator.

7. The apparatus as defined in claim 6 wherein said controllable optical vector modulator includes an optical splitter having an input and a plurality of outputs for splitting said received optical signal into a plurality of similar optical signals which are supplied on a one-to-one basis to said plurality of optical vector modulator output ports, an optical combiner having a plurality of inputs and an output and a plurality of controllable waveguides coupling predetermined ones of said optical splitter outputs to said optical combiner input ports, and wherein said opto-electronic control signals are supplied to control at least one of said plurality of controllable waveguides.

8. The apparatus as defined in claim 1 wherein said predetermined criteria includes an opto-electronic least means square (LMS) process.

9. The apparatus as defined in claim 8 wherein said control signal generator includes an optical detector having prescribed characteristics and being responsive to said received optical signal for generating a version of said received optical signal and apparatus utilizing said version of said received optical signal in accordance with said opto-electronic LMS process to generate amplitude and/or phase adjustment values for each tap of said controllable optical modulator.

10. The apparatus as defined in claim 9 further including an interferometer supplied with said optical input signal to and said optical output signal from said optical vector modulator, a differential amplifier supplied with electronic versions of outputs from said interferometer for generating a difference signal, said difference signal being supplied to said utilization apparatus to be employed in said opto-electronic LMS process to generate said amplitude and/or phase adjustment values.

11. The apparatus as defined in claim 9 wherein said optical detector includes a photodiode for generating a current version of said received optical, and said signal utilization apparatus includes an amplifier for generating a voltage version of said current signal, a slicer for generating a sliced version of said voltage signal in accordance with a supplied adjustable threshold level, an algebraic combiner supplied with said voltage signal and an output from said slicer for generating an error signal and a control signal generator responsive to said error signal for generating said amplitude and/or phase adjustment signals in accordance with said opto-electronic LMS process.

12. A method for use in an adaptive optical equalizer including a controllable optical modulator comprising the steps of:

adaptively controlling said controllable optical modulator to modulate a supplied optical signal to generate an equalized optical output signal;

converting, in accordance with predetermined first criteria, said equalized optical output signal to an electronic signal version;

utilizing said electronic signal version to generate, in accordance with second predetermined criteria, amplitude and/or phase control signals;

feeding back said control signals to adaptively control said controllable optical modulator; and

employing each control signal to adjust the amplitude and/or phase of a corresponding optical signal propagating on a corresponding optical waveguide of a parallel array of waveguides of said controllable optical modulator.

13. The method as defined in.claim 12 wherein said controllable optical modulator is configured to operate as a controllable optical finite impulse response (FIR) filter, and wherein said parallel array of waveguides form parallel optical taps of said controllable optical FIR filter.

14. The method as defined in claim 12 wherein said controllable optical modulator is configured to operate as a controllable optical vector modulator.

15. The method as defined in claim 14 wherein in operation of said controllable optical vector modulator a supplied optical signal is split into a plurality of similar optical signals corresponding in number to a number of said parallel waveguides in said array, said plurality of similar optical signals being supplied on a one-to-one basis to said plurality of optical waveguides, in response to said control signals adjusting the amplitude and/or phase of said optical signals being transported in said optical waveguides, optically combining said adjusted optical signals from said plurality of optical waveguides to generate said equalized optical output signal.

16. The method as defined in claim 1 wherein said second predetermine criteria includes use of an opto-electronic least means square (LMS) process.

17. The method as defined in claim 16 wherein said control signals are generated by optical detecting said output optical signal from said controllable optical modulator to generate an electronic signal version, and processing said electronic signal version in accordance with said opto-electronic LMS process to generate said amplitude and/or phase control signals.

18. The method as defined in claim 17 further including the steps of generating sum and difference optical signals of an optical signal supplied to the input of said controllable optical modulator and said optical output signal generated by said controllable optical modulator, converting said sum and difference optical signals into electronic versions, differentially amplifying said sum and difference signal versions to generate a difference signal, utilizing said difference signal in said opto-electronic LMS process for generating said amplitude and/or phase adjustment control signals.

19. The method as defined in claim 17 wherein said optical detecting includes utilizing a photodiode to convert said output signal from said controllable optical modulator to an electronic signal having predetermined characteristics, converting said photo diode output signal into a voltage electronic signal, slicing said voltage signal in accordance with a supplied threshold level, algebraically combining said voltage signal and a sliced version of said voltage signal to generate an error signal and utilizing said error signal in said opto-electronic LMS process for generating said amplitude and/or phase adjustment control signals.

20. The method as defined in claim 12 wherein said controllable optical modulator comprises arrayed waveguide gratings.