SINGLE USER DETECTION BASE STATION

Info

Publication number: 20070019590
Type: Application
Filed: Sep 12, 2006
Publication Date: Jan 25, 2007
Patent Grant number: 7414999
Applicant: INTERDIGITAL TECHNOLOGY CORPORATION (Wilmington, DE)
Inventors: Parthapratim De (Camp Bell, CA), Ariela Zeira (Huntington, NY), Jung-Lin Pan (Selden, NY), Raj Mani Misra (Fremont, CA)
Application Number: 11/530,919

Abstract

A transmitter site transmits a plurality of data signals over a shared spectrum in a code division multiple access communication system. Each transmitted data signal experiences a similar channel response. A combined signal of the transmitted data signals is received. The combined signal is sampled at a multiple of the chip rate. The channel response for the combined signal is determined. A first element of a spread data vector is determined using the combined signal samples and the estimated channel response. Using a factor from the first element determination, remaining elements of the spread data vector are determined. The data of the data signals is determined using the determined elements of the spread data vector.

Description

Description

This application is a continuation of U.S. patent application Ser. No. 10/052,943, filed on Nov. 7, 2001, which claims priority to U.S. Provisional Patent Application No. 60/246,947, filed on Nov. 9, 2000, which applications are incorporated herein by reference.

BACKGROUND

The invention generally relates to wireless communication systems. In particular, the invention relates to data detection in a wireless communication system.

FIG. 1 is an illustration of a wireless communication system 10. The communication system 10 has base stations 12₁to 12₅(12) which communicate with user equipments (UEs) 14₁to 14₃(14). Each base station 12 has an associated operational area, where it communicates with UEs 14 in its operational area.

In some communication systems, such as code division multiple access (CDMA) and time division duplex using code division multiple access (TDD/CDMA), multiple communications are sent over the same frequency spectrum. These communications are differentiated by their channelization codes. To more efficiently use the frequency spectrum, TDD/CDMA communication systems use repeating frames divided into time slots for communication. A communication sent in such a system will have one or multiple associated codes and time slots assigned to it. The use of one code in one time slot is referred to as a resource unit.

Since multiple communications may be sent in the same frequency spectrum and at the same time, a receiver in such a system must distinguish between the multiple communications. One approach to detecting such signals is multiuser detection. In multiuser detection, signals associated with all the UEs 14, users, are detected simultaneously. Approaches for implementing multiuser detection include block linear equalization based joint detection (BLE-JD) using a Cholesky or an approximate Cholesky decomposition. These approaches have a high complexity. The high complexity leads to increased power consumption, which at the UE 14₁results in reduced battery life. Accordingly, it is desirable to have alternate approaches to detecting received data.

SUMMARY

A code division multiple access user equipment is used in receiving a plurality of data signals over a shared spectrum. Each received data signal experiences a similar channel response. A combined signal of the received data signals is received over the shared spectrum. The combined signal is sampled at a multiple of the chip rate. A channel response is estimated as a channel response matrix at the multiple of the chip rate. A padded version of a spread data vector of a size corresponding to the multiple chip rate using a column of the channel response matrix, the estimated channel response matrix, the samples and a fourier transform. The spread data vector is estimated by eliminating elements of the padded version so that the estimated spread data vector is of a size corresponding to the chip rate.

BRIEF DESCRIPTION OF THE DRAWING(S)

FIG. 1 is a wireless communication system.

FIG. 2 is a simplified transmitter and a single user detection receiver.

FIG. 3 is an illustration of a communication burst.

FIG. 4 is a flow chart of an extended forward substitution approach to single user detection (SUD).

FIG. 5 is a flow chart of an approximate banded Cholesky approach to SUD.

FIG. 6 is a flow chart of a Toeplitz approach to SUD.

FIG. 7 is a flow chart of a fast fourier transform (FFT) approach applied to the channel correlation matrix for SUD.

FIG. 8 is a flow chart of a FFT approach to SUD using effective combining.

FIG. 9 is a flow chart of a FFT approach to SUD using zero padding.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

FIG. 2 illustrates a simplified transmitter 26 and receiver 28 using single user detection (SUD) in a TDD/CDMA communication system, although the single user detection approaches are applicable to other systems, such as frequency division duplex (FDD) CDMA. In a typical system, a transmitter 26 is in each UE 14 and multiple transmitting circuits 26 sending multiple communications are in each base station 12. The SUD receiver 28 may be at a base station 12, UEs 14 or both. SUD is typically used to detect data in a single or multicode transmission from a particular transmitter. When all the signals are sent from the same transmitter, each of the individual channel code signals in the multicode transmission experience the same channel impulse response. SUD is particularly useful in the downlink, where all transmissions originate from a base station antenna or antenna array. It is also useful in the uplink, where a single user monopolizes a timeslot with a single code or multicode transmission.

The transmitter 26 sends data over a wireless radio channel 30. A data generator 32 in the transmitter 26 generates data to be communicated to the receiver 28. A modulation/spreading sequence insertion device 34 spreads the data and makes the spread reference data time-multiplexed with a midamble training sequence in the appropriate assigned time slot and codes for spreading the data, producing a communication burst or bursts.

A typical communication burst 16 has a midamble 20, a guard period 18 and two data bursts 22, 24, as shown in FIG. 3. The midamble 20 separates the two data fields 22, 24 and the guard period 18 separates the communication bursts to allow for the difference in arrival times of bursts transmitted from different transmitters 26. The two data bursts 22, 24 contain the communication burst's data.

The communication burst(s) are modulated by a modulator 36 to radio frequency (RF). An antenna 38 radiates the RF signal through the wireless radio channel 30 to an antenna 40 of the receiver 28. The type of modulation used for the transmitted communication can be any of those known to those skilled in the art, such as quadrature phase shift keying (QPSK) or M-ary quadrature amplitude modulation (QAM).

The antenna 40 of the receiver 28 receives various radio frequency signals. The received signals are demodulated by a demodulator 42 to produce a baseband signal. The baseband signal is sampled by a sampling device 43, such as one or multiple analog to digital converters, at the chip rate or a multiple of the chip rate of the transmitted bursts. The samples are processed, such as by a channel estimation device 44 and a SUD device 46, in the time slot and with the appropriate codes assigned to the received bursts. The channel estimation device 44 uses the midamble training sequence component in the baseband samples to provide channel information, such as channel impulse responses. The channel impulse responses can be viewed as a matrix, H. The channel information is used by the SUD device 46 to estimate the transmitted data of the received communication bursts as soft symbols.

The SUD device 46 uses the channel information provided by the channel estimation device 44 and the known spreading codes used by the transmitter 26 to estimate the data of the desired received communication burst(s). Although SUD is explained using the third generation partnership project (3GPP) universal terrestrial radio access (UTRA) TDD system as the underlying communication system, it is applicable to other systems. That system is a direct sequence wideband CDMA (W-CDMA) system, where the uplink and downlink transmissions are confined to mutually exclusive time slots.

The receiver 28 receives using its antenna 40 a total of K bursts that arrive simultaneously, 48. The K bursts are superimposed on top of each other in one observation interval. For the 3GPP UTRA TDD system, each data field of a time slot corresponds to one observation interval.

For one observation interval, the data detection problem is viewed as per Equation 1.
r=H·d+n Equation 1
r is the received samples. H is the channel response matrix. d is the spread data vector. The spread data matrix contains the data transmitted in each channel mixed with that channel's spreading code.

When the received signal is oversampled, multiple samples of each transmitted chip are produced resulting in received vectors, r₁, . . . , r_N, (48). Similarly, the channel estimation device 44 determines the channel responses, H₁, . . . , H_N, corresponding to the received vectors, r₁, . . . , r_N, (50). For twice the chip rate, Equation 1 becomes Equation 2. $\begin{matrix} [\begin{matrix} r_{1} \\ r_{2} \end{matrix}] = [\begin{matrix} H_{1} \\ H_{2} \end{matrix}] \cdot d + n & Equation 2 \end{matrix}$
r₁is the even samples (at the chip rate) and r₂is the odd samples (offset half a chip from the r₁samples). H₁is the channel response for the even samples and H₂is the channel response for the odd samples.

Equation 1 becomes Equation 3 for a multiple N of the chip rate. $\begin{matrix} [\begin{matrix} r_{1} \\ r_{2} \\ ⋮ \\ r_{N} \end{matrix}] = [\begin{matrix} H_{1} \\ H_{2} \\ ⋮ \\ H_{N} \end{matrix}] \cdot d + n & Equation 3 \end{matrix}$
r₁, r₂. . . r_Nare the multiple of the chip rate samples. Each offset by 1/N of a chip. H₁, H₂. . . H_Nare the corresponding channel responses. Although the following discussion focuses on a receiver sampling at twice the chip rate, the same approaches are applicable to any multiple of the chip rate.

For twice the chip rate sampling, matrices H₁and H₂are (N_S+W−1) by N_Sin size. N_Sis the number of spread chips transmitted in the observation interval and W is the length of the channel impulse response, such as 57 chips in length. Since the received signal has N_Sspread chips, the length of r₁and r₂is N_S. Equation 2 is rewritten as Equation 4. $\begin{matrix} \begin{matrix} r_{1} (0) \\ r_{1} (1) \\ | \\ r_{1} (W - 1) \\ r_{1} (N_{S} - 1) \\ \dots \\ r_{2} (0) \\ r_{2} (1) \\ | \\ r_{2} (W - 1) \\ r_{2} (N_{S} - 1) \end{matrix}] = [\begin{matrix} h_{1} (0) & 0 & 0 & \dots & \dots & \dots & \dots & \dots \\ h_{1} (1) & h_{1} (0) & 0 & \dots \\ | & | \\ h_{1} (W - 1) & h_{1} (W - 2) & \dots & \dots & h_{1} (1) & h_{1} (0) & 0 & 0 \\ | & | & | & | & | & | \\ 0 & 0 & \dots & h_{1} (W - 1) & h_{1} (W - 2) & \dots & h_{1} (1) & h_{1} (0) \\ \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots \\ h_{2} (0) & 0 & 0 & \dots & \dots & \dots & \dots & \dots \\ h_{2} (1) & h_{2} (0) & 0 & \dots \\ | & | \\ h_{2} (W - 1) & h_{2} (W - 2) & \dots & \dots & h_{2} (1) & h_{2} (0) & 0 & 0 \\ | & | & | & | & | & | \\ 0 & 0 & \dots & h_{2} (W - 1) & h_{2} (W - 2) & \dots & h_{2} (1) & h_{2} (0) \\ \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots \end{matrix}] \cdot d + n & Equation 4 \end{matrix}$
r₁(i), r₂(i), h₁(i) and h₂(i) is the i^thelement of the corresponding vector matrix r₁, r₂, H₁and H₂, respectively.

One approach to determine the spread data vector is extended forward substitution, which is explained in conjunction with FIG. 4. For extended forward substitution, the received data vector is rearranged so that each even sample is followed by its corresponding odd sample. A similar rearrangement is performed on the channel response matrix, as shown in Equation 5a. $\begin{matrix} \begin{matrix} r_{1} (0) \\ r_{2} (0) \\ r_{1} (1) \\ r_{2} (1) \\ | \\ r_{1} (W - 1) \\ r_{2} (W - 1) \\ | \\ r_{1} (N_{S} - 1) \\ r_{2} (N_{S} - 1) \end{matrix}] = [\begin{matrix} h_{1} (0) & 0 & 0 & \dots \\ h_{2} (0) & 0 & 0 & \dots \\ h_{1} (1) & h_{1} (0) & 0 & \dots \\ h_{2} (1) & h_{2} (0) & 0 & \dots \\ | & | & | & \dots \\ h_{1} (W - 1) & h_{1} (W - 2) & \dots & \dots & h_{1} (1) & h_{1} (0) & 0 & 0 \\ h_{2} (W - 1) & h_{2} (W - 2) & \dots & \dots & h_{2} (1) & h_{2} (0) & 0 & 0 \\ | & | & | & | & | & | & | \\ 0 & 0 & \dots & h_{1} (W - 1) & h_{1} (W - 2) & \dots & h_{1} (1) & h_{1} (0) \\ 0 & 0 & \dots & h_{2} (W - 1) & h_{2} (W - 1) & \dots & h_{2} (1) & h_{2} (0) \end{matrix}] \cdot [\begin{matrix} d (0) \\ d (1) \\ | \\ | \\ d (N_{S} - 1) \end{matrix}] + n & Equation 5 a \end{matrix}$

Similarly, for an N-multiple of the chip rate sampling, Equation 5b is the arrangement. $\begin{matrix} [\begin{matrix} r_{1} (0) \\ r_{2} (0) \\ | \\ r_{N} (0) \\ r_{1} (1) \\ r_{2} (1) \\ | \\ r_{N} (1) \\ | \\ r_{1} (N_{S} - 1) \\ r_{2} (N_{S} - 1) \\ | \\ r_{N} (N_{S} - 1) \end{matrix}] = [\begin{matrix} h_{1} (0) & 0 & \dots & 0 & 0 \\ h_{2} (0) & 0 & \dots & 0 & 0 \\ | & | & \dots & | & | \\ h_{N} (0) & 0 & \dots & 0 & 0 \\ h_{1} (1) & h_{1} (0) & \dots & 0 & 0 \\ h_{2} (1) & h_{2} (0) & \dots & 0 & 0 \\ | & | & \dots & | & | \\ h_{N} (1) & h_{N} (0) & \dots & 0 & 0 \\ | & | & \dots & | & | \\ 0 & 0 & \dots & h_{1} (1) & h_{1} (0) \\ 0 & 0 & \dots & h_{2} (1) & h_{2} (0) \\ | & | & \dots & | & | \\ 0 & 0 & \dots & h_{N} (1) & h_{N} (0) \end{matrix}] . [\begin{matrix} d (0) \\ d (1) \\ | \\ | \\ d (N_{S} - 1) \end{matrix}] + n & Equation 5 b \end{matrix}$
d(i) is the i^thelement of the spread data vector, d. The length of the spread data vector is N_S. Using extended forward substitution, the zero-forcing solution to determine d(0),dˆ(0), is per Equations 6a and 7a, (52). $\begin{matrix} [\begin{matrix} h_{1} (0) \\ h_{2} (0) \end{matrix}] \cdot d (0) = [\begin{matrix} r_{1} (0) \\ r_{2} (0) \end{matrix}] & Equation 6 a \\ \hat{d} (0) = {[h_{1} (0) h_{2} (0)] [\begin{matrix} h_{1} (0) \\ h_{2} (0) \end{matrix}]}^{- 1} [h_{1} (0) h_{2} (0)] [\begin{matrix} r_{1} (0) \\ r_{2} (0) \end{matrix}] & Equation 7 a \end{matrix}$

Equation 6a is the general formula for d(0). Equation 7a is the zero forcing solution for dˆ(0). Similarly, for N-multiple of the chip rate, Equations 6b and 7b are used. $\begin{matrix} [\begin{matrix} h_{1} (0) \\ h_{2} (0) \\ | \\ h_{N} (0) \end{matrix}] d (0) = [\begin{matrix} r_{1} (0) \\ r_{2} (0) \\ | \\ r_{N} (0) \end{matrix}] & Equation 6 b \\ \hat{d} (0) = {[h_{1} (0) \dots h_{N} (0)] [\begin{matrix} h_{1} (0) \\ ⋮ \\ h_{N} (0) \end{matrix}]}^{- 1} [h_{1} (0) \dots h_{N} (0)] [\begin{matrix} r_{1} (0) \\ ⋮ \\ r_{N} (0) \end{matrix}] & Equation 7 b \end{matrix}$

In solving Equation 7a and 7b, for later use v^His determined as illustrated by Equation 8 for the v^Hfor Equation 7a and stored, (52). $\begin{matrix} v^{H} = {[h_{1} (0) h_{2} (0)] [\begin{matrix} h_{1} (0) \\ h_{2} (0) \end{matrix}]}^{- 1} [h_{1} (0) h_{2} (0)] & Equation 8 \end{matrix}$
dˆ(0) is determined using v^Hper Equation 9. $\begin{matrix} d \hat{} (0) = v^{H} [\begin{matrix} r_{1} (0) \\ r_{2} (0) \end{matrix}] & Equation 9 \end{matrix}$

Using the Toplitz structure of the H matrix, the remaining spread data elements can be determined sequentially using zero forcing per Equation 10a, (54). $\begin{matrix} d \hat{} (i) = v^{H} {[\begin{matrix} r_{1} (i) \\ | \\ r_{N} (i) \end{matrix}] - [\begin{matrix} h_{1} (i) \\ | \\ h_{N} (i) \end{matrix}] \hat{d} (0) - \sum_{j = 1}^{i - 1} [\begin{matrix} h_{1} (j) \\ | \\ h_{N} (j) \end{matrix}] \cdot d (i - j - 1) & Equation 10 b \end{matrix}$

For an N-multiple of the chip rate, Equation 10b is used. $\begin{matrix} d^{^} (i) = v^{H} {[\begin{matrix} \begin{matrix} r_{1} (i) \\ | \end{matrix} \\ r_{N} (i) \end{matrix}] - [\begin{matrix} \begin{matrix} h_{1} (i) \\ | \end{matrix} \\ h_{N} (i) \end{matrix}] \hat{d} (0) - \sum_{j = 1}^{i - 1} [\begin{matrix} \begin{matrix} h_{1} (j) \\ | \end{matrix} \\ h_{N} (j) \end{matrix}] \cdot d (i - j - 1) & Equation 10 b \end{matrix}$

After the spread data vector is determined, each communication burst's data is determined by despreading, such as by mixing the spread data vector with each burst's code, (56).

The complexity in using the extended forward substitution approach, excluding despreading, is summarized in Table 1.

TABLE 1 Calculating v^H 4 multiplications & 1 reciprocal Calculating d{circumflex over ( )}(0) 2 multiplications Calculating d{circumflex over ( )}(1) 4 multiplications Calculating each up to 2 multiplications d{circumflex over ( )}(W − 1) Calculating each d{circumflex over ( )}(i) from 2W + 2 multiplications d{circumflex over ( )}(w) to d{circumflex over ( )}(N_S− 1) Total Number of Multiplications 2N_S+ (W − 1) · W + 2W . . . (N_S− W + 1) Total Number of Calculations 2N_S+ (W − 1) · W + 2W . . . (N_S− W + 1) + 5

For a TDD burst type II, N_Sis 1104 and W is 57, solving for d using extended forward substitution 200 times per second requires 99.9016 million real operations per second (MROPS) for twice the chip rate sampling or 49.95 MROPs for chip rate sampling.

Another approach to estimate data is an approximate banded Cholesky approach, which is explained in conjunction with FIG. 5. A cross correlation matrix R is determined so that it is square, (N_Sby N_S), and banded per Equation 11, (58).
R=H^HH Equation 11
(·)^Hindicates the hermetian function. H is of size 2(N_S+W−1) by N_S. Equation 11 is rewritten as Equation 12a for twice the chip rate sampling. $\begin{matrix} R = [H_{1}^{H} H_{2}^{H}] \cdot [\begin{matrix} H 1 \\ H 2 \end{matrix}] = H_{1}^{H} H_{1} + H_{2}^{H} H_{2} & Equation 12 a \end{matrix}$

For an N-multiple of the chip rate, Equation 12b is used. $\begin{matrix} R = [H_{1}^{H}, H_{2}^{H} - - H_{N}^{H}] [\begin{matrix} \begin{matrix} \begin{matrix} H_{1} \\ H_{2} \end{matrix} \\ | \end{matrix} \\ H_{N} \end{matrix}] or R = \sum_{i = 1}^{N} H_{i}^{H} H_{i} & Equation 12 b \end{matrix}$

Using Equation 12a or 12b, the resulting R is of size N_Sby N_Sand banded as illustrated in Equation 13 for twice the chip rate sampling, W=3 and N_S=10. $\begin{matrix} R = [\begin{matrix} R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\ R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 & 0 \\ R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 \\ 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 \\ 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 \\ 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 \\ 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 \\ 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} \\ 0 & 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} \\ 0 & 0 & 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} \end{matrix}] & Equation 13 \end{matrix}$

In general, the bandwidth of R is per Equation 14.
p=W−1 Equation 14

Using an approximate Cholesky approach, a sub-block of R, R_sub, of size N_colby N_colis used. A typical size of R_subis 2W−1 by 2W−1, although other sized matrices may be used. The sub-block, R_sub, is decomposed using Cholesky decomposition per Equation 15, (60).
R_sub=G G^H Equation 15

The Cholesky factor G is of size N_colby N_col. An illustration of a 5x 5G matrix with W=3 is per Equation 16. $\begin{matrix} G = [\begin{matrix} G_{11} & 0 & 0 & 0 & 0 \\ G_{21} & G_{22} & 0 & 0 & 0 \\ G_{31} & G_{32} & G_{33} & 0 & 0 \\ 0 & G_{42} & G_{43} & G_{44} & 0 \\ 0 & 0 & G_{53} & G_{54} & G_{33} \end{matrix}] & Equation 16 \end{matrix}$
G_ijis the element of the G matrix at the i^thcolumn and j^throw. The G matrix is extended to an N_Sby N_Smatrix, G_full, by right shifting the bottom row of G by one element for each row after the last row of G, (62). For N_S=10, the illustration of Equation 16 is expanded per Equation 17, (62). $G_{full} = [\begin{matrix} G_{11} & 0 \\ G_{21} & G_{22} & 0 \\ G_{31} & G_{32} & G_{33} & 0 \\ 0 & G_{42} & G_{43} & G_{44} & 0 \\ 0 & G_{53} & G_{54} & G_{55} & 0 \\ 0 & G_{53} & G_{54} & G_{55} & 0 \\ 0 & G_{53} & G_{54} & G_{55} & 0 \\ 0 & G_{53} & G_{54} & G_{55} & 0 \\ 0 & G_{53} & G_{54} & G_{55} & 0 \\ 0 & G_{53} & G_{54} & G_{55} \end{matrix}]$

The spread data vector is determined using forward and backward substitution, (64). Forward substitution is used to determine y per Equation 18a for twice the chip rate sampling and Equation 18b for a multiple N of the chip rate sampling.
G_fully=H₁^Hr₁+H₂^Hr₂ Equation 18a
G_fully=H₁^Hr+H₂^Hr₂+ . . . +H_N^Hr_n Equation 18b
Backward substitution is subsequently used to solve for the spread data vector per Equation 19.
G_full^Hd=y Equation 19

After the spread data vector, d, is determined, each burst's data is determined by despreading, (66).

The complexity of approximate Cholesky decomposition, excluding despreading, for twice the chip rate sampling is per Table 2.

TABLE 2 Operation Number of Calculations Calculating H^HH W(W + 1) Calculating Cholesky Decomposition

\frac{{N_{col} (W - 1)}^{2}}{2} + \frac{3 N_{col} (W - 1)}{2} - \frac{{(W - 1)}^{3}}{3} - {(W - 1)}^{2} - \frac{2 (W - 1)}{3}

Calculating H^Hr 2N_SW Forward Substitution

[N_{S} - \frac{(W - 1)}{2}] W and the reciprocal of N_{S}

real numbers Backward Substitution

[N_{S} - \frac{(W - 1)}{2}] W and the reciprocal of N_{S}

real numbers

For a TDD burst type II, N_Sis 1104 and for W is 57, performing approximate banded Cholesky 200 times per second at twice the chip rate requires 272.56 MROPS. By contrast, an exact banded Cholesky approach requires 906.92 MROPS. For chip rate sampling, the approximate banded Cholesky approach requires 221.5 MROPS.

Another approach for data detection uses a Toeplitz approach, (Levinson-Durbin type algorithm), which is explained in conjunction with FIG. 6. The R matrix of Equation 12a and 12b is reproduced here. $\begin{matrix} R = [H_{1}^{H} H_{2}^{H}] \cdot [\begin{matrix} H 1 \\ H 2 \end{matrix}] = H_{1}^{H} H_{1} + H_{2}^{H} H_{2} & Equation 12 a \end{matrix}$
For an N-multiple of the chip rate, Equation 12b is used. $\begin{matrix} R = [H_{1}^{H}, H_{2}^{H} - - H_{N}^{H}] [\begin{matrix} H_{1} \\ H_{2} \\ | \\ H_{N} \end{matrix}] or R = \sum_{i = 1}^{N} H_{i}^{H} H_{i} & Equation 12 b \end{matrix}$

The R matrix is symmetric and Toeplitz with a bandwidth of p=W−1, (68). A left most upper corner of the R matrix, R(k), being a k by k matrix is determined as shown in Equation 20. $\begin{matrix} R (k) = [\begin{matrix} R_{0} & R_{1} & \dots & R_{k - 1} \\ R_{1} & R_{0} & ⋮ \\ ⋰ \\ R_{k - 1} & R_{k - 2} & \dots & R_{0} \end{matrix}] & Equation 20 \end{matrix}$

Additionally, another vector R_kis determined using elements of R, per Equation 21, (72). $\begin{matrix} R_{k} = [\begin{matrix} R_{1} \\ R_{2} \\ ⋮ \\ R_{k} \end{matrix}] . & Equation 21 \end{matrix}$

Bolding indicates a matrix including all elements up to its subscript. At stage k+1, the system is solved per Equation 22.
R(k+1)d(k+1)=[H^Hr]_k+1 Equation 22
[H^Hr]_k+1is the first (k+1) components of H^Hr. d(k+1) is broken into a vector d₁(k+1) of length k and a scalar d₂(k+1) as per Equation 23. $\begin{matrix} d (k + 1) = [\begin{matrix} d_{1} (k + 1) \\ d_{2} (k + 1) \end{matrix}] . & Equation 23 \end{matrix}$

The matrix R(k+1) as decomposed as per Equation 24. $\begin{matrix} R (k + 1) = [\begin{matrix} R (k) & ⋮ & E_{k} R_{k} \\ \dots & ⋮ & \dots \\ R_{k}^{H} E_{k} & ⋮ & R_{0} \end{matrix}] . & Equation 24 \end{matrix}$

E_kis an exchange matrix. The exchange matrix operates on a vector by reversing that vectors elements.

Using the Yule-Walker equation for linear prediction, Equation 25 results, (78). $\begin{matrix} [\begin{matrix} R (k - 1) & ⋮ & E_{k - 1} R_{k - 1} \\ \dots & ⋮ & \dots \\ R_{k - 1}^{H} E_{k - 1} & ⋮ & R_{0} \end{matrix}] [\begin{matrix} y_{1} (k) \\ y_{2} (k) \end{matrix}] = - [\begin{matrix} R_{k - 1} \\ R_{k} \end{matrix}] & Equation 25 \end{matrix}$

Using order recursions, Equations 26, 27 and 28 result. $\begin{matrix} y_{1} (k) = y (k - 1) + y_{2} (k) E_{k - 1} y (k - 1) & Equation 26 \\ y_{2} (k) = - \frac{[R_{k} + R_{k - 1}^{H} E_{k - 1} y (k - 1)]}{[1 + R_{k - 1}^{H} y (k - 1)]} & Equation 27 \\ y (k) = [\begin{matrix} y_{1} (k) \\ y_{2} (k) \end{matrix}] & Equation 28 \end{matrix}$

Using y(k), d(k+1) is determined per Equations 29, 30 and 31, (74). $\begin{matrix} d_{1} (k + 1) = d (k) + d_{2} (k + 1) E_{k} y (k) & Equation 29 \\ d_{2} (k + 1) = [\frac{{(H^{H} r)}_{k + 1} - R_{k}^{H} E_{k} d (k)}{1 + R_{k}^{H} y (k)}] & Equation 30 \\ d (k + 1) = [\begin{matrix} d_{1} (k + 1) \\ d_{2} (k + 1) \end{matrix}] & Equation 31 \end{matrix}$
(H^Hr)_k+1is the (k+1)^thelement of H^Hr.

After properly initializing the recursions, the recursions are computed for k=1, 2, . . . , N_S·d(N_S) is a solution to Equation 32, (74).
Rd=H^Hr Equation 32

The spread data vector d is despread with the burst's channelization codes to recover the data, (76).

The banded structure of R affects the recursions as follows. R(2) and R₂are per Equation 33. $\begin{matrix} R (2) = [\begin{matrix} R_{0} & R_{1} \\ R_{1} & R_{0} \end{matrix}], R_{2} = [\begin{matrix} R_{1} \\ R_{2} \end{matrix}] & Equation 33 \end{matrix}$

The inner product calculations in Equations 27 and 30 require two multiplications each. To illustrate, the R matrix of Equation 20 for k=6 is per Equation 34. $\begin{matrix} R (6) = [\begin{matrix} R_{0} & R_{1} & R_{2} & 0 & 0 & 0 \\ R_{1} & R_{2} & 0 & 0 \\ R_{2} & R_{2} & 0 \\ 0 & R_{2} & R_{2} \\ 0 & 0 & R_{2} & R_{1} \\ 0 & 0 & 0 & R_{2} & R_{0} \end{matrix}], R_{6} = [\begin{matrix} R_{1} \\ R_{2} \\ 0 \\ 0 \\ 0 \\ 0 \end{matrix}] & Equation 34 \end{matrix}$

The number of non-zero elements in the vector R₆is equal to the bandwidth, p, of the matrix R. When the inner product of R₆^HE₆y(k) in Equation 27 and the inner product R₆^HE₆d(k) in Equation 30 are calculated, only p (and not k) multiplications are required. For the recursions of Equations 26 and 29, no computational reductions are obtained.

Table 3 shows the complexity in implementing the Toeplitz approach. The total MROPs for the Toeplitz approach for a TDD burst type is 909.4656 MROPs e the chip rate sampling and 858.4668 MROPs for chip rate sampling.

Another approach for data detection uses fast fourier transforms (FFTs), explained using FIG. 7. If chip rate sampling is used, the channel matrix H is square except for edge effects. Using a circulant approximation for the H matrix, a FFT of the received vector, r, and the channel vector, H, to obtain the data estimate is

For a multiple of the chip rate sampling, such as twice the chip rate, the H is not square or circulant. However, a submatrix, shown by dotted lines, of the channel correlation matrix R =H^HH matrix of Equation 13, (84), as shown for Equation 35a is circulant. $\begin{matrix} R = [\begin{matrix} R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 & 0 & 0 \\ R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 & 0 \\ \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots \\ R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 & 0 \\ 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 & 0 \\ 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 & 0 \\ 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 & 0 \\ 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} & 0 \\ 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} & R_{2} \\ \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots & \dots \\ 0 & 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} & R_{1} \\ 0 & 0 & 0 & 0 & 0 & 0 & 0 & R_{2} & R_{1} & R_{0} \end{matrix}] & Equation 35 a \end{matrix}$

For an N-multiple of the chip rate sampling, the channel correlation matrix is determined per Equation 35b. $\begin{matrix} R = [H_{1}^{H} H_{2}^{H} -- H_{N}^{H}] [\begin{matrix} H_{1} \\ H_{2} \\ ❘ \\ H_{N} \end{matrix}] = \sum_{i - 1}^{N} H_{i}^{H} H_{i} & Equation 35 b \end{matrix}$

By approximating the R matrix as being circulant, Equations 36, 37 and 38 are used. $\begin{matrix} R^{H} = D Δ D^{H} Δ = diag ({D (R)}_{1}) & Equation 36 \\ d^{⋀} = diag (D [\begin{matrix} R_{0} \\ R_{1} \\ R_{2} \\ 0 \\ ⋮ \\ 0 \end{matrix}]) & Equation 37 \end{matrix}$
(R)₁is the first column of the R matrix expanded to a diagonal matrix. Although described using the first column, this approach can be modified to use any column of the R matrix, (86). However, it is preferred to use a column having the most non-zero elements of any column, such as R₂, R₁, R₀, R₁, R₂. These columns are typically any column at least W columns from both sides, such as any column between and including W and N_S−W−1. Equations 38 and 39 are used in a zero-forcing equalization approach.
Rdˆ=H^Hr Equation 38
dˆ=R⁻¹(H^Hr) Equation 39

Since D is an orthogonal discrete fourier transform (DFT) matrix, Equations 40, 41 and 42 result. $\begin{matrix} D^{H} D = N_{S} I & Equation 40 \\ D^{- 1} = (\frac{1}{N_{S}}) D^{H} & Equation 41 \\ R^{- 1} = \frac{1}{N_{S}} D^{H} Δ^{- 1} \frac{1}{N_{S}} D & Equation 42 \end{matrix}$

Accordingly, dˆ can be determined using a fourier transform per Equations 43, 44 and 45a. $\begin{matrix} R^{- 1} = \frac{1}{N_{S}^{2}} D^{H} Δ^{- 1} [D (H^{H} r)] & Equation 43 \\ D^{H} d^{⋀} = \frac{1}{N_{S}} Δ^{- 1} [F (H^{H} r)] & Equation 44 \\ F (d^{⋀}) = \frac{F (H^{H} r)}{N_{S} F ({(R)}_{1})} & Equation 45 a \end{matrix}$
(·)₁is the first column, although an analogous equation can be used for any column of R. F(·) denotes a fourier transform function. F(H^Hr) is preferably calculated using FFTs per Equation 45b.
F(H^Hr)=N_c[F(h₁)F(r₁)+ . . . +F(h_N)F(r_N)] Equation 45b
Taking the inverse fourier transform F⁻¹(·), of the result of Equation 45a produces the spread data vector, (88). The transmitted data can be recovered by despreading using the appropriate codes, (90).

The complexity of this FFT approach is shown in Table 4.

TABLE 4 Functions Executed Once Per Burst Calculation # of Calculations MROPs Calculating H^HH 1.3224 F([R]₁) · N_Slog₂N_S

11160 \times \frac{100}{10^{6}}

4.4640 Functions Executed 38 Twice Per Burst Calculating H^Hr by FFT Calculating Equation 45 0.8832 F⁻¹(d) · N_Slog₂N_S 8.9280 Total 55 MROPS

The FFT approach is less complex than the other approaches. However, a performance degradation results from the circulant approximation.

Another approach to use FFTs to solve for the data vector for a multiple of the chip rate sampling combines the samples by weighing, as explained using FIG. 8. To illustrate for twice the chip rate sampling, r₁is the even and r₂is the odd samples. Each element of r₁, such as a first element r₁(0), is weighted and combined with a corresponding element of r₂, such as r₂(0) per Equation 46.
r_eff(0)=W₁r₁(0)+W₂r₂(0) Equation 46
r_eff(0) is the effectively combined element of an effectively combined matrix, r_eff. W₁and W₂are weights. For N-times the chip rate sampling, Equation 47 is used.
r_eff(0)=W₁r₁(0)+ . . . +W_nr_n(0) Equation 47
Analogous weighting of the channel response matrices, H₁to H_n, is performed to produce H_eff, (92). As a result, Equation 3 becomes Equation 48.
r_eff=H_effd+n Equation 48
The resulting system is an N_Sby N_Ssystem of equations readily solvable by FFTs per Equation 49, (94). $\begin{matrix} F (d) = \frac{F (r_{reff})}{F ({(H_{eff})}_{1})} & Equation 49 \end{matrix}$

Using the inverse fourier transform, the spread data vector is determined. The bursts'data is determined by despreading the spread data vector using the bursts'code, (96). Although Equation 49 uses the first column of H_eff, the approach can be modified to use any representative column of, H_eff.

Another approach using FFTs uses zero padding, which is explained using FIG. 9. Equation 5 is modified by zero padding the data vector so that every other element, such as the even elements are zero, (98). The modified d matrix is d¹⁸. The H matrix is also expanded to form H¹⁸. H matrix is expanded by repeating each column to the right of that column and shifting each element down one row and zero padding the top of the shifted column. An illustration of such a system for twice the chip rate sampling, W=3 and N_S=4 is per Equation 49a. $\begin{matrix} [\begin{matrix} r_{1} (0) \\ r_{2} (0) \\ r_{1} (1) \\ r_{2} (1) \\ r_{1} (2) \\ r_{2} (2) \\ r_{1} (3) \\ r_{2} (3) \end{matrix}] = [\begin{matrix} h_{1} (0) \\ h_{2} (0) & h_{1} (0) \\ h_{1} (1) & h_{2} (0) & h_{1} (0) \\ h_{2} (1) & h_{1} (1) & h_{2} (0) & h_{1} (0) \\ h_{1} (2) & h_{2} (1) & h_{1} (1) & h_{2} (0) & h_{1} (0) & h_{2} (0) \\ h_{2} (2) & h_{1} (2) & h_{2} (1) & h_{1} (1) & h_{2} (0) & h_{1} (0) & h_{2} (0) \\ 0 & h_{2} (2) & h_{1} (2) & h_{2} (1) & h_{1} (1) & h_{2} (0) & h_{1} (0) & h_{2} (0) \\ 0 & 0 & h_{2} (2) & h_{1} (2) & h_{2} (1) & h_{1} (1) & h_{2} (1) & h_{1} (1) \end{matrix}] \cdot [\begin{matrix} d (0) \\ 0 \\ d (1) \\ 0 \\ d (2) \\ 0 \\ d (3) \\ 0 \end{matrix}] + n & Equation 49 a \end{matrix}$

For an N-multiple of the chip rate, Equation 49b is used as shown for simplicity for N_S=3. $\begin{matrix} [\begin{matrix} r_{1} (0) \\ r_{2} (0) \\ ❘ \\ r_{N} (0) \\ r_{1} (1) \\ r_{2} (1) \\ ❘ \\ r_{N} (1) \\ r_{1} (2) \\ r_{2} (2) \\ ❘ \\ r_{N - 1} (2) \\ r_{N} (2) \end{matrix}] = [\begin{matrix} h_{1} (0) & 0 & -- & 0 \\ h_{2} (0) & h_{1} (0) & -- & 0 \\ ❘ & ❘ & -- & ❘ \\ h_{N} (0) & h_{N - 1} (0) & -- & 0 \\ h_{1} (1) & h_{N} (0) & -- & 0 \\ h_{2} (1) & h_{1} (1) & -- & 0 \\ ❘ & ❘ & -- & ❘ \\ h_{N} (1) & h_{N - 1} (1) & -- & 0 \\ 0 & h_{N} (1) & -- & 0 \\ 0 & 0 & -- & 0 \\ ❘ & ❘ & -- & ❘ \\ 0 & 0 & -- & h_{2} (0) \\ 0 & 0 & -- & h_{1} (0) \end{matrix}] \cdot [\begin{matrix} d (0) \\ 0 \\ | \\ 0 \\ 0 \\ d (1) \\ 0 \\ 0 \\ | \\ 0 \\ d (2) \\ 0 \\ | \\ 0 \\ 0 \end{matrix}] + n & Equation 49 b \end{matrix}$

In general, the H^˜ matrix for an N multiple is (N N_S) by (N N_S). The matrix H^˜is square, Toeplitz and approximately circulant and is of size 2N_Sby 2N_S. The zero forcing solution is per Equation 50, (100). $\begin{matrix} F (d^{~}) = \frac{F (r)}{F ({(H^{~})}_{1})} & Equation 50 \end{matrix}$

A column other than the first column may be used in an analogous FFT. Furthermore, since any column may be used, a column from the original channel response matrix, H or an estimated expanded column of the H^˜ derived from a column of H. Using every N^thvalue from d^˜, d is estimated. Using the appropriate codes, d is despread to recover the data, (102).

Claims

1. A method of receiving a plurality of data signals that are concurrently transmitted at a predetermined chip rate from a transmitter site over a shared spectrum in a code division multiple access communication system, the method comprising:

receiving a combined signal of the plurality of concurrently transmitted data signals over the shared spectrum;

sampling the combined signal at a multiple of the chip rate;

estimating a channel response for the combined signal at the multiple of the chip rate;

determining a spread data vector using the combined signal samples and the estimated channel response; and

estimating data of the data signals using the spread data vector.

2. The method of claim 1 wherein the determining of a spread data vector includes:

determining a first element of a spread data vector using the combined signal samples and the estimated channel response; and

using a factor from the first element determination to determine remaining elements of the spread data vector.

3. The method of claim 2 wherein the factor is derived using the channel response and the factor is stored prior to the remaining elements determination.

4. The method of claim 3 wherein the multiple of the chip rate is N-multiple of the chip rate and the factor is vH and a first element of each channel response matrix corresponding to each N-multiple of the chip rate is h1(0), h2(0)... hN (0) and v H = { [ h 1 ⁡ ( 0 ) ⁢ … ⁢ ⁢ h N ⁡ ( 0 ) ] ⁡ [ h 1 ⁡ ( 0 ) ⋮ h N ⁡ ( 0 ) ] } - 1 ⁡ [ h 1 ⁡ ( 0 ) ⁢ … ⁢ ⁢ h N ⁡ ( 0 ) ].

5. The method of claim 4 wherein an ith element of the spread data vector, dˆ(i), other than the first element is determined per d ⋀ ⁡ ( i ) = v H ⁢ { [ r 1 ⁡ ( i ) ❘ r N ⁡ ( i ) ] - [ h 1 ⁡ ( i ) ❘ h N ⁡ ( i ) ] ⁢ d ^ ⁡ ( 0 ) - ∑ j = 1 i - 1 ⁢ [ h 1 ⁡ ( j ) ❘ h N ⁡ ( j ) ] · d ⁡ ( i - j - 1 ).

6. The method of claim 1 wherein the data estimating is by despreading the spread data vector.

7. The method of claim 1 wherein the determining of a spread data vector includes:

determining a cross correlation matrix using the estimated channel response;

selecting a subblock of the cross correlation matrix;

determining a Cholesky factor for the subblock;

extending the Cholesky factor; and

determining the spread data vector using the extended Cholesky factor, a version of the channel response and the samples.

8. The method of claim 7 wherein the channel response is estimated as a channel response matrix and the cross correlation matrix is the hermetian of the channel response matrix multiplied by the channel response matrix.

9. The method of claim 8 wherein the multiple is twice the chip rate sampling, the channel response matrix has even matrix samples H1 and odd matrix samples H2.

10. The method of claim 8 wherein the multiple is N-times the chip rate sampling, the channel response matrix has N sets of matrix samples, H1, H2,... HN.

11. The method of claim 7 wherein the subblock has 2W−1 by 2W−1 elements of the cross correlation matrix and W is a length of the impulse response.

12. The method of claim 7 wherein the determining the spread data vector uses forward and backward substitution.

13. The method of claim 1 wherein the determining of a spread data vector includes:

determining a cross correlation matrix using the estimated channel response; and

determining the spread data vector using order recursions by determining a first spread data estimate using an element from the cross correlation matrix and recursively determining further estimates using additional elements of the cross correlation matrix.

14. The method of claim 13 wherein the spread data estimates are determined by combining a scalar and a vector portions of the spread data estimates.

15. The method of claim 13 wherein the spread data vector determining is performed using Yule-Walker equations.

16. The method of claim 13 wherein the first spread data estimate is determined using an element of an upper left corner of the cross correlation matrix.

17. The method of claim 1 wherein the determining of a spread data vector includes:

determining a column of a channel correlation matrix using the estimated channel response; and

determining the spread data vector using the determined column, the estimated channel response, the received combined signal and a fourier transform.

18. The method of claim 17 wherein the determined column is a first column of the channel correlation matrix.

19. The method of claim 17 wherein a length of an impulse response of the combined signal is W and the determined column is at least W−1 columns from edges of the channel correlation matrix.

20. The method of claim 17 wherein the determining the spread data vector uses a fourier transform of a hermetian of a channel response matrix multiplied by the received combined signal.

21. The method of claim 17 wherein the determining the spread data vector uses a fourier transform of the determined column and the determined column fourier transform is multiplied by a number of spread chips transmitted in the data signals.

22. The method of claim 1 wherein the determining of a spread data vector includes:

combining the multiple chip rate samples as effective chip rate samples;

combining the multiple chip rate estimated channel response as an effective chip rate channel response; and

determining a spread data vector using the effective samples, the effective channel response and a fourier transform.

23. The method of claim 22 wherein the multiple chip rate samples and the multiple chip rate estimated channel response are weighted prior to combining.

24. The method of claim 22 wherein the effective chip rate channel response is an effective channel response matrix.

25. The method of claim 22 wherein the determining of the spread data vector uses a column of a channel response matrix derived using the effective chip rate response.

26. The method of claim 22 wherein the column is a first column of the channel response matrix.

27. The method of claim 22 wherein the spread data vector determining further uses an inverse fourier transform.

28. The method of claim 1 wherein the estimating of the channel response is as a channel response matrix for the combined signal at the multiple of the chip rate and the determining of a spread data vector includes:

determining a padded version of a spread data vector of a size corresponding to the multiple chip rate using a column of the channel response matrix, the estimated channel response matrix, the samples and a fourier transform; and

estimating the spread data vector by eliminating elements of the padded version so that the estimated spread data vector is of a size corresponding to the chip rate.

29. The method of claim 28 wherein the multiple of the chip rate is an N-multiple of the chip rate and the estimated spread data vector comprises elements of the padded version spaced by N elements.

30. The method of claim 29 further comprising determining an expanded version of the channel response matrix by adding N−1 columns for every column of the channel response matrix.

31. A wireless communication station configured for use in receiving a plurality of data signals concurrently transmitted at a predetermined chip rate from a transmitter site over a shared spectrum in a code division multiple access communication system, the communication station comprising:

a receiver configured to receive a combined signal of the plurality of concurrently transmitted data signals over the shared spectrum;

a sampling device configured to sample the combined signal at a multiple of the chip rate;

channel response estimation circuitry configured to estimate a channel response for the combined signal at the multiple of the chip rate;

spread data vector determination circuitry configured to determine a spread data vector using the combined signal samples and the estimated channel response; and

data signal estimation circuitry configured to estimate data of the data signals using the spread data vector.

32. The communication station of claim 31 wherein the spread data vector determination circuitry is configured to determine a spread data vector by determining a first element of a spread data vector using the combined signal samples and the estimated channel response and by using a factor from the first element determination to determine remaining elements of the spread data vector.

33. The communication station of claim 32 wherein the spread data vector determination circuitry is configured to derive the factor using the channel response and to store the factor prior to the remaining elements determination.

34. The communication station of claim 33 wherein the multiple of the chip rate is N-multiple of the chip rate and the spread data vector determination circuitry is configured to determine the factor as vH and a first element of each channel response matrix corresponding to each N-multiple of the chip rate as h1(0), h2(0)... hN(0) and v H = { [ h 1 ⁡ ( 0 ) ⁢ … ⁢ ⁢ h N ⁡ ( 0 ) ] ⁡ [ h 1 ⁡ ( 0 ) ⋮ h N ⁡ ( 0 ) ] } - 1 ⁡ [ h 1 ⁡ ( 0 ) ⁢ … ⁢ ⁢ h N ⁡ ( 0 ) ].

35. The communication station of claim 34 wherein the spread data vector determination circuitry is configured to determine an ith element of the spread data vector, dˆ(i), other than the first element per d ⋀ ⁡ ( i ) = v H ⁢ { [ r 1 ⁡ ( i ) ❘ r N ⁡ ( i ) ] - [ h 1 ⁡ ( i ) ❘ h N ⁡ ( i ) ] ⁢ d ^ ⁡ ( 0 ) - ∑ j = 1 i - 1 ⁢ [ h 1 ⁡ ( j ) ❘ h N ⁡ ( j ) ] · d ⁡ ( i - j - 1 ).

36. The communication station of claim 31 wherein the data signal estimation circuitry is configured to estimate data by despreading the spread data vector.

37. The communication station of claim 31 wherein the spread data vector determination circuitry is configured to determine a cross correlation matrix using the estimated channel response, to select a subblock of the cross correlation matrix, to determine a Cholesky factor for the subblock, to extend the Cholesky factor and to determine the spread data vector using the extended Cholesky factor, a version of the channel response and the samples.

38. The communication station of claim 37 wherein the channel response estimation circuitry is configured to estimate the channel response as a channel response matrix and the spread data vector determination circuitry is configured to determine the cross correlation matrix as the hermetian of the channel response matrix multiplied by the channel response matrix.

39. The communication station of claim 38 where the multiple is twice the chip rate sampling, wherein the channel response estimation circuitry is configured to estimate the channel response as a channel response matrix having even matrix samples H1 and odd matrix samples H2.

40. The communication station of claim 38 where the multiple is N-times the chip rate sampling, wherein the channel response estimation circuitry is configured to estimate the channel response as a channel response matrix having N sets of matrix samples, H1, H2,... HN.

41. The communication station of claim 37 wherein the spread data vector determination circuitry is configured to determine the subblock with 2W−1 by 2W−1 elements of the cross correlation matrix and W as a length of the impulse response.

42. The communication station of claim 37 wherein the spread data vector determination circuitry is configured to determine the spread data vector using forward and backward substitution.

43. The communication station of claim 31 wherein the spread data vector determination circuitry is configured to determine a cross correlation matrix using the estimated channel response and to determine the spread data vector using order recursions by determining a first spread data estimate using an element from the cross correlation matrix and recursively determining further estimates using additional elements of the cross correlation matrix.

44. The communication station of claim 43 wherein the spread data vector determination circuitry is configured to determine the spread data estimates by combining a scalar and a vector portions of the spread data estimates.

45. The communication station of claim 43 wherein the spread data vector determination circuitry is configured to determine the spread data vector using Yule-Walker equations.

46. The communication station of claim 43 wherein the spread data vector determination circuitry is configured to determine the first spread data estimate using an element of an upper left corner of the cross correlation matrix.

47. The communication station of claim 31 wherein the spread data vector determination circuitry is configured to determine a column of a channel correlation matrix using the estimated channel response and to determine the spread data vector using the determined column, the estimated channel response, the received combined signal and a fourier transform.

48. The communication station of claim 47 wherein the spread data vector determination circuitry is configured to determine the determined column as a first column of the channel correlation matrix.

49. The communication station of claim 47 wherein the spread data vector determination circuitry is configured to determine a length of an impulse response of the combined signal as W and the determined column as at least W−1 columns from edges of the channel correlation matrix.

50. The communication station of claim 47 wherein the spread data vector determination circuitry is configured to determine the spread data vector using a fourier transform of a hermetian of a channel response matrix multiplied by the received combined signal.

51. The communication station of claim 47 wherein the spread data vector determination circuitry is configured to determine the spread data vector using a fourier transform of the determined column where the determined column fourier transform is multiplied by a number of spread chips transmitted in the data signals.

52. The communication station of claim 31 wherein the spread data vector determination circuitry is configured to combine the multiple chip rate samples as effective chip rate samples, to combine the multiple chip rate estimated channel response as an effective chip rate channel response, and to determine a spread data vector using the effective samples, the effective channel response and a fourier transform.

53. The communication station of claim 52 wherein the spread data vector determination circuitry is configured to weight the multiple chip rate samples and the multiple chip rate estimated channel response prior to combining.

54. The communication station of claim 52 wherein the spread data vector determination circuitry is configured to determine the effective chip rate channel response as an effective channel response matrix.

55. The communication station of claim 52 wherein the spread data vector determination circuitry is configured to determine the spread data vector using a column of a channel response matrix derived using the effective chip rate response.

56. The communication station of claim 52 wherein the spread data vector determination circuitry is configured to determine the column as a first column of the channel response matrix.

57. The communication station of claim 52 wherein the spread data vector determination circuitry is configured to determine the spread data vector using an inverse fourier transform.

58. The communication station of claim 31 wherein:

the channel response estimation circuitry is configured to estimate the channel response as a channel response matrix for the combined signal at the multiple of the chip rate and

the spread data vector determination circuitry is configured to determine a padded version of a spread data vector of a size corresponding to the multiple chip rate using a column of the channel response matrix, the estimated channel response matrix, the samples and a fourier transform, and to estimate the spread data vector by eliminating elements of the padded version so that the estimated spread data vector is of a size corresponding to the chip rate.

59. The communication station of claim 58 where the multiple of the chip rate is an N-multiple of the chip rate, wherein the spread data vector determination circuitry is configured to determine the estimated spread data vector as elements of the padded version spaced by N elements.

60. The communication station of claim 59 wherein the spread data vector determination circuitry is configured to determine an expanded version of the channel response matrix by adding N−1 columns for every column of the channel response matrix.

61. The communication station of claim 31 configured as a base station for a third generation partnership project (3GPP) universal terrestrial radio access (UTRA) system.

62. The communication station of claim 31 configured as a user equipment for a third generation partnership project (3GPP) universal terrestrial radio access (UTRA) system.