METHOD AND APPARATUS TO PROVIDE POWER CONVERSION WITH HIGH POWER FACTOR

Abstract

A power converter circuit rectifies a line voltage and applies the rectified voltage to a stack of capacitors. Voltages on the capacitors are coupled to a plurality of regulating converters to be converted to regulated output signals. The regulated output signals are combined and converted to a desired DC output voltage of the power converter. Input currents of the regulating converters are modulated in a manner that enhances the power factor of the power converter.

Description

FIELD

The subject matter described herein relates generally to electrical circuits and systems and more particularly to power conversion circuits and systems.

BACKGROUND

The power factor (PF) at the input port of a circuit is a parameter that is related to a ratio between the amount of real power drawn by the circuit (e.g., the average power entering the input port of the circuit) and the amount of apparent power drawn by the circuit (e.g., the product of the root-mean-square, or RMS, voltage at the input port and the RMS current at the input port). In an ac-dc converter, high power factor is generally desired to best convey real power from an alternating current (ac) input to a direct current (dc) output. For example, in a power converter used as a grid interface, a high PF is desired to maintain high power quality. A common technique to achieve high power factor is to cascade a “Power Factor Correction” (PFC) rectifier circuit with a de-dc converter. The PFC circuit shapes a sinusoidal input current and buffers twice-line-frequency ac power on a capacitor with a certain average do voltage. The subsequent dc-dc converter then regulates the load voltage from this buffered voltage.

A boost converter is often selected for use in a PFC circuit because of its high power factor capability (about 0.99). However, a boost converter will typically require the use of high-voltage rated switch, diode, and capacitor components (e.g., 2-3 times peak voltage of about 200-400V). Even worse, in the case of a low do voltage load application, the subsequent dc-dc converter would need a high step-down conversion ratio from the high stepped-up buffered voltage to the low do voltage. Therefore, high voltage stress will exist on the components of the dc-dc converter, deteriorating the efficiency.

Instead of a boost converter, one alternative for a PFC circuit is a buck converter, which draws a clipped current waveform (i.e., draws current when the input ac voltage magnitude exceeds the do buffer voltage). Often a clipped-sine current waveform is drawn, yielding a ˜0.9-0.95 power factor for typical ac inputs. A benefit of a PFC having a buck topology is the reduced voltage stress and conversion ratio for the corresponding dc-dc converter. However, the active components in the buck PFC circuit still need to be operated directly from the ac line voltage, and thus the buck PFC needs to be designed with high voltage (e.g., 300-600V) active devices. This high voltage requirement also typically results in low achievable switching frequencies and the need for large passive components. Moreover, in both typical boost and buck PFC circuits, the twice-line frequency energy is stored on an output capacitor with small twice-line-frequency voltage ripple, requiring a peak capacitive energy storage rating that is large compared to the amount of twice-line-frequency energy that needs to be buffered. Consequently, PFC circuits with buck converters often display low efficiency and low power density.

New power conversion architectures and methods are needed that are capable of achieving high power factor with one or more of high switching frequency, low component voltage stress, low buffer energy storage requirements, high power density, and/or high efficiency.

SUMMARY

A new power conversion architecture is provided that is capable of achieving high power factor. The power conversion architecture is suitable for use in, for example, grid interface power converters and other power converters where high power factor is an important operating characteristic. The power conversion architecture is capable of providing high power factor while also achieving one or more of high switching frequency, low component voltage stress, low peak buffer energy storage requirement, high power density, and/or high efficiency. The architecture can significantly decrease the voltage stress of the active and passive devices within the converter, thus enabling high-frequency operation and small size. Moreover good power factor can be achieved while dynamically buffering twice-line-frequency energy with relatively small capacitors. In some implementations, an ac-dc power converter architecture is provided that is suitable for use as a single-phase grid interface at high switching frequency (e.g., above 3 MHz). Features, concepts, and circuits described herein may also be used to provide bi-directional power converter circuits and dc-ac circuits.

A prototype power converter was developed for use in light emitting diode (LED) driver operations. Testing has shown that the prototype implementation achieves an efficiency of 93.3% and a power factor of 0.89, while maintaining high frequency operation and small size (with a displacement power density of 130 W/in³). As will be described in greater detail, the circuits, systems, concepts, and features described herein can be used to provide power converters with different regulating and combining converter topologies, and can likewise be adapted to the requirements of a variety of different applications, voltage ranges and power levels.

In accordance with one aspect of the concepts, systems, circuits, and techniques described herein, a power converter circuit comprises: a line frequency rectifier circuit configured to rectify an alternating current (ac) input voltage of the power converter circuit; a stack of capacitors to receive an output signal of the line frequency rectifier circuit, the stack of capacitors having at least two stacked capacitors; a set of regulating converters coupled to the stack of capacitors, each regulating converter in the set of regulating converters being configured to draw current from a corresponding one of the stacked capacitors and to generate a regulated voltage at an output thereof in response thereto; a power combining converter circuit configured to combine the output signals of the set of regulating converters and to convert the combined signal to a desired direct current (dc) output voltage; and a controller configured to controllably modulate the input current of one or more of the regulating converters in a manner that enhances power factor in the power converter circuit.

In some embodiments, the controller is configured to controllably modulate the input currents of the regulating converters in a manner that regulates the voltage or current at one or outputs of the regulating converters and/or at one or more system outputs.

In some embodiments, the controller is configured to controllably modulate the input currents of the regulating converters in a manner that results in a power factor in the power converter circuit that is greater than or equal to 0.8.

In some embodiments, the controller is configured to controllably modulate the input current of all of the regulating converters in the set in a manner that enhances power factor in the power converter circuit.

In some embodiments, the set of regulating converters includes at least one inverted resonant-transition buck converter.

In some embodiments, the set of regulating converters includes at least one inverted resonant-transition buck converter.

In some embodiments, the set of regulating converters includes at least one flyback converter.

In some embodiments, the power combining converter circuit includes a switched capacitor circuit.

In some embodiments, the power combining converter circuit includes an interleaved switched capacitor circuit.

In some embodiments, the power combining converter circuit includes an indirect dc-dc converter topology.

In some embodiments, the power combining converter circuit includes a buck-boost converter.

In some embodiments, the power combining converter circuit includes a multi-winding flyback converter.

In some embodiments, the power combining converter circuit is configured to combine the outputs of the regulating converters to a single output.

In some embodiments, the stack of capacitors has three or more stacked capacitors and the set of regulating converters has one converter for each of the three or more stacked capacitors.

In some embodiments, the controller is configured to adapt a quantity of capacitors and regulating converters that are operative at a particular time based on a predetermined criterion.

In some embodiments, the controller is configured to adapt a quantity of capacitors and regulating converters that are operative at a particular time based on a line voltage currently being used.

In some embodiments, the sum of the voltages across the capacitors of the stack of capacitors is substantially equal to the output voltage of the line frequency rectifier.

In accordance with another aspect of the concepts, systems, circuits, and techniques described herein, a machine implemented method of operating a power converter circuit comprises: rectifying an alternating current (ac) line voltage; applying the rectified ac line voltage across a stack of capacitors, the stack of capacitors having at least two stacked capacitors; for each of the capacitors in the stack of capacitors, applying a corresponding voltage to an input of a corresponding regulating converter to convert the voltage to a regulated output signal for each of the regulating converters; modulating an input current of one or more of the regulating converters in a manner that increases power factor of the power converter circuit; and combining the regulated output signals of the regulating converters to generate a combined signal that is a desired direct current (dc) output voltage.

In some embodiments, combining the regulated output signals of the regulating converters to generate a combined signal includes providing electrical isolation between the regulated output signals and the combined signal.

In some embodiments, rectifying an ac line voltage includes full-wave rectifying the ac line voltage.

In some embodiments, the method further comprises adapting the number of capacitors in the stack of capacitors that are active based on a predetermined criterion.

In some embodiments, the method further comprises adapting the number of capacitances in the stack of capacitors that are active based on a line voltage level currently being used.

In some embodiments, modulating an input current of one or more of the regulating converters includes modulating input currents of all regulating converters that are currently active.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features may be more fully understood from the following description of the drawings in which:

FIG. 1 is a schematic diagram illustrating an example grid interface power converter architecture in accordance with an embodiment;

FIG. 2 is a schematic diagram illustrating a simplified circuit model of the front end of the grid interface power converter architecture of FIG. 1 in accordance with an embodiment;

FIG. 3 are waveform diagrams illustrating the two-phase nature of capacitor stack voltage and rectifier input current in accordance with an embodiment;

FIGS. 4, 5, 6, and 7 are waveform diagrams illustrating various current and voltage waveforms associated with a power converter circuit operating with input current modulated as a clipped sinusoid and with C₁=1 μF, C₂=50 μF, P_out=30 W, V_c1,i=10 volts, and V_c2,i=97 volts (Example 1);

FIG. 8 is a diagram illustrating the input power and the output power associated with the power converter circuit operating in accordance with Example 1;

FIGS. 9, 10, 11, and 12 are waveform diagrams illustrating various current and voltage waveforms associated with a power converter circuit operating with input current modulated as a clipped sinusoid and with C₁=1 μF, C₂=50 μF, P_out=30 W, V_c1,i=35 volts, and V_c2,i=81 volts (Example 2);

FIG. 13 is a diagram illustrating the input power and the output power associated with the power converter circuit operating in accordance with Example 2;

FIGS. 14, 15, 16, and 17 are waveform diagrams illustrating various current and voltage waveforms associated with a power converter circuit operating with input current modulated as a folded clipped sinusoid and with C₁=1 μF, C₂=50 μF, P_out=30 W, V_c1,i=35 volts, and V_c2,i=73 volts (Example 3);

FIG. 18 is a diagram illustrating the input power and the output power associated with the power converter circuit operating in accordance with Example 3;

FIGS. 19, 20, 21, and 22 are waveform diagrams illustrating various current and voltage waveforms associated with a power converter circuit operating with input current modulated as a folded clipped sinusoid and with C₁=1 μF, C₂=50 μF, P_out=20 W, V_c1,i=35 volts, and V_c2,i=71 volts (Example 4);

FIG. 23 is a diagram illustrating the input power and the output power associated with the power converter circuit operating in accordance with Example 4;

FIGS. 24, 25, 26, and 27 are waveform diagrams illustrating various current and voltage waveforms associated with a power converter circuit operating with input current modulated as a folded clipped sinusoid and with C₁=1 μF, C₂=50 μF, P_out=10 W, V_c1,i=35 volts, and V_c2,i=81 volts (Example 5);

FIG. 28 is a diagram illustrating the input power and the output power associated with the power converter circuit operating in accordance with Example 5;

FIG. 29 is a schematic diagram illustrating an example grid interface power converter architecture that includes more than two capacitors in the capacitor stack in accordance with an embodiment;

FIG. 30 is a schematic diagram illustrating an example grid interface power converter circuit that uses inverted resonant-transition buck converters as regulating converters and an interleaved switched capacitor circuit as a power combining converter in accordance with an embodiment;

FIG. 31 is a schematic diagram illustrating a control arrangement that may be used with the grid interface power converter circuit of FIG. 30 and other converters;

FIG. 32 is a waveform diagram illustrating measured voltage waveforms for a prototype converter at startup and during operation over an ac line cycle;

FIG. 33 are waveform diagrams illustrating measured ac input voltage and measured AC input current for the prototype converter;

FIG. 33A is a schematic diagram illustrating an example inverted resonant buck converter that may be used with the grid interface power converter circuit of FIG. 30 and other converters;

FIG. 33B is a plot illustrating an example operation of the inverted resonant buck converter of FIG. 33A;

FIG. 33C is schematic diagram illustrating an example control circuit that may be used to control the inverted resonant buck converter of FIG. 33A and other converters;

FIG. 34 is a schematic diagram illustrating an example grid interface power converter circuit that uses inverted resonant-transition buck converters as regulating converters and a buck-boost converter as a power combining converter in accordance with an embodiment;

FIG. 35 is a schematic diagram illustrating an example grid interface power converter circuit that uses inverted resonant-transition buck converters as regulating converters and a multi-winding flyback converter as a power combining converter in accordance with an embodiment; and

FIG. 36 is a flowchart illustrating a method of operating a power converter circuit in accordance with an embodiment.

DETAILED DESCRIPTION

FIG. 1 is a schematic diagram illustrating an example grid interface power converter architecture 10 in accordance with an embodiment. As shown, the grid interface power converter architecture 10 is characterized by a line-frequency rectifier 12, a stack of capacitors 14, a set of regulating converters 16, and a power-combining converter 18 (or a set of power combining converters). The line-frequency rectifier 12 has an input that is coupled to a grid voltage (Vin) and an output that is coupled to the stack of capacitors 14. The line-frequency rectifier 12 rectifies the grid voltage to generate a DC voltage at the output. In at least one implementation, the line-frequency rectifier 12 includes a full-bridge diode rectifier, although other rectifier configurations (such as synchronous full bridge rectifier, semi-bridge rectifier, voltage-doubler rectifier, switchable voltage-doubler/full-bridge rectifier, valley-fill rectifier) can alternatively be used. The total voltage across the capacitor stack 14 is equal to the rectifier output voltage. The capacitor stack 14 provides a portion of (or all of) any needed twice-line-frequency energy buffering, such that the power converter can provide high power factor. One or more of the capacitors in the capacitor stack 14 may have a voltage that varies over a wide range as the line voltage varies over the line cycle. One or more of the capacitors in the capacitor stack 14 may have larger value(s) than the other(s), and buffer a larger portion of the twice-line-frequency energy. The rectifier input current waveform may approximate a clipped sine wave (or, may be a different current waveform providing a desired power factor and/or instantaneous power processing), and the total capacitor stack voltage may closely follow the amplitude of the line voltage over a portion of the line cycle.

Although described above and in other places herein as a “stack of capacitors” or a “capacitor stack,” it should be appreciated that any one or more of the capacitors in the depicted “stack” can be implemented using multiple capacitors that are interconnected in a particular manner to achieve a single effective capacitance value. Thus, as used herein, the phrases “stack of capacitors” and “capacitor stack” are intended to mean a stack of capacitors where each “capacitor” may be associated with a single capacitor (as in FIG. 1) or multiple interconnected capacitors, or multiple capacitors interconnected by switches.

The set of regulating converters 16 have their inputs coupled to the capacitors of the capacitor stack 14 and provide regulated output voltages based thereon. The currents drawn by the regulating converters are modulated to draw energy from the capacitors, such that the currents drawn from the capacitor stack result in an input current waveform to the rectifier that provides high power factor while achieving the total needed energy transfer. In some embodiments described herein, the regulating converters 16 may be implemented as resonant-transition discontinuous-mode inverted buck converters. This topology enables high-frequency (HF) or very-high-frequency (VHF) operation of the regulating converters with high efficiency, low device voltage stress, small component size, and good control capability. Other types of regulating converters may alternatively be used.

The power-combining converter 18 has a plurality of inputs coupled to the regulating converter outputs. The power-combining converter 18 draws energy from the regulating converter outputs and delivers the combined power to the converter system output. The power-combining converter 18 may provide one or more of: voltage balancing among the regulating converter outputs, isolation, voltage transformation, and additional regulation of the output. The power-combining converter 18 may be a regulating or non-regulating converter structure, and may be optionally implemented as, for example, a switched capacitor circuit, a multi-input isolated converter, or as a set of single-input converters which each receive a corresponding regulating converter output and collectively supply a single power combining converter output. Other power combining converter configurations may alternatively be used.

Before introducing specific circuit topology examples for the regulating converters and the power combining converter(s), a mathematical description of steady state circuit behavior and how high power factor is achieved will be made.

FIG. 2 is a simplified circuit model 40 of the front end of the grid interface power converter architecture 10 of FIG. 1. In the model 40 of FIG. 2, two current sources 42, 44 model the average current drawn by the regulating converters (e.g., average over a switching period). The circuit cycles in two phases across a half line cycle, as shown in the FIG. 3. During phase 1, the input ac voltage is lower than the total voltage of the stacked capacitors and the full-bridge rectifier is off. During this phase, capacitors C₁ and C₂ are discharged by the regulating converters and the voltage across the capacitor stack decreases. When the input ac voltage amplitude starts to exceed the total voltage of stacked capacitors, the full-bridge turns on and the circuit enters phase 2. During phase 2, the capacitor stack voltage tracks the rectified input AC voltage, and the input current follows the sum of the currents into C₁ and regulating converter 1, as well as C₂ and regulating converter 2. When the total capacitor stack voltage discharges slowly enough by the regulating convertors that it does not follow the decreasing envelope of the ac input voltage, the total capacitor stack voltage becomes higher than the rectified ac voltage. The full-bridge then turns off and the circuit repeats the cycle starting phase 1 (see V_C1+V_C2 in FIG. 3).

Considering continuous current draw from the regulating converters (e.g. local average over a switching cycle of the regulating converters), the mathematical expressions for each phase are as follows:

• • phase 1— full-bridge is off and conducts zero input current

$\begin{array}{cc}\begin{array}{cc}{i}_{C1}\left(t\right)=C_1\frac{v_{c1}\left(t\right)}{t}=-i_1\left(t\right)& v_{C1}\left(t\right)=V_{c1,i}\int \frac{i_1\left(t\right)}{C_1}\left(t\right)\end{array}& \left(1\right)\\ \begin{array}{cc}{i}_{C2}\left(t\right)=C_2\frac{v_{c2}\left(t\right)}{t}=-i_2\left(t\right)& v_{C2}\left(t\right)=V_{c2,i}\int \frac{i_2\left(t\right)}{C_2}\left(t\right)\end{array}& \left(2\right)\\ P_0\left(t\right)=v_{C1}\left(t\right)i_1\left(t\right)+v_{C2}\left(t\right)i_2\left(t\right)& \left(3\right)\end{array}$

• • phase 2—full-bridge is on and conducts i_in(t) input current

$\begin{array}{cc}{v}_{\mathrm{in}}\left(t\right)=V\sin \omega t=v_{C1}\left(t\right)+v_{C2}\left(t\right)& \left(4\right)\\ i_{\mathrm{in}}\left(t\right)=i_{C1}\left(t\right)+i_1\left(t\right)=i_{C2}\left(t\right)+i_2\left(t\right)& \left(5\right)\\ P_0\left(t\right)=v_{C1}\left(t\right)i_1\left(t\right)+v_{C2}\left(t\right)i_2\left(t\right)& \left(6\right)\\ \begin{array}{cc}{i}_{C1}\left(t\right)=C_1\frac{v_{c1}\left(t\right)}{t}& v_{C1}\left(t\right)=V_{c1,i}-\int \frac{i_1\left(t\right)}{C_1}\left(t\right)\end{array}& \left(7\right)\\ \begin{array}{cc}{i}_{C2}\left(t\right)=C_2\frac{v_{c2}\left(t\right)}{t}& v_{C2}\left(t\right)=V_{c2,i}-\int \frac{i_2\left(t\right)}{C_2}\left(t\right)\end{array}& \left(8\right)\\ i_1\left(t\right)=\frac{1}{\frac{1}{C_1}-\frac{1v_{c1}\left(t\right)}{C_2v_{c1}\left(t\right)}}\left\{\left(\frac{1}{C_1}+\frac{1}{C_2}\right)i_{\mathrm{in}}\left(t\right)-\omega V\cos \omega t-\frac{P_0\left(t\right)}{C_2v_2}\right\}& \left(9\right)\\ i_2\left(t\right)=\frac{1}{v_{c2}\left(t\right)}\left(P_0\left(t\right)-v_{c1}\left(t\right)i_1\left(t\right)\right)& \left(10\right)\end{array}$

where i_C1(t) and i_C2(t) are the instantaneous currents through C₁ and C₂, respectively; v_C1(t) and v_C2(t) are the instantaneous voltages across C₁ and C₂, respectively; i₁(t) and i₂(t) are the currents associated with current sources 42, 44, respectively; P₀ is the output power of the simplified circuit model; and V_c1,i and V_c2,i are the initial values of v_C1(t) and v_C2(t), respectively.

Equations 9 and 10 above (which are derived from the Equations 4-8) show the currents i₁ and i₂ during phase 2 (i.e., the average current through the regulating converters 1 and 2). These equations for phase 2 show that a certain shape of input current is achieved through proper i₁ and i₂ current modulation across the line cycle. It should also be noted that the currents i₁ and i₂ are related to the instantaneous voltages across the capacitors C₁ and C₂, respectively.

Small time step numerical simulations executed over the line cycle with initial values verified the steady-state energy delivery and high power factor capability of the proposed system. The numerical simulation results, which were based on the equations above, demonstrate the effectiveness of the proposed system. In the simulation, ideal power combining may be assumed (i.e., 100% efficiency of the power combining converter). For certain capacitance values for the stack capacitors C₁ and C₂ and a given power level, there are some proper i₁ and i₂ current combinations over the line cycle that enable steady state operation with high power factor. Because there is a relationship between capacitor stack size, power level, capacitor stack voltage ripple, regulating converter voltage stress, and power factor, setting desired operating points and choosing proper regulating circuit topology are essential. The examples below briefly illustrate how the current of each regulating converter may be modulated, how the capacitor stack voltage varies, and the shape of the input current under certain requirements. In these examples, the stack capacitor values are selected as C₁=1 μF and C₂=50 μF among various sets of other parameter values.

Example 1

C₁=1 μF, C₂=50 μF, Pout=30 W, and i_in=clipped-sinusoid.

FIGS. 4, 5, 6, and 7 show the current and voltage waveforms of the model 40 of FIG. 2 when C₁=1 μF, C₂=50 μF, P_out=30 W, V_c1,i=10 volts, and V_c2,i=97 volts. The initial voltage of each capacitor is decided by predefined calculation or by internal feedback control circuitry. A sinusoidal input voltage of 170V_pk is assumed. First of all, to achieve high power factor an ideal sinusoidal current waveform is desired when the full-bridge conducts. Currents i₁(t) and i₂(t) are calculated and modulated to achieve this using equations 9 and 10. The simulation shows that with the current modulation of each regulating converter, as shown in FIG. 4, the voltages across the capacitors C₁ and C₂ vary as shown in FIG. 5. It should be noted that the capacitor stack buffers twice-line-frequency energy and both of the capacitors C₁ and C₂ maintain steady-state operation over the line cycle (i.e., the capacitor stack voltage is the same at the beginning and end of the line cycle). As described above, the full-bridge rectifier turns on and off over the line cycle. FIG. 6 illustrates the capacitor stack voltage waveform and the input voltage (v_in) waveform. FIG. 7 shows the actual clipped sinusoidal input current waveform at the rectifier input. As a result, the variations in power drawn from the line voltage are totally buffered by the stacked capacitors and the regulating converters, and the total output power of the two regulating converters is constant, as shown in FIG. 8. From this simulation, it is shown that reasonably high power factor (i.e., around 0.93) can be achieved.

Example 2

C₁=1 μF, C₂=50 μF, Pout=30 W, 35≦v_C1(t), v_C2(t)≦100, and i_in=clipped-sinusoid.

In this example, additional current control considerations are made when several restrictions are required from the regulating converter topology. First of all, it is supposed that the regulating converter to be used is only operable across a 35-100V input voltage range and only delivers power in one direction (i.e., power always flows from the capacitor stack to the power combining circuit and the regulating converter cannot draw negative current from the input). In this circumstance, modulating the current and operating the regulating converter as described in the first example above is not allowed because of the wide-range variation of v_c1 from FIG. 5 (i.e., v_c1 varies 10-80V, which is too wide a range for regulating converter 1 to operate).

FIGS. 9, 10, 11, and 12 illustrate the current and voltage waveforms satisfying the regulating converter requirements with C₁=1 μF, C₂=50 μF, P_out=30 W, V_c1,i=35 volts, and V_c2,i=81 volts. In this scenario, the calculated i₂ current would ideally go negative, requiring bidirectional power flow from the regulating converter. However, it is assumed here that i₂ should be kept positive due to the regulating converter requirements. Hence, as shown in FIG. 9, the modulating current i₂ is assigned to sit at the zero current level instead, and the resulting voltage and current waveforms are plotted as shown in FIGS. 10, 11, and 12.

As shown in FIG. 12, the input current waveform does not resemble an ideal clipped sine wave, due to the positive-clipped i₂ current waveform (i.e., the input current waveform has a small deviation from an ideal clipped sinusoidal waveform when current i₂ sits at zero). This deviation, however, has only a small effect on the clipped sinusoidal waveform shape, and thus reasonably high power factor of about 0.91 is achieved in this example.

It should be noted that because of the positive-clipped i₂ current modulation, the whole AC input energy is not buffered thoroughly in the capacitor stack, as shown in FIG. 13. To convey fixed power to the system output, the residual ac ripple energy of Pout should be buffered either at the output capacitors of the regulating converters or at the system output capacitor. Since the amount of this ac ripple energy is greatly reduced in comparison to the ac input energy, a small additional capacitance can be used for this further buffering.

Example 3

C₁=1 μF, C₂=50 μF, Pout=30 W, 35≦v_C1(t), v_C2(t)≦100, and i_in=folded-clipped-sinusoid.

This example proposes another shape of the input current waveform to achieve high power factor and ac (twice line frequency) power buffering. The capacitor stack values (C₁=1 μF and C₂=50 μF) and power level are the same as the previous example. However, instead of a clipped sinusoidal input current, a folded-clipped sinusoidal input current waveform is assumed (i.e., as shown in FIG. 17, the input current waveform is folded backward above a specific current level). In addition, V_c1,i=35 volts and V_c2,i=73 volts are used. FIGS. 14, 15, 16, and 17 illustrate the resulting i₁ and i₂ current waveforms and the voltage waveforms across each capacitor stack.

It should be noted that compared to the ac ripple on the buffered output power in FIG. 13 (which is based upon an approximate clipped sinusoidal input current waveform), the folded-clipped sinusoidal input current waveform yields better ac power buffering capability as shown in FIG. 18. Due to this folded clipped sinusoidal current waveform, the power factor is reduced to 0.9, which is nonetheless sufficient for many applications.

To compare and clarify the three examples discussed above, if the modulation currents i₁ and i₂ are always positive for a certain desired input current waveform, complete ac buffering is possible in the capacitor stack. On the other hand, with an approximate clipped sinusoidal input current waveform, if current i₁ or i₂ is assigned to be zero (instead of a negative current level), the capacitor stack does not totally buffer the ac input energy as described in the case 2 example. In this case, the folded clipped sinusoidal input current waveform helps the capacitor stack buffer the ac input energy, as shown in FIG. 18. However, there is a trade-off between the power factor and the ac energy buffer capability. Among various options of input current waveform, one can balance goals of high power factor, high power density, and requirements placed upon the regulating converter topology.

Example 4

C₁=1 μF, C₂=50 μF, Pout=20 W, 35≦v_C1(t), v_C2(t)≦100, and i_in=folded-clipped-sinusoid.

In this example, additional simulation results were obtained for operating at a different power level with the same capacitor stack values. The initial state voltage was assumed to be V_c1,i=35 volts and V_c2,i=71 volts and the step-wise simulations are presented in FIGS. 19, 20, 21, 22, and 23. In this example scenario, a power factor of 0.9 is achieved.

Example 5

C₁=1 μF, C₂=50 μF, Pout=10 W, 35≦v_C1(t), v_C2(t)≦100, and i_in=folded-clipped-sinusoid.

In this example, additional simulation results were obtained for operating at an even lower power level of 10 W with the same capacitor stack values. The initial voltages across the capacitors were set to V_c1,i=35 volts and V_c2,i=81 volts. FIGS. 24, 25, 26, 27, and 28 illustrate the resulting current, voltage, and power waveforms over the half line cycle. As before, simulation shows that a power factor of 0.9 can be achieved with sufficient input ac energy buffering operation.

It should be noted that in addition to the examples described above, numerous other techniques for modulating the current between each regulating converter with various input current waveforms may be used in different implementations. The results provided herein show that power factors up to 0.9 (and above) are achievable using the disclosed techniques, circuits, and concepts with complete energy buffering and constrained converter operation. In many applications (e.g., LED lighting for residential applications), power factors of 0.7 and above are considered sufficient, in other applications power factors of 0.8 are sufficiently high, and in still other applications (e.g., LED lighting for commercial applications) power factors of 0.9 are considered adequate. Consequently, the techniques presented herein provide means to achieve sufficient power factor in many applications.

The power converter architectures of the present disclosure have several possible advantages. One potential benefit is the decreased voltage stress to the components in the regulating converter and the power combining converter. In comparison to the common grid interface converter which directly confronts the grid voltage, each regulating converter of the proposed architecture operates at a fraction of the grid voltage because of the stacked capacitor structure (with an increased number of “stack” capacitors, this voltage stress may be further reduced.) The power combining converter, tied to the regulated outputs of the regulating converters, operates with low voltage stress range as well in some implementations.

Furthermore, power converter architectures disclosed herein are capable of providing better overall converter efficiency in many cases over prior structures. For example, in a traditional PFC circuit, such as a boost or flyback PFC circuit, the DC-DC converter stage after the PFC can dissipate a lot of power due to the high conversion ratio from the buffer capacitor voltage after the PFC circuit (e.g., around 300-400V) to the system output voltage. In the proposed architecture, on the other hand, each regulating converter can operate under much higher efficiency because of the reduced conversion ratio from each capacitor stack voltage to the regulated output voltage (e.g., about one quarter the conversion ratio in some implementations). In addition, the power combining converter of the present architecture, which may also operate from the low regulated voltage, may also provide high efficiency.

The grid interface power converter approach described herein is, in many cases, suitable for substantial miniaturization through a combination of architecture, circuit topology, and greatly increased switching frequencies. To enhance miniaturization, increases in switching frequency may be necessary because the values of inductors and capacitors vary inversely with switching frequency. However, the sizes of passive components do not necessarily decrease monotonically with frequency, owing to magnetic-core loss, voltage breakdown, and heat transfer limits. Consequently, achieving substantial miniaturization through high frequency operation further relies upon appropriate passives design and careful selection of circuit topology to minimize the demands placed upon the passive components, especially the magnetic components. It should also be noted that the capacitor stack structure can help the top regulating converter operate at high frequency, providing ac ground at the regulating converter switching frequency (i.e., the voltage across each capacitor stack changes so slowly across the line cycle that even the top regulating converter can be easily operated at high frequency).

In addition to miniaturized passive component size from high frequency operation, the behavior of grid interface architectures of the present disclosure suggests the use of smaller energy-buffer capacitors and smaller inductor component levels. Considering that the total energy stored in a capacitor

$\frac{1}{2}{\mathrm{CV}}^2$

and that the ac energy sloshes back and forth into and out of the buffer capacitor across the line cycle, a large capacitor is required to buffer this ac sloshing energy with low voltage variation across the capacitor

$\left(i.e.,\frac{1}{2}{C\left(V+\Delta V\right)}^2-\frac{1}{2}{\mathrm{CV}}^2=\mathrm{ac}\mathrm{sloshing}\mathrm{energy}\right)$

In the proposed grid interface architecture, the stacked capacitors right after the full bridge rectifier buffer the sc energy, and can be controlled to large voltage variations by active current modulation of the two regulating converter legs. This large voltage variation, as already shown in the simulation results, allows smaller capacitors to be used (e.g., about 30-200 μF for designs operating at tens of Watts from 120 Vac) to store the ac sloshing energy.

Additionally, a smaller inductor can be used with the reduced impedance level at the regulating stage as compared to a conventional converter. Converters operating at high voltages and low currents operate at high impedance levels, and consequently utilize relatively large inductors and small capacitors (e.g., characteristic impedance

$Z_0=\sqrt{L/C}$

scales as V/I). Furthermore, inductor and capacitor values scale down with increasing resonant frequency

$\left(e.g.,f=1/\sqrt{\mathrm{LC}}\right).$

Thus, for a given topology, increasing frequency beyond a certain point may lead to capacitance values that are too small to be practically achievable (i.e., given parasitics), placing a practical bound on frequency and miniaturization. For miniaturization of converters at relatively high voltage and low power, it is preferable to select system architectures and circuit topologies that require relatively low characteristic impedance values (i.e., small inductances and large capacitances) to reduce constraints on scaling up in frequency. The proposed architecture roughly divides the input voltage range of each regulating converter by two (or more), and thus decreases the inductance level by four with a given capacitance. Consequently, the entire converter with two regulating converters can be designed with smaller inductor values and size. As will be described in greater detail, in some embodiments, more than two capacitors may be used in the capacitor stack, further reducing required individual inductance values and enabling higher frequencies to be employed.

In at least one implementation, bi-directional grid interface power converters are provided that can convert from ac to dc and also from dc to ac. Such bi-directional converters require proper device and topology selection and control. In at least one embodiment, for example, a line frequency bridge may be provided that also serves as an inverter for dc to ac conversion (e.g., by replacing diodes with active devices). Control circuitry is also provided to control operation of the inverter. Power converter circuits that only serve as do-to-ac converters may also be produced.

As described previously, the grid interface power converter architecture 10 of FIG. 1 comprises a line-frequency bridge rectifier 12 with a stack of two capacitors 14 across its output; a pair of regulating converters 16; and a power combining converter 18 having two inputs and an output that supplies the system output. It should be understood that certain modifications and variations may be made to this architecture without departing from the spirit and scope of the disclosure. For example, modifications may be made to the number of each different component in the architecture, to the configuration for interconnecting the different components, and to the types of circuit elements used to implement each component. Likewise, one or more elements may be added to and/or removed from the architecture in different implementations.

FIG. 29 is a schematic diagram illustrating a grid interface power converter architecture 80 that includes a capacitor stack 82 having three capacitors.

Correspondingly, the power converter architecture 80 also includes three regulating converters 84, 86, 88 which are each coupled at an output thereof to a power combining converter 90. In some implementations, the number of capacitors and sub-regulating-converters that are active may be allowed to vary dynamically or adapt over time based on a predetermined criterion (e.g., based on whether the line voltage is currently 120 or 240 V_rms or based upon the ac rms voltage amplitude within a range This approach provides greater flexibility of operation and narrower component operating ranges than could otherwise be achieved. The variation may be provided, for example, by providing circuit structures that allow one or more of the stack capacitors to be controllably shorted out. Other techniques may alternatively be used. It should be appreciated that this architecture may be further extended to use even more capacitors in the capacitor stack. Such an extension might be desirable in designs for “universal input” (e.g., operable from 85-265 V_ac,rms).

FIG. 30 is a schematic diagram illustrating a prototype grid interface power converter circuit 100 in accordance with an embodiment. The prototype grid interface power converter circuit 100 was built and tested and found to operate at a high performance level. As illustrated, the grid interface power converter circuit 100 includes: a line-frequency rectifier 102, a capacitor stack 104, a set of regulating converters 106, and a power-combining converter 108. To achieve high efficiency and high power density, the regulating converters 106 were designed as inverted resonant-transition buck converters and the power combining converter 108 was designed as an interleaved switched capacitor circuit. The regulating converters 106 were each designed with a single switch, diode, and small inductor and operate in the HF frequency range (i.e., 3- 30 MHz). The buck converters are “inverted” in the sense that they were designed with “common positives.” An advantage of this is that the active switch control ports are each referenced to slowly-moving node voltages, simplifying the task of driving them at high frequencies.

For much of their operating range, the regulating converters 106 act like quasi-square-wave zero-voltage switching (ZVS) buck converters with a low ratio of switching to resonant frequency. Outside of this range, the converters do not operate with perfect zero-voltage switching, but rather provide near ZVS operation with low turn-on loss of the active switch. Each regulating converter takes as an input one of the capacitor voltages from the capacitor stack and provides a regulated voltage across its output capacitor. This regulating converter design has several benefits. First, it operates with ZVS or near-ZVS soft switching across the 35-100 V wide input voltage range. The single common referenced switch (referenced to a slowly-moving potential) makes it suitable for operation at HF. Second, it requires only a single, small-valued inductor. Furthermore, it has very fast response (near single cycle) to input voltage transients and changes in the output current command. Finally, for a given input voltage, the output current is roughly proportional to transistor on-time, allowing a variety of control schemes to be employed.

The power combining converter 108 in the embodiment of FIG. 30 is implemented as an “interleaved” switched capacitor circuit. The interleaved switched-capacitor circuit is an effective choice for high efficiency and power density, as the converter doesn't need to provide regulation. The switched capacitor circuit draws energy from the two regulating converter outputs and supplies a single system output (which is also the output of one of the regulating converters 106). Because the switched capacitor circuit transfers charge without voltage regulation, and is designed with switches and capacitors, it can be operated with high efficiency at low frequency, with small converter size. In the switched capacitor power combining converter 108 of FIG. 30, the capacitors C_f1 and C_f2 transfer charge from capacitor C_R1 to capacitor C_R2 and supply the combined power to the load. Because the load is connected across the output of one of the regulating converters, the switched capacitor power combining converter 108 only processes a portion of overall system energy. Moreover, if C_f1 and C_f2 are selected as much larger than C_R1 and C_R2, partial “soft charging” of the energy transfer capacitors can be achieved.

To test the prototype power converter circuit 100, an example system was implemented based on the topology describe above. The system was designed to supply 30 W to a 35V DC output from a 120 V_rsm AC 60 Hz input. The converter circuit 100 can buffer the AC energy with C1=1 μF and C2=50 μF at the 30 W power level. To select practical capacitor values, it was recognized that the capacitance of high-k ceramic capacitors degrades with bias voltage. Consequently, a 1 μF, 100V, X7R capacitance was used for C1 and a (nominal) 225 μF, 100 V X7S ceramic capacitor was used for C2. For the regulating converters, the prototype circuit used a gallium nitride (GaN) switch (EPC 2012) for Q_R1 and Q_R2; a Schottky diode (STPS30120DJF) for diode D; and an 800 nH inductance using 10 turns on a Micrometals P68-106 core for L. For the power combining convertor 108, the prototype circuit used a GaN switch (EPC 2012) for the switches, and a 20 μF, 100V, X7R ceramic capacitor for C_f1 and C_f2. For control, the prototype used an Atmel ATtiny 1634 microcontroller. With the above values, the regulating converters were able to operate in the range of 5-10 MHz under soft-switching or near-soft-switching conditions over their input voltage and power range. The power combining convertor 108 operates at a fixed switching frequency of 30 kHz (although a variable frequency, e.g., proportional to operating power, could be employed).

Among other possible control functions, the microcontroller was used to control the average current of each regulating converter and to synchronize operation over the line cycle. The microcontroller was also used to, for example, detect the zero-voltage crossing of the ac line voltage and reset the time state. The microcontroller may also monitor the capacitor stack voltage (e.g., the input voltage of each regulating converter) and send updated pro-defined switch-turn-on duration information to each regulating converter at periodic intervals (e.g., every 80 μS or so) in response thereto. In addition, the microcontroller may generate a 30 kHz 50% duty ratio switching signal (with dead time) for the combining converter. As will be appreciated, the functions of the microcontroller may vary from implementation to implementation based on the specific types of circuits that are used as the regulating converters and the power-combining converter.

The prototype converter was designed to support up to 30 W output at 35V from 120 Vac. FIG. 32 shows measured voltage waveforms of the prototype converter at startup and during operation over the ac line cycle, when powered from an Agilent 6812B ac power supply. The waveform labeled V_C1 represents the voltage across capacitor C₁ and the waveform labeled V_C2 represents the voltage across capacitor C₂. As shown, V_C2 fluctuates about 50 V over the line cycle in this prototype converter as it buffers the line-frequency energy. FIG. 33 shows the measured input AC voltage and current for the prototype converter. The prototype converter displayed 93.3% efficiency with 0.89 power factor for a 35Vdc, 30 W load.

The volume of the prototype converter was measured and the “box power density” was calculated to be 25 W/in³. It is notable that this is much higher than the approximately 5 W/in³ found for typical commercial LED drivers at this power rating, even though the component layout was not optimized for box volume power density. The displacement volume was 0.23 in³ yielding a “displacement power density” of 130 W/in³. This illustrates the high power density achieved, and that layout can readily provide a greatly improved “box power density.”

FIG. 31 is a diagram illustrating how a microcontroller may be coupled to various elements of a power converter circuit in accordance with an embodiment. Although illustrated in FIG. 31 as a microcontroller, it should be appreciated that many other types of processors and/or circuits may be used to provide control functions in a power converter in different embodiments. For example, in various embodiments, the control functions may be performed by: a general purpose microprocessor, a digital signal processor (DSP), a reduced instruction set computer (RISC), a complex instruction set computer (CISC), an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a programmable logic array (PLA), a microcontroller, an embedded controller, a multi-core processor, a processor complex, and/or others, including combinations of the above. One or more analog control circuits may also be used in addition to, or as an alternative to, digital processing structures. For example, the controls for the inverted resonant-transition buck converters may use comparators to detect voltage ringdown for soft switching, and may employ analog time-based control to realize the PWM. In various embodiments, techniques, systems, and circuits described herein may be implemented using any combination of hardware, software, and firmware.

There are various way to control the proposed circuit architecture. As a first option, the control circuitry can save the predefined current waveform for various power levels, and modulate regulating converters to draw predefined current waveform. As an alternative option, the separate feedback control circuits for each regulating converter can be utilized to regulate each output of regulating converter, and the proposed system is controlled by selecting which regulating converter to turn on. The input current waveform is decided by the power level and the instantaneous operating voltage of each regulating converter.

There are also various ways to control the inverted resonant buck converter of the proposed circuit architecture, like example inverted resonant buck converter shown in FIG. 33A. Illustrated in FIGS. 33B and 33C, for example, are an example operation of the inverted resonant buck converter shown in FIG. 33A and associated control circuitry for achieving said operation, respectively.

In operation, as shown in FIG. 33B, the inverted resonant buck converter (e.g., of FIG. 33A) cycles through four phases of a HF frequency cycle (denoted by phase 1, phase 2, phase 3, and phase 4), like the HF frequency cycle shown in FIG. 7. In particular, in phase 1 of the HF frequency cycle, a switch (e.g., S_hf, shown in FIG. 33A) of the inverted resonant buck converter is toggled on and, as a result, inductor current (e.g., i_L) of inductor (e.g., L, shown in FIG. 33A) ramps up linearly. In phase 2, the switch (e.g., S_hf, shown in FIG. 33A) is toggled off and, as a result, switch drain to source voltage (e.g., v_ds, shown in FIG. 33A) of the inverted resonant buck converter increases to a same voltage level as the input voltage (e.g., V_int, shown in FIG. 33A), which is substantially the same as or similar to an HF regulation stage input voltage. In phase 3, a diode (e.g., D, shown in FIG. 33A) begins to conduct current and, as a result, current flowing through the inductor (e.g., L, shown in FIG. 33A) ramps down to zero. In phase 4, the switch (e.g., S_hf, shown in FIG. 33A) is toggled off and, as a result, the diode (e.g., D, shown in FIG. 33A) stops conducting current and the inductor (e.g., L, shown in FIG. 33A) rings with a net capacitance observed at a switch drain node (not shown) of the inverted resonant buck converter, wherein the net capacitance is representative of switch output capacitance combined with diode output capacitance. Additionally, in phase 4 current flowing through the inductor (e.g., L, shown in FIG. 33A) ramps to a negative current value and the switch drain to source voltage (e.g., vds, shown in FIG. 33A) ramps down to or near a potential at or around zero volts (e.g., down to V_int-2V_out). Subsequently, the switch (e.g., S_hf, shown in FIG. 33A) is toggled on and the cycle repeats again starting at phase 1.

FIG. 33C illustrates example control circuitry for achieving the example operation discussed above in conjunction with FIG. 33B, particularly in achieving a particular peak current in inductor L of the inverted buck converter shown in FIG. 33A and other similar converters. As shown, the example control circuitry comprises a microcontroller, a capacitor (C), a plurality of resistive loads (R), a logic gate, and two comparators, U₁ and U₂. It is to be appreciated that, in alternate embodiments, the control circuitry may, for example, comprise greater than or fewer than two comparators.

The control circuitry is configured to adjust output power (e.g., V_out, shown in FIG. 33A) of the inverted resonant buck converter by altering on-time (e.g., t_on, shown in FIG. 33B) of a switch (e.g., S_hf, shown in FIG. 33A) of the inverted resonant buck converter in any given phase (e.g., phase 4) of the HF frequency cycle. In particular, by specifying a peak amount of current (e.g., i_L) flowing through inductor (e.g., L, shown in FIG. 33A) on a cycle by cycle basis (e.g., complete cycle through phases 1 to 4 of the HF frequency cycle), average amount of current flowing through the inverted resonant buck converter can be set with a wide bandwidth range, especially when the control circuitry cycles at a frequency in the multi-megahertz range. Additionally, by specifying a peak amount of current flowing through the inductor (e.g., L, shown in FIG. 33A) on a cycle by cycle basis, short circuit protection of the inverted resonant buck converter can be achieved.

With knowledge of input voltage (e.g., V_int, shown in FIG. 33A) and peak current of the inductor (e.g., L, shown in FIG. 33A) of the inverted resonant buck converter, average input current can be set for any given load (R) with a constant output voltage (e.g., V_out, shown in FIG. 33A). Additionally and alternatively, with knowledge of input voltage (e.g., V_int, shown in FIG. 33A) and output voltage (e.g., V_out, shown in FIG. 33A), average input current and average output current can be determined. The aforementioned can be found particularly helpful in providing for power factor correction to the proposed circuit architecture discussed above. Specifically, the average input current can be set proportional to the input voltage (e.g., V_int, shown in FIG. 33A) and, in doing so, generate a unity of power factor for power factor correction. If power factor correction is not required, however, arbitrary input current waveforms can be suitably used for the average input current.

It is to be appreciated that the control circuitry of FIG. 33C can also be implemented in an open loop or forward feedback arrangement. As such, with there being no need to know input voltage (e.g., V_int, shown in FIG. 33A) or output voltage (e.g., V_out, shown in FIG. 33A) values of the inverted resonant buck converter, and hence no need to measure said input or output voltage values, cost, complexity, and power loss often associated with conventional control circuitry for inverted resonant buck converters (e.g., of FIG. 33A) and the like can be significantly reduced.

FIGS. 34 and 35 are schematic diagrams illustrating a number of grid interface power converter circuits in accordance with different embodiments. In each of these embodiments, a different combination of structures is used for the regulating converters and the power-combining converter. In the embodiment of FIG. 34, for example, the regulating converters are implemented as inverted resonant-transition buck converter and the power-combining converter is implemented as a buck-boost converter. In the embodiment of FIG. 35, the regulating converters are again implemented as inverted resonant-transition buck converters, but the power-combining converter is implemented as flyback converter. It will be recognized that other topologies may likewise be used. Various regulating converters can be employed, such as resonant transition buck converters, soft-switched synchronous buck converters, quasi-square-wave synchronous buck converters, tapped-inductor converters, indirect converters, etc.

Likewise the power combining converter could comprise a converter that takes energy from one (or more) regulating converter multi-input indirect converter outputs and provides it to a single regulating converter output. This could be done with a variety of isolated converter types, including flyback converters or resonant converters. The power combining converter may be realized as an isolated multi-winding converter (e.g., multi-input converter) with input-side switches of different inputs operated synchronously. This may be done with a flyback topology (as in FIG. 35), as a different multi-input PWM converter, as a multi-input resonant converter (e.g., multi-input LLC converter), etc.

FIG. 36 is a flowchart illustrating a method 120 of operating a power converter circuit in accordance with an embodiment. First, an ac line voltage is rectified (block 122). In some embodiments, full wave rectification is performed, but other rectification techniques may alternatively be used. The rectified ac line voltage is then applied to a stack of two or more capacitors (block 124). For each capacitor in the stack, a voltage is applied to an input of a corresponding regulating converter, which converts the voltage to a regulated output signal (block 126). Input currents of one or more of the regulating converters are modulated in a manner that increases the power factor of the power converter circuit (block 128). In some implementations, input currents of all active regulating converters are modulated. In some embodiments, the input currents are modulated in a manner that results in an input current of the power conversion circuit taking the form of a clipped sinusoid or a folded clipped sinusoid. The regulated output powers of the regulating converters are combined to form a combined power and the combined power is converted to a desired dc output voltage (block 130). In some embodiments, the number of capacitances in the stack that are active is adapted over time based on a predetermined criterion (e.g., line voltage level being used, etc.).

Although various circuits and systems described herein are referred to as grid interface power converters, it should be appreciated that the techniques, principles, features, and structures described herein also have application in systems that do not involve a grid interface. That is, the concepts described herein have application in any number of different power conversion scenarios and may involve conversion from ac to dc, conversion from dc to ac, or both.

Having described example embodiments of the invention, it will now become apparent to one of ordinary skill in the art that other embodiments incorporating their concepts may also be used. The embodiments contained herein should not be limited to disclosed embodiments but rather should be limited only by the spirit and scope of the appended claims. All publications and references cited herein are expressly incorporated herein by reference in their entirety.

Claims

1. A power converter circuit comprising:

a line frequency rectifier circuit configured to rectify an alternating current (ac) input voltage of the power converter circuit;

a stack of capacitors coupled to receive an output signal of the line frequency rectifier circuit, the stack of capacitors having at least two stacked capacitors;

a set of regulating converters coupled to the stack of capacitors, each regulating converter in the set of regulating converters being configured to draw current from a corresponding one of the stacked capacitors and to generate a regulated voltage output power at an output thereof in response thereto; and

a power combining converter circuit configured to combine the regulated voltage output powers of the set of regulating converters and to convert the combined power to a desired direct current (dc) output voltage.

2. The power converter circuit of claim 1, further including:

a controller configured to controllably modulate input current of one or more of the regulating converters in the set of regulating converters in a manner that results in a power factor at the ac input of the power converter circuit that is greater than or equal to 0.8 for at least some operating conditions.

3. The power converter circuit of claim 1, further including:

a controller configured to controllably modulate input current of all of the regulating converters in the set of regulating converters in a manner that enhances power factor at the ac input of the power converter circuit.

4. The power converter circuit of claim 1, wherein:

the set of regulating converters includes at least one inverted resonant-transition buck converter or synchronous resonant-transition buck converter.

5. The power converter circuit of claim 1, wherein:

the set of regulating converters includes at least one buck converter.

6. The power converter circuit of claim 1, wherein:

the stack of capacitors buffers a substantial portion of a twice-line-frequency energy.

7. The power converter circuit of claim 6, wherein:

one or more capacitors in the stack of capacitors has a capacitance value that is substantially greater than capacitors in the stack of capacitors not including the one or more capacitors, wherein the one or more capacitors buffers a greater portion of the twice-line frequency than the capacitors in the stack of capacitors not including the one or more capacitors.

8. The power converter circuit of claim 1, wherein

the power combining converter circuit includes a switched capacitor circuit.

9. The power converter circuit of claim 1, wherein:

the power combining converter circuit provides electrical isolation.

10. The power converter circuit of claim 1, wherein:

the power combining converter circuit includes a buck-boost converter.

11. The power converter circuit of claim 1, wherein:

the power combining converter circuit includes a flyback converter.

12. The power converter circuit of claim 1, wherein:

the power combining converter circuit is configured to combine the regulated voltage output powers of the regulating converters to a single output.

13. The power converter circuit of claim 1, wherein:

the stack of capacitors has three or more stacked capacitors and the set of regulating converters has one converter for each of the three or more stacked capacitors.

14. The power converter circuit of claim 12, wherein:

the controller is configured to adapt a quantity of capacitors and regulating converters that are operative at a particular time.

15. The power converter circuit of claim 12, wherein:

the controller is configured to adapt a quantity of capacitors and regulating converters that are operative at a particular time based, at least in part, on a line voltage currently being used.

16. A machine implemented method of operating a power converter circuit comprising:

rectifying an alternating current (ac) line voltage;

applying the rectified ac line voltage across a stack of capacitors, the stack of capacitors having at least two stacked capacitors;

for each of the capacitor in the stack of capacitors, applying a corresponding voltage to an input of a corresponding regulating converter to convert the voltage to a regulated output signal;

modulating an input current of one or more of the regulating converters in a manner that increases power factor of the power converter circuit; and

combining the regulated output signals of the regulating converters to generate a combined signal and converting the combined signal to a desired direct current (dc) output voltage.

17. The method of claim 16, wherein:

applying the rectified ac line voltage across a stack of capacitors comprises storing energy in the stack of capacitors, wherein the energy stored in the stack of capacitors is utilized to buffer a substantial portion of twice-line-frequency energy processed by the system.

18. The method of claim 16, further comprising:

adapting a select number of capacitors in the stack of capacitors that are active based on a predetermined criterion.

19. The method of claim 16, further comprising:

adapting a select number of capacitors in the stack of capacitors that are active based on a line voltage level currently being used.

20. The method of claim 16, further comprising:

providing electrical isolation between the regulated output signals of the regulating converters and the dc output voltage of the machine implemented method of operating a power converter circuit.

21. The method of claim 16, further comprising:

providing switched capacitive energy transfer to combine the output signals of the regulating converters to provide the do output voltage of the system.