Multiphase converter with zero voltage switching

Abstract

A multiphase DC-to-DC converter includes at least two phase circuits each having upper and lower power switches and a front-end inductor that is operative for forming a resonant tank circuit with the phase circuits to ensure zero voltage switching and minimizing power losses.

Description

RELATED APPLICATION

This application is based upon prior filed copending provisional application Ser. No. 60/538,091 filed Jan. 21, 2004.

FIELD OF THE INVENTION

The present invention relates to the field of electronic circuits, and more particularly, to DC-to-DC converters and switch mode power supplies for example, Buck converters.

BACKGROUND OF THE INVENTION

DC-to-DC converters typically are designed as switching-regulated power supplies also known as a switch-mode power supplies. Some DC-to-DC converters raise voltage from a lower input voltage (a step-up converter), and others lower voltage from a higher input voltage (a step-down converter). One type of step-down switch mode power supply that lowers the voltage is known as a Buck converter, a type of switch mode regulator/or switching power supply. These devices resemble a linear power supply in some respects, but in other ways they are much different. A switching power supply typically includes an energy-storage inductor, and sometimes a non-linear regulator network. This type of power supply can incorporate a regulation system in which a control element, for example, power MOSFET switches, are switched on and off rapidly. The on/off pulses can be controlled by an oscillator/error amplifier/pulse-width modulator network as a controller. Thus, in a more common variety of switching regulator, the transistor switch, for example, the MOSFET, is a control element.

During an ON cycle, energy can be pumped into an inductor and stored in magnetic fields. When the control element is turned OFF, the energy is stored and the inductor is directed by a diode into a filter and load. Various sampling circuits can sample the output voltage and feed a sample to an input of an error amplifier as part of a controller. The sample voltage can be compared with a reference voltage and an error amplifier can increase its output control voltage, which is sent to a pulse-width modulator. The pulse-width modulator produces a modified ON/OFF signal, for example, sometimes a square wave whose time is determined by the input error voltage.

More specific examples of DC-to-DC converters as switch mode power supplies are disclosed in commonly assigned, published U.S. patent application nos. 2003/0038614 and 2004/0070382, the disclosures which are hereby incorporated by reference in their entirety. As noted before, a Buck converter is a specific type of step-down, DC-to-DC converter.

To power various microprocessors, and more particularly the next generation microprocessors, requiring about one volt and up to 1,000 amps current, the number of phases in a multiphase Buck converter has been increasing, sometimes requiring as many as eight phases. The optimum number of phases can be determined by the output current, system efficiency, transient requirements, thermal management, cost of capacitors, MOSFET performance, size restriction and overall system costs. A controller for Buck converters is complicated and typically is designed as a multiphase PWM control integrated circuit with companion gate drivers, e.g., the HIP6301, HIP6601B, HIP6602B, HIP6603B, or HIP6604B and external MOSFET's, for example as manufactured by the assignee of the present invention, Intersil Americas Inc.

Multiphase power conversion is an improvement over earlier single phase converter configurations and is used to satisfy the increasing current demands of modern microprocessors. Multiphase converters distribute the power and load current, which results in smaller and lower cost transistors with fewer input and output capacitors. This occurs because of higher effective conversion frequency with higher frequency ripple current and phase interleaving. Each phase circuit typically includes a lower MOSFET and an upper MOSFET as power switches. The requirement for decreasing the size of the converter along with the requirement for higher power densities requires an increase in the switching frequency used in the power converter. The use of a high switching frequency, in these multiphase DC-to-DC converters, and especially Buck converters, however, leads to switching losses, stresses on the power component, and EMI generation.

SUMMARY OF THE INVENTION

The present invention is advantageous and improves the efficiency of a switch mode power supply, DC-to-DC converter because it is operable for zero voltage switching and can be used for non-isolated high input/low voltage output voltage converters, such as a Buck converter. The present invention uses a resonant tank circuit for a multiphase topology. The Vout/Vin DC transfer function depends on N number of phases. With the present invention, it is possible to achieve higher than normal output ripple cancellation than with existing Buck topologies.

In the present invention, the duty cycle is no longer a function of only the time ON but it is a function of the time ON and the number of phases (N). The present invention detects the zero crossing, for example, using a PWM controller or other Buck controller. In the present invention, it is possible to create a zero voltage across the upper MOSFET before the MOSFET's are turned ON or OFF. A resonant tank is created that achieves zero voltage across the MOSFET's before they are turned ON or OFF as part of the improved topology. The front-end inductor creates a desired resonant tank circuit.

Typically the MOSFET has an inherent parasitic capacitance as part of a resonant tank. If the inherent parasitic capacitance is too small, it is possible to add a capacitor. A diode can also be added if the intrinsic diode capability of a MOSFET is insufficient.

In accordance with the present invention, the inductor at the front end does not allow the current to increase until a MOSFET is fully ON. There is no overlapping of current until the MOSFET turns ON. As to the inductor, its transition is smoother and the diode is slowly turning OFF instead of switching. Thus, it can be seen that there is zero current across the upper MOSFET and zero voltage across the lower MOSFET. The inductor resonates with any capacitors of the upper MOSFET's. Because of the resonant tank circuit, the time ON is fixed, but the time ON can vary according to what the controller signals. A total period for each phase is changing and time is variable, notably because the time ON is variable by the controller. The present invention is also operable because there is a time period when all lower MOSFET's are ON, and that time period is taken advantage of because of the resonance.

In accordance with the present invention, a multiphase DC-to-DC converter includes at least two phase circuits, each having upper and lower power switches and a front end inductor operative for forming a resonant tank circuit with the phase circuits to ensure zero voltage switching and minimizing power losses. The converter includes a controller operative with the phase circuit for detecting a zero volt crossing. The controller could be a PWM controller or other Buck controller. The resonant tank circuit is created to achieve zero voltage across the power switches, which typically are formed as field effect transistors. The converter could include a feedback signal processing circuit operative with each phase circuit and an output capacitor operative with the voltage output from the phase circuits. A capacitor can be operative with at least each upper power switch and lower power switch. A diode can also be operative with the upper power switch and lower power switch. These capacitors and diodes can be added if the intrinsic capacitance or diode function of the power switch is not enough to form the resonant tank circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

Other objects, features and advantages of the present invention will become apparent from the detailed description of the invention which follows, when considered in light of the accompanying drawings in which:

FIG. 1 is a schematic circuit diagram of a multiphase switch mode power supply, i.e., a DC-to-DC converter, and showing a front-end inductor to form a single resonant tank for a multiphase topology of the present invention.

FIG. 2 is a timing diagram for N phases in accordance with the present invention.

FIG. 3 is a graph showing Vout/Vin as a function of theta (θ) and N.

FIG. 4 is a graph showing Vout/Vin as a function of theta (θ) and N.

FIG. 5 is a schematic circuit diagram illustrating an example of a circuit function relative to Mode 1 for the multiphase DC-to-DC converter of the present invention.

FIG. 6 is another schematic circuit diagram similar to the circuit shown in FIG. 5, but showing a Mode 2 operation.

FIG. 7 is a graph showing time relative to VSW and VSW_Vin and phase 1, phase 2 and phase N.

FIG. 8 is a graph showing simulation results for three phases that all switch at zero voltage.

FIG. 9 is a schematic circuit diagram showing a two-phase circuit similar to that shown in FIG. 1, but showing greater details of functional components, which could be intrinsic to the power components.

FIG. 9A is an equivalent circuit of FIG. 9, showing current IO1 and IO2 in respective phase structures (circuits).

FIG. 10 is an equivalent circuit of the present invention showing its function prior to Mode 1.

FIG. 11 is an equivalent circuit of the present invention showing its function at Mode 1.

FIG. 12A is an equivalent circuit of the present invention showing its function at Mode 2.

FIG. 12B are formulas depicting operation of the circuit shown in FIG. 12A.

FIG. 13A is an equivalent circuit of the present invention showing its function at Mode 3.

FIG. 14 is a graph of a state plane diagram in accordance with the present invention.

FIGS. 15-18 are examples of equivalent circuits similar to the circuits shown in respective FIGS. 10, 11, 12A, and 13A and showing functional operation with different modes and the times.

FIG. 19 is a graph showing the conservation of energy relative to time in accordance with the present invention.

FIG. 20 is a three-dimensional graph showing duty versus the number of phases and the time ON.

FIG. 21 is a graph depicting results of a spice test.

FIG. 22 is a schematic circuit diagram showing an example of a spice model set-up that can be used for modeling the present invention.

FIG. 23 is a graph showing a state plane, full load diagram.

FIG. 24 is a graph showing a no load state diagram.

FIG. 25 are graphs comparing efficiencies of hard and the soft switching of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention will now be described more fully hereinafter with reference to the accompanying drawings, in which preferred embodiments of the invention are shown. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the invention to those skilled in the art. Like numbers refer to like elements throughout, and prime notation is used to indicate similar elements in alternative embodiments.

The present invention improves the overall efficiency of the DC-to-DC converter system because zero voltage switching can be used for non-isolated high input voltage, and low output voltage power converters, for example, “Buck converters.”

The need for decreasing the size of the converter, along with the need for higher power densities, implies an increase of switching frequency used in the power converter. The use of high switching frequency, however, leads to switching losses, imparted stresses on the power components, and EMI generation. To overcome this disadvantage, soft switching zero voltage switching is used in the present invention.

FIG. 1 is a fragmentary, block diagram of a portion of the multiphase “Buck” converter 30 as a DC-to-DC converter that includes an output inductor 32 coupled between the load for Vout and a node between the high and low side power switches 34, 36 that are connected together. The high and low side power switches 34, 36 are also termed upper and lower power switches. The different phases circuits 40 are cascaded and terminate at phase N as illustrated, to form phase circuits, 40, 40a . . . 40N. The phase circuits include appropriate inputs and outputs 42, 44. PWM drivers 50 are operative with the power switches 34, 36 and could each include a feedback signal processing circuit 52. Capacitors 54, 55 can be placed in parallel with the power switches as illustrated, including an output capacitor 56 connected in parallel across the load. The power switches each could have intrinsic capacitance, and capacitors may not be required.

In accordance with the present invention, to have zero volt switching, an inductor 60 is placed in front of the normal switching circuit at the front end of the circuit, as illustrated, and receives input voltage from the input voltage source 61. The control scheme is also changed to detect zero voltage, as will be explained in further detail below. The inductor 60 is resonating with the capacitors 54 of the upper MOSFET's in each of the phases.

FIG. 9 is a schematic circuit diagram similar to FIG. 1 but showing in greater detail a first and second phase structure on circuit 40, 40a, which are cascaded. Also, each of the field effect transistors 34, 36 as a power device includes a diode 62, 63 labelled D1 in the first phase circuit and D2 in the second phase circuit and given the designations up or down for association with upper or lower power switches. The diodes could be body diodes. The field effect transistors 34, 36 would have some intrinsic diode capability, but additional diodes as body diodes, could be added as necessary for achieving desired inductance as explained below. FIG. 9a shows an equivalent circuit structure without the output and showing IO1 and IO2 in the respective phase structures as currents. Similar circuit components are given the same reference numerals except with the letter “a” in the cascaded or second phase circuit or structure shown in FIGS. 9 and 9a. The upper and lower switches throughout the drawings are given the designation up and low, with the first phase circuit switches as S1 and the second phase circuit switches as S2. Upper and lower capacitors and diodes are given an association in some of the drawings to distinguish upper and lower and the phase circuits.

FIG. 2 shows a timing diagram of the present invention using the front-end inductor 60 for n-phases with time ON and time OFF shown relative to phases 1, 2 and phase n such that total time as used with a duty cycle is equal to the number of phases n times the time ON plus the number of phases times the time OFF.

FIG. 3 shows a graph such that the vertical axis shows Vout over Vin, and the horizontal axis shows the number of phases on the right side and and theta (θ) on the left as time ON over time OFF. Thus, FIG. 3 shows that the voltage OUT divided by the voltage IN is a function of theta (θ) and “N”, i.e., the number of phases.

FIG. 4 shows in greater detail that the Vout over the Vin as a function of theta (θ) and N.

FIG. 5 shows a conceptual schematic circuit drawing of the function of input inductor of FIG. 1 in mode 1 with the upper capacitor and showing current flow.

FIG. 6 is a similar drawing but showing a mode 2 operation. FIG. 7 is a graph showing the zero volt switch point as a predetermined time and the various VSW points and upper_drive relative to different phases. FIG. 8 shows the simulation results for three phases, all shown switching at zero voltage.

With reference again to FIGS. 1 through 8, assume that there are N phases, such as shown in FIG. 1, and that the output inductor 60 is large enough to assume constant current source. At t=O the cycle starts. The upper MOSFET 34 of phase 1 is on and all the (N−1) upper MOSFET 36 are off, while all the (N−1) lower MOSFET are on. The operation will differ depending on the modes and the time (+). With mode 1, at t=ton, the phase 1 upper MOSFET will be turned off and the body diode 63 of its lower MOSFET turned on. As a result, the lower MOSFET of phase 1 would be turned on at zero voltage. After that time, all the power MOSFET of the N phases are on and the upper MOSFET's are off. At mode 2, with N*Cr capacitor will start resonating with Lr 60 and the resonant time is toff, when the VSWIN and VSW are equal phase 2 can be turned on at zero voltage. The cycle continues for all N phases. When the switch turns off and all the power MOSFET are ON, the next mode of operation starts.

The circuit was simulated as shown in the functional circuit representations shown in FIGS. 5 and 6 and the following results were obtained as shown in the simulation graph of FIG. 8. As can be seen from the simulation results in this graph, the three phases shown all switch at zero voltage. The steady state analysis of the converter shows that: $\underset{\square }{\frac{\mathrm{Vout}}{\mathrm{Vin}}=\frac{1}{n\times \theta +n}}$

Where θ is ton/toff, for example, as shown in FIG. 3, for a higher input voltage and lower output voltage, it is possible to use more phases to achieve a practical duty cycle without the requirement for the down stage. The two stage (or phase) circuit is shown in FIG. 9, and the equivalent of that two phase circuit is shown in FIG. 9A. As illustrated, two phase structures are shown with the addition of a diode 62, 63 for each power switching phase structure, including a D1 up diode 62 and a D1 low diode 63 and parallel with the capacitors 54, 56 and parallel with the power switches 34, 36. As illustrated, the CR1_up capacitor 54 is in parallel with D1_up diode 62. For the other phase structure, the CR2_up 54a is in parallel with the D2_up 62a. Upper and lower switches are illustrated in each phase structure. Functional operation of the circuit prior to Mode 1, t<to, is shown in FIG. 10.

At t_o, as shown in FIG. 11, the slup 34 is turned on for Mode 1 with the following initial condition: $\begin{array}{c}{i}_{\mathrm{lr}}\left(0\right)=I_{\mathrm{ro}}\mathrm{and}{V}_{\mathrm{Cr1up}}\left(0\right)=0\\ i_{\mathrm{lr}}\left(t\right)=\frac{\mathrm{Vin}}{L}t+I_{\mathrm{ro}}\end{array}$
At t₂, the inductor current reaching the output current (Mode 2), which is reflected in the functional drawing of FIG. 12A. This condition can be explained by the formula shown in FIG. 12B. At Mode 3, both switches 34, 36 are ON, as best shown in the functional circuit diagram of FIG. 13A, with the initial condition explained by the formula shown in FIG. 13.

A state plane diagram is shown in FIG. 14 and shows the various centers of operation for Mode 2 and Mode 3.

The present invention allows zero voltage switching. Referring again to FIG. 9, an example of a two-stage circuit for zero voltage switching is illustrated and the function in the circuit can be expressed as: $\begin{array}{c}{I}_{\mathrm{o1}}=I_{\mathrm{o2}}=\frac{\mathrm{Io}}{2}\\ \mathrm{VD1\_low}=\mathrm{VD2\_low}=0\\ \mathrm{Cr1\_up}=\mathrm{Cr2\_up}=\mathrm{Cr}\end{array}$

When the mode of operation is t<t_o=0, the mode of operation can be expressed as a circuit function in FIG. 15. The voltage across the upper switch of phase 1 (S1_up) is zero. Switch S1_up OFF, S2_up is OFF, S1_low is On, S2_Low is On and D1_low and D2_low are OFF. Mode 1 of operation O<t<t is represented as shown in FIG. 16. This circuit function can be expressed as S1_up On, S1_low off, S2_up Off, S2_low On, D1_low On, and D2_Low Off. For purposes of this example: $\begin{array}{ccccc}Z=\sqrt{\frac{L_r}{C_r}}& & & & {\omega }_o=\frac{1}{\sqrt{L_r{C}_r}}\\ i_r\left(0\right)=I_{\mathrm{r1}}& & & & \end{array}$

The normalized value of V is $\frac{V}{V_{\mathrm{in}}}$
for I(t) is $\frac{i\left(t\right)Z}{V_{\mathrm{in}}}$
This results in: $\begin{array}{ccccc}{i}_r\left(t\right)=\frac{V_{\mathrm{in}}t}{L_r}+I_{\mathrm{r1}}& & & & i_{\mathrm{lrn}}\left(t\right)={\omega }_{o}t+I_{\mathrm{r1n}}\end{array}$

Mode 2 of operation, t₁<t<t₂ is shown by way of example to the circuit of FIG. 17. The circuit function is operative, S1_up On, S1_low off, S2_up off, S2_low On, D1_low Off, and D2_Low Off. At an initial condition: $\begin{array}{cc}{i}_r\left(t_1\right)=\frac{\mathrm{Io}}{2}& V_{\mathrm{Cr}}\left(\mathrm{t1}\right)=0\end{array}$
At a normalized solution: $\begin{array}{ccccc}{V}_{\mathrm{in}}-L_r\frac{\partial i_r\left(t\right)}{\partial t}-V_{\mathrm{Cr}}\left(t\right)=0& & & & V_{\mathrm{crn}}\left(t\right)=1-\cos \left({\omega }_{o}t\right)\\ i_r\left(t\right)-C_r\frac{\partial V_{\mathrm{Cr}}\left(t\right)}{\partial t}-\frac{I_o}{2}=0& & & & i_{\mathrm{rn}}\left(t\right)=\sin \left({\omega }_{o}t\right)+\frac{I_{\mathrm{on}}}{2}\end{array}$

Mode 3 of operation corresponds to t₂<t<t₃ as is shown in FIG. 18 where a representative circuit diagram is illustrated. The circuit function is operative for: S1_up OFF, S1_low OFF, S2_up OFF, S2_low On, D1_Low Off, and D2_Low Off. The initial condition is:
i_r(t₂)=I_r2 V_Cr(t2)=Vm
The normalized solution is: $\begin{array}{c}{V}_{\mathrm{in}}-L_r\frac{\partial i_r\left(t\right)}{\partial t}-V_{\mathrm{Cr}}\left(t\right)=0\\ i_r\left(t\right)-C_r\frac{\partial V_{\mathrm{Cr}}\left(t\right)}{\partial t}-\frac{I_o}{2}=0\\ V_{\mathrm{crn}}\left(t\right)=1+\sin \left(\frac{{\omega }_o}{\sqrt{2}}t\right)\frac{I_{\mathrm{r2n}}}{\sqrt{2}}+\cos \left(\frac{{\omega }_o}{\sqrt{2}}t\right)\left(\mathrm{Vmn}-1\right)\\ i_{\mathrm{rn}}\left(t\right)=\sqrt{2}\sin \left(\frac{{\omega }_o}{\sqrt{2}}t\right)\left(1-\mathrm{Vmn}\right)+\cos \left(\frac{{\omega }_o}{\sqrt{2}}t\right)I_{\mathrm{r2n}}\end{array}$

The state plane diagram for this type of function is shown in FIG. 14, which shows center of operation for Mode 2 and Mode 3. Points are shown for center of operation of Mode 2, the graph for Mode 2 during time ON, the center of operation for Mode 3, and the graph when tie is OFF, and a point for Mode 1 during inductor charge. Analytical solutions are shown below: $I_{r1n}=\frac{\sqrt{2}\left(\cos \left(\mathrm{wo}\mathrm{ton}\right)-\cos \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)\right)}{\sin \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)}$ $I_{\mathrm{rn}2}=\frac{\left(\cos \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)\cos \left(\mathrm{wo}\mathrm{ton}\right)-1\right)\sqrt{2}}{\sin \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)}$ $V_{\mathrm{mn}}=1-\cos \left(\mathrm{wo}\mathrm{ton}\right)$ $I_{\mathrm{on}}=2\frac{\begin{array}{c}-\sin \left(\mathrm{wo}\mathrm{ton}\right)\sin \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)-\\ \sqrt{2}+\sqrt{2}\cos \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)\cos \left(\mathrm{wo}\mathrm{ton}\right)\end{array}}{\sin \left(\frac{1}{2}\mathrm{wo}\sqrt{2}\mathrm{toff}\right)}$ $\mathrm{Ts}=2\mathrm{ton}+2\mathrm{toff}+\frac{I_{\mathrm{on}}-I_{r1n}}{{\omega }_o}$

Simplified equations for the circuit functions can be expressed as using: $\theta ={\omega }_o\mathrm{toff}\mathrm{and}\beta ={\omega }_o\mathrm{ton}$ $I_{r1n}=\frac{\sqrt{2}\left(\cos \left(\beta \right)-\cos \left(\frac{\theta }{\sqrt{2}}\right)\right)}{\sin \left(\frac{\theta }{\sqrt{2}}\right)}$ $I_{r2n}=\frac{\sqrt{2}\left(\cos \left(\frac{\theta }{\sqrt{2}}\right)\cos \left(\beta \right)-1\right)}{\sin \left(\frac{\theta }{\sqrt{2}}\right)}$ $I_{\mathrm{on}}=2\frac{\sqrt{2}\left(\cos \left(\frac{\theta }{\sqrt{2}}\right)\cos \left(\beta \right)-1\right)}{\sin \left(\frac{\theta }{\sqrt{2}}\right)}-2\sin \left(\beta \right)$ $V_{\mathrm{mn}}=1-\cos \left(\beta \right)$ $\mathrm{Ts}{\omega }_o=2\beta +2\theta +I_{\mathrm{on}}-I_{r1n}$

The circuit functions with conservation of energy are expressed as: $P_{\mathrm{in\_n}}=\frac{2}{\mathrm{Ts}}\left({\int }_0^{\mathrm{to}}{i}_{\mathrm{rn}}\left(t\right)\partial t+{\int }_{\mathrm{to}}^{t1}{i}_{\mathrm{rn}}\left(t\right)\partial t+{\int }_{t1}^{\mathrm{Ts}}{i}_{\mathrm{rn}}\left(t\right)\partial t\right)$ $P_{\mathrm{Out\_n}}=I_{\mathrm{on}}D$
Where D is: $\frac{V_o}{V_{in}}\iff \mathrm{Duty}\mathrm{Cycle}$
Using the conservation of energy, it is possible to obtain D as a function of β and θ:
D=f(β, θ)

A graphical example of this conservation of energy is shown in the graph of FIG. 19, where L_r=10 nH, C_r=10 nF, n=2, toff=90 nS.

Generalized solutions for the duty, time ON and number of phases (n) are shown in the three-dimensional graph of FIG. 20 showing duty on the vertical axis (y) and the number of phases and time ON on the lower axis x and Z. The spice result is shown in the graph of FIG. 21 showing time on the horizontal axis and the voltage on the vertical axis. The V-switch, gate drive, and next phase gate drive are shown.

A spice model set-up circuit is shown in FIG. 22. The spice model set-up shows various integrated circuits as U51 and U50 operative with various components and IC's. The spice model, of course, is a computerized modeling technique for the design of integrated circuits. By entering details of the circuit using the spice model as illustrated, it is possible to check for frequency and phase response of the circuit and check the circuit response over a set period of time as a transient analysis as compared to an AC analysis. There are also different analyses to check effects of temperature variations and noise. By using the spice model as shown in FIG. 22, the design was tested “on paper” and then prototyped.

A graph showing a state plane full load is shown in FIG. 23 showing Modes 1, 2 and 3. A no load state diagram is shown in FIG. 24. FIG. 25 shows an efficiency comparison between hard and soft switching.

Many modifications and other embodiments of the invention will come to the mind of one skilled in the art having the benefit of the teachings presented in the foregoing descriptions and the associated drawings. Therefore, it is understood that the invention is not to be limited to the specific embodiments disclosed, and that modifications and embodiments are intended to be included within the scope of the appended claims.

Claims

1. A multiphase DC-to-DC converter comprising:

at least two phase circuits, each having upper and lower power switches; and

a front-end inductor operative for forming a resonant tank circuit with said phase circuits to ensure zero voltage switching and minimizing power losses.

2. A multiphase DC-to-DC converter according to claim 1 and further comprising a controller operative with a phase circuit for detecting a zero volt crossing.

3. A multiphase DC-to-DC converter according to claim 2 wherein said controller comprises a PWM controller.

4. A multiphase DC-to-DC converter according to claim 1 wherein said resonant tank circuit is created to achieve zero voltage across said power switches.

5. A multiphase DC-to-DC converter according to claim 1 wherein said upper and lower power switches comprise field effect transistors.

6. A multiphase DC-to-DC converter according to claim 1 and further comprising a feedback signal processing circuit operative with each phase circuit.

7. A multiphase DC-to-DC converter according to claim 1 and further comprising an output capacitor operative with a voltage output from said phase circuits.

8. A multiphase DC-to-DC converter according to claim 1, and further comprising an output inductor operative within each phase circuit.

9. A multiphase DC-to-DC converter comprising:

at least two phase circuits, each having upper and lower power switches and a capacitor operative with at least each upper power switch; and

a front-end inductor operative for forming a resonant tank circuit with said upper power switches and capacitors to ensure zero voltage switching and minimizing power losses.

10. A multiphase DC-to-DC converter according to claim 9 wherein an upper capacitor is connected in parallel to a respective upper power switch.

11. A multiphase DC-to-DC converter according to claim 9 and further comprising a diode operative with each upper power switch to enhance zero voltage switching.

12. A multiphase DC-to-DC converter according to claim 9 and further comprising a diode operative with each lower power switch to enhance zero voltage switches.

13. A multiphase DC-to-DC converter according to claim 9 and further comprising a lower capacitor operative with each lower power switch.

14. A multiphase DC-to-DC converter according to claim 9 and further comprising a controller operative with a phase circuit for detecting a zero volt crossing.

15. A multiphase DC-to-DC converter according to claim 14 wherein said controller comprises a PWM controller.

16. A multiphase DC-to-DC converter according to claim 9 wherein said resonant tank circuit is created that achieves zero voltage across said power switches.

17. A multiphase DC-to-DC converter according to claim 9 wherein said upper and lower power switches comprise field effect transistors.

18. A multiphase DC-to-DC converter according to claim 9 and further comprising a feedback signal processing circuit operative with each phase circuit.

19. A multiphase DC-to-DC converter according to claim 9 and further comprising an output capacitor operative with a voltage output from said phase circuits.

20. A multiphase DC-to-DC converter according to claim 9, and further comprising an output inductor operative with phase circuits.

21. A method for regulating a multiphase DC-to-DC converter comprising:

forming a resonant tank circuit with phase circuits using a front end inductor; and

switching power switches in the phase circuits at zero volts to ensure zero voltage switching and minimizing power losses.

22. A method according to claim 21 wherein the method further comprises detecting a zero volt crossing.

23. A method according to claim 21 wherein the method further comprises forming a resonant tank circuit with capacitors operative with said power switches.

24. A method according to claim 21 wherein the method further comprises forming a resonant tank circuit with diodes operative with said power switches.