Direct-conversion transmitting circuit and integrated transmitting/receiving circuit
A transmitter is provided which includes a transmitting circuit that does not require a high-performance low noise VCO restricting cost reduction thereof and that can reduce the number of parts without requiring an RF filter. A direct conversion that does not require a transmission VCO is applied to the transmitting circuit. In order to achieve noise reduction in a receiving band, low-pass filters are provided at IQ input sections of a modulator that converts IQ signals into RF signals. In comparison with a conventional transmitter using offset PLL, an external VCO required in addition to an RF integrated circuit, a power amplifier, and a front end circuit is reduced. Even in current transistor performance, by using a filter having rapid waveform characteristics such as a SAW more inexpensive than the VCO, or the like, it is possible to provide a GSM/GSM 1800/GSM1900 triple band transmitter.
This is a continuation application of U.S. application Ser. No. 10/073,029, filed Feb. 12, 2002, the entire disclosure of which is hereby incorporated by reference.
BACKGROUND OF THE INVENTIONThe present invention relates to mobile communication equipment, and particularly to a direct-conversion transmitting circuit suitable for large scale integration and to an integrated transmitting/receiving circuit using the same.
According to rapid spread of the mobile communication equipment, requests for miniaturization and lower cost thereof have increased. Because of this, it is expected to apply a voltage control type oscillator (VCO), or an integrated circuit whose filter number is reduced and whose integration is enhanced. What is given as one conventional example of transmitting equipment is “RF Circuits Technique of Dual-Band Transceiver IC for GSM and DCS1800 applications” published in pages 278 to 281 of manuscripts for IEEE 25th European Solid-State Circuits Conference on 1999 by K. Takikawa et al.
As an important item on a transmitting circuit design, reduction of noise leakage into receiving frequency band has been given.
The PLL section 1814 is characterized by including a mixer 1801, and converts a frequency of an output signal of the VCO 1800 operated by an RF frequency, into an IF frequency fIF (270 MHz), by means of a mixer 1801, and outputs amounts of error between the IF signal and the output signal of the VCO 1800 by means of the phase comparator 1802. A frequency of the output error signal is lowered up to a baseband signal band that is the same as the I and Q input signals. High frequency noise of the error signal is suppressed by the low-pass filter 1803. A cutoff frequency of a PLL closed loop of a filter, the PLL closed loop which is denoted by reference numeral 1816, is about 1.6 MHz in a signal band of 200 kHz, and a noise of 20 MHz is greatly suppressed. Because of this, the noise generated in band which is a 20 MHz distant from an output signal of the VCO 1800 is greatly suppressed. Therefore, even if an output of the VCO 1800 is directly connected to a power amplifier PA, it is possible to suppress noises of receiving band up to −79 dBm/100 kHz (−129 dBm/Hz) or less without newly connecting a filter to an RF signal.
In a transmitter using the offset PLL, although a portion 1817 enclosed in solid line shown in
As described previously, a conventional example applying an offset PLL has been used as a transmitter because requiring no external filter. However, in the transmitter applying the offset PLL, there has been the limit of cost reduction because an external VCO with low noise is required.
An object of the present invention is to provide a direct-conversion transmitting circuit that does not require a low noise VCO having high performance and restricting the cost reduction in order to achieve reduction of the number of parts thereof, and that does not require an expensive and external high frequency filter such as a surface acoustic wave (SAW) filter or the like.
An object of the present invention is also to provide a transmitter/receiver using a direct-conversion transmitting circuit.
In order to solve problems described above, a transmitting circuit according to the present invention has an element of a circuit that uses a direct-conversion requiring no transmission VCO, and that provides a low-pass filter with each of I and Q (hereinafter, referred to as IQ) input portions of a modulator for converting IQ signals to a RF signal in order to achieve noise reduction in receiving band. An integrated transmitting/receiving circuit according to the present invention uses this direct-conversion transmitting circuit in a transmitting circuit section thereof. Concrete descriptions will be made in the following embodiments.
BRIEF DESCRIPTION OF THE DRAWINGS
Preferred embodiments of a direct-conversion transmitting circuit and an integrated transmitting/receiving circuit using the same according to the present invention will be described in detail below, with reference to the accompanying drawings.
First Embodiment A first embodiment of the present invention will be described with reference to FIGS. 1 to 8 and
Here, problems, which arise in a direct-conversion transmitting circuit and should be solved, will be clarified with reference to
A direct-up-conversion transmitter has the same circuit constitution as the conventional IF section 1815 shown in
In
In order to suppress the turn noise, low-pass filters 404 and 405 are installed at each output section of the AD converters. An output of the low-pass filter 404 is shown in a rectangle denoted by reference number 414, and suppresses a signal equal to or more than a cutoff frequency fCUTOFF of the low-pass filter 404, and includes a signal 415 and a noise 416 which is equal to or less than the cutoff frequency. The low-pass filter 405 also has the same output as the low-pass filter 404. Output signals of the low-pass filters 404 and 405 are applied to an I input 108 and a Q input 109 of the transmitting circuit, respectively. An optimal output signal amplitude of the AD converter is about 2 Vpp in normal differential. On the other hand, an optimal input level of the mixer depends on a circuit constitution and, for example, is 0.8 Vpp, and so is different from the output signal level of the AD converter. Besides, since optimal bias levels thereof are also different from each other, attenuators 103 and 104 each including a shift function of a bias level are required.
Since generating noises, an output of the attenuator 103 includes a signal 419, noise 420 equal to or less than the cutout frequency of the low-pass filter, and further a noise 421 generated by the attenuator as shown in a rectangle 418. The attenuator 104 also has the same output as the attenuator 103.
Each noise of the attenuators 103 and 104 extends within a wide band. In the mixers 101 and 102, the IQ signals containing noises are converted into RF frequencies that each regard a carrier frequency “fc” as a center. Local signals having the same frequency as the carrier frequencies whose a phase difference is 90° and which have the same frequencies are applied to two mixers 101 and 102. Each local signal is generated by a phase shifter 100 in accordance with an output of an oscillator 105. As typical phase shifters, there are two types of one using a CR filter and one using a frequency divider.. However, in the case of one using the frequency divider, an oscillation frequency of the oscillator 105 is twice as high as the carrier frequency.
An output of the modulator 106 composed of the mixers 101 and 102 becomes a signal modulated in both sides by regarding the carrier frequency “fc” as a center, as indicated in rectangle 422. This modulation signal includes a modulator output signal 424, and a thermal noise 425 caused by the AD converter for a modulator output, and a noise 426 caused by the attenuator for a modulator output. In particular, the noise 421 of the attenuator extends within a wide band, and the modulated noise also exists as the noise 426 caused by the attenuator in a wide band.
A modulation signal is further amplified by a driver amplifier 107, and is output via an output terminal 132. This signal includes a wide band noise therein. Thus, in order to reduce a noise within a receiving band which is 20 MHz distant from a transmission band, a high frequency (RF) filter 430 which has a rapid waveform characteristic and which applies a SAW (surface acoustic wave) device, dielectric resonator, or the like is required.
In order to eliminate the high frequency filter, it is required to reduce a wide band noise each generated by the attenuators 103 and 104. Due to this, as shown in
In the attenuators 103 and 104, similarly to the example shown in
Specification of the case of the GSM is shown in
In order to make performance checks on a direct-up-conversion transmitting circuit adopted in the present embodiment, the circuit shown in
In order to investigate effects of a low-pass filter in a direct-up-conversion transmitting circuit proposed in the present embodiment, evaluation oriented to a GSM1800 has been carried out about the case of no use of a filter and the case where the filter has cutoff frequencies of 4.9 MHz and 440 kHz.
When the IQ input level is −17 dBV, an output thereof is −7 dBm. In comparison with respective characteristics under this case, the case of no use of filter has a level of −142 dBc/Hz (−149 dBm/Hz) while the case of use of a filter having a cutoff frequency of 440 kHz has a level of −156 dBc/Hz (−163 dBm/Hz). Therefore, it is understood that the case of use of a filter having a cutoff frequency of 440 kHz is improved by 14 dB. The similar results have been obtained even in the case of GSM transmission frequency band. In this trial testing, although performances relative to the GSM specification are insufficient, it is considered that improvement of device characteristics by use of a SiGe (silicon/germanium) bipolar transistor, and the like, can be achieved, and thereby effects of the present invention are expected.
Although it is possible to reduce the noise level in receiving band by lowering the cutoff frequency of the filter, degradation of phase precision of a modulation signal is considered due to an affect of frequency characteristics of a group delay.
As described above, according to the constitution of the first embodiment that is the present invention in which a low-pass filter is connected immediately before the IQ inputs of each mixer circuit constituting a direct-up-conversion transmitting circuit, it is understood that the spur level in receiving band for an output can be eminently improved.
Second Embodiment A second embodiment of the present invention will be described with reference to
The Sallen-Key filter can be composed of a Butterworth type filter or a Chabyshev type filter by selecting an element value. The filter output drives a mixer input stage transistor 1002, and is converted into a high frequency signal by means of two groups of differential pairs 1006, and is fetched from respective connection ends between load resistances 1007 and the differential pairs 1006 of a mixer. Here, although the second order filter is shown, the third order filter can easily be used instead of the second order filter. However, as far as the GSM system is concerned, as is evident from
A third embodiment of the present invention will be described with reference to
First, a mixer carrier leak will be described here. This mixer functions as a multiplier. As shown in formula (1), a modulation wave fc(t) is generated by multiplying a baseband input signal f(t) and a local signal cos(2πfc).
fc(t)=f(t)×cos(2πfc) (1)
When a DC offset a is added to the mixer input, as shown in formula (2), a single term of a carrier signal is generated and this causes degradation of modulation precision.
fc(t)=f(t)×cos(2πfc)+αcos(2πfc) (2)
In order to correct the DC offset, in the present embodiment respective channels of the I and Q are provided with a bias correction circuit 1103 consisting of an AD converter ADC for detecting an offset, a DA converter DAC for generating a correction bias, and a control section CNT which carries out control for minimizing an offset and simultaneously stores correction conditions. Correction is carried out within time from supply of power to beginning of transmission. The control section CNT is composed of a control register and a logic circuit and the like. By the present embodiment, a direct-up-conversion transmitter reducing an effect of a DC offset can be achieved.
Fourth Embodiment A fourth embodiment according to the present invention will be described with reference to
The AD converter ADC is selectively connected to the I and Q signal lines by means of a switch SW. The DA converter DAC is provided exclusively for each of the I and Q signal lines. The control section CNT is also provided exclusively for each of the I and Q signal lines, and thereby each of the DC offsets is independently controlled. Since correction cannot be made for the I and Q simultaneously, correction time of the present embodiment is required about twice further than that of the third embodiment. By the present embodiment, a direct-up-conversion transmitter reducing an effect of the DC offset can be achieved with a small circuit scale.
Fifth Embodiment A fifth embodiment according to the present invention will be described with reference to
The I input signal 108 is converted to current from voltage at a differential input circuit composed of a PNP type transistor 1303. The transistors 1301 and 1302 each have a current mirror structure, and current is returned to two groups of differential pairs 1006 for mixer, and a mixer output is supplied from a connection end connected to each load resistor 1007. A low-pass filter 1300 is composed of resistors R1 and R2 connected in series to bases of the transistors 1301 and 1302, and capacitors C1 and C2 connected to a grounding terminal, and an emitter of the transistor 1302, respectively. In addition, emitters of the transistors 1301 and 1302 are grounded via resistors R3 and R4, respectively. By the present embodiment, it is possible to achieve a direct-conversion transmitter capable of corresponding to a large baseband input signal.
Sixth Embodiment A sixth embodiment according to the present invention will be described with reference to
In contrast, in the present embodiment, MOSFETs 1400 and 1401 each having a gate in which no DC current flows are applied to a current mirror section, and thereby generation of a DC offset caused by deviation of potential fall at the resistors R1 and R2 is suppressed. In addition, a transistor constituting the attenuator 103 is changed from the PNP transistor 1303 to a P type MOSFET 1402. This is because input impedances thereof are increased and driving thereof can be achieved by using small amount of power.
By the present embodiment, since a large resistor can be applied to a filter, capacitive value thereof can be reduced. As a result, a low noise direct-conversion transmitter having small element area can be achieved.
Seventh Embodiment A seventh embodiment according to the present invention will be described with reference to
A transmitting circuit applies any of the embodiments introduced previously. The IQ transmission signals are adjusted to a desired signal level at the attenuator 103, and a wide band noise generated by the attenuator 103 is suppressed at the low-pass filter 130. The signals whose noises are suppressed at the filters 130 are converted into modulation signals having RF frequency, by the modulator 106 composed of a group of mixers 101 and 102. These mixers operate within respective ranges from 880 to 915 MHz in the case of GSM, from 1710 to 1785 MHz in the case of the GSM1800, and from 1850 to 1910 MHz in the case of the GSM1900. The mixers 101 and 102 each are driven by means of a local signal having a 90° phase difference generated by the frequency divider 100.
The output signal of the modulator 106 is amplified by a GSM driver amplification circuit 1500 or a driver circuit 1501 compatible with the GSM1800 and GSM1900. A band pass filter 1502 such as a SAW filter having rapid waveform characteristics, or the like is connected to an output of the GSM driver circuit, and thereby residual noise in receiving band which is 20 MHz distant is eliminated. Here, although the SAW filter having rapid waveform characteristics is connected to the GSM output in accordance with the testing results shown in the first embodiment, this filter can replace an inexpensive LC filter according to higher performance of the circuit. The output signal of the filter is amplified by a power amplifier module (PA module) 801.
A simple LC filter 1503 is connected to an output of the driver amplification circuit 1501 compatible with the GSM1800 and GSM1900, and the signal thereof is amplified by the power amplifier module 801 after high harmonics are eliminated. Here, the power amplifier module 801 packages the GSM modulator and the modulator compatible with the GSM1800 and GSM1900. The amplified signal is transmitted from an antenna via a low-pass filter (LPF) 1504 that eliminates high harmonics generated by an output of the amplifier in the power amplifier module 801, and via a transmission/reception changeover switch (S/W) 1505.
A voltage control oscillator (RF VCO) 1515 receives and constantly oscillates a feedback loop by means of a synthesizer (RF PLL Synth) 1516, and generates a transmission/reception signal as follows.
GSM reception: An oscillator 1515 oscillates within a range from 3700 to 3840 MHz. The output of this oscillator is frequency-divided into two sections by means of a frequency divider 1517, and further is frequency-divided into two sections by the frequency divider 1512. Thereby, a local signal for a GSM reception, which drives the mixers 1510 and 1511, is obtained.
GSM1800 reception: The oscillator 1515 oscillates within a range from 3610 to 3760 MHz. The output of this oscillator is directly connected to the frequency divider 1512 without passing through the frequency divider 1517, and is frequency-divided into two sections by a switch 1518. Thereby, the local signal for the GSM1800 reception, which drives the mixers 1510 and 1511, is obtained.
GSM1900 reception: The oscillator 1515 oscillates within a range from 3860 to 3980 MHz. The output of this oscillator is directly connected to the frequency divider 1512 without passing through the frequency divider 1517, and is frequency-divided into two section by the switch 1518. Thereby, the local signal for the GSM1900 reception, which drives the mixers 1510 and 1511, is obtained.
GSM transmission: The oscillator 1515 oscillates within a range from 3520 to 3660 MHz. The output of this oscillator is frequency-divided into two sections by means of a frequency divider 1519, and further is frequency-divided by means of the frequency divider 100. Thereby, a local signal for the GSM transmission, which drives the modulator 106, is obtained.
GSM1800 transmission: The oscillator 1515 oscillates within a range from 3420 to 3570 MHz. The output of this oscillator is directly connected to the frequency divider 100 without passing through the frequency divider 1519, and is frequency-divided into two sections by means of a switch 1520 without passing through a frequency divider 1519. Thereby, a local signal for the GSM1800 transmission, which drives the modulator 106, is obtained.
GSM1900 transmission: The oscillator 1515 oscillates within a range from 3700 to 3820 MHz. The output of this oscillator is directly connected to the frequency divider 100 without passing through the frequency divider 1519, and is frequency-divided into two sections by the switch 1520. Thereby, a local signal for the GSM1900 transmission, which drives the modulator 106, is obtained.
In order to make such operations, the oscillator 1515 operates within a range from 3420 to 3980 MHz. By the present embodiment, a direct-conversion circuit can be achieved for both of transmission and reception by using one voltage control oscillator.
Eighth Embodiment An eighth embodiment according to the present invention will be described with reference to
When the output of the oscillator 1515 is inputted into the two-frequency divider, two waveforms of outputs 1 and 2 are generated. One of two rising edges 2007 and 2008 in the waveforms of the outputs 1 and 2 is synchronized with a rising edges 2005 of the input of the frequency divider (that is, the output of the oscillator), and the other is synchronized with a falling edge 2006 of the input of the frequency divider.
If the output of the oscillator 1515 has a duty ratio of 50%, a phase difference between these two outputs is 90°. In the case where the duty ratio is shifted from 50%, an error occurs in the phase difference. When the waveform of the output 1 is further frequency-divided into two sections, waveforms of outputs 3 and 4 are obtained. Rising edges 2009 and 2010 of any of the waveforms are also synchronized with the rising edge 2005 of the oscillator output, and a signal precisely having a phase difference of 90° phase difference can be generated without depending on the duty ratio of the oscillation waveforms.
Therefore, although a signal having a precise phase difference can be generated relative to the GSM in the sixth embodiment described above, there occurs each error depending on the duty ratio of the oscillation waveforms in the GSM1800 and the GSM1900. The transceiver IC 160 employing a direct-conversion for both of transmission and reception, which is applied to the triple band of the GSM/GSM1800/GSM1900 in the present embodiment shown in
A ninth embodiment according to the present invention will be described with reference to
By the present embodiment described above, a 4-band compatible transceiver IC can be achieved by using the small number of external elements.
As described above, several preferred embodiments of the present invention has been described. However, the present invention is not limited to these embodiments, and of course various design modifications thereof can be made without departing from the spirit of the present invention.
According to the present invention, in comparison with a conventional transmitter applying offset PLL, even if a required external VCO in addition to an RF integrated circuit, a power amplifier and a front end circuit is reduced and current transistor performance is maintained, then a GSM/GSM1800/GSM 1900 triple band transmitter/receiver can be achieved by using one filter having rapid waveform characteristics, such as a SAW or the like more inexpensive than the VCO. Further, by improving transistor characteristics, a triple band or quadrant band transmitter/receiver can be formed without using expensive external parts.
Claims
1. A direct-conversion transmitting circuit comprising:
- a modulator to modulate I and Q signals into a transmitting frequency signal, the I and Q signals being inputted from a base band circuit to said modulator;
- a first driver amplification circuit coupled to an output node of said modulator to amplify a first transmitting frequency signal being modulated into a first frequency band through said modulator;
- a second driver amplification circuit coupled to an output node of said modulator to amplify a second transmitting frequency signal being modulated into a second frequency band through said modulator, the second frequency band being higher than the first frequency band;
- first and second low-pass filters being coupled at output nodes thereof to input nodes of said modulator;
- first and second gain/bias adjusters being coupled at output nodes thereof to input nodes of said first and second low-pass filters, respectively; and
- wherein said modulator comprises first and second mixers, and a first phase shifter,
- wherein high frequency output terminals of said first and second mixers are connected to each other,
- wherein an output terminal of said first low-pass filter is connected to an input terminal of said first mixer, and an input terminal of said first low-pass filter is connected to an output terminal of said first gain/bias adjuster to suppress a noise generated by said first gain/bias adjuster,
- wherein an output terminal of said second low-pass filter is connected to an input terminal of said second mixer, and an input terminal of said second low-pass filter is connected to an output terminal of said second gain/bias adjuster to suppress a noise generated by said second gain/bias adjuster,
- wherein a first output terminal of said first phase shifter is connected to a local signal input terminal of said first mixer, and a second output terminal of said first phase shifter is connected to a local signal input terminal of said second mixer,
- wherein an input signal generated from an output signal of a first AD converter is applied to an input terminal of said first gain/bias adjuster to reduce difference in gain and bias levels between an input signal of said first mixer and an output signal of said first AD converter, and
- wherein an input signal generated from an output signal of a second AD converter is applied to an input terminal of said second gain/bias adjuster to reduce difference in gain and bias levels between an input signal of said second mixer and an output signal of said second AD converter.
2. The direct-conversion transmitting circuit according to claim 1, wherein said first phase shifter is comprised of a frequency divider circuit.
3. The direct-conversion transmitting circuit according to claim 1, wherein each circuit of said first and second low-pass filters is comprised of a filter whose order is at least a second order.
Type: Application
Filed: Sep 29, 2006
Publication Date: Jan 25, 2007
Inventors: Satoshi Tanaka (Kokubunji), Masamichi Tanabe (Tokyo), Yasuyuki Okuma (Kokubunji), Taizo Yamawaki (Tokyo), Koichi Yahagi (Takasaki), Hiroaki Matsui (Takasaki), Robert Henshaw (Cambridge)
Application Number: 11/529,326
International Classification: H04B 1/04 (20060101); H01Q 11/12 (20060101);