COULOMB COUNTING USING ANALOG-TO-FREQUENCY CONVERSION

Abstract

In an analog-to-frequency converting circuit, a set of switches receive a first sense signal indicative of a current and provides a second sense signal that alternates between an original version of the first sense signal and a reversed version of the first sense signal, under control of a switching signal. An integral comparing circuit integrates the second sense signal to generate an integral value and generates a train of trigger signals. Each trigger signal is generated when the integral value reaches a preset reference. A compensation circuit compensates for the integral value with a predetermined value in response to each trigger signal. A control circuit generates the switching signal such that a time interval during which the second sense signal is the original version and a time interval during which the second sense signal is the reversed version are substantially the same.

Description

BACKGROUND

A conventional method for counting charges passing in and out of a battery (referred to as “coulomb counting”) includes generating a sense voltage linearly proportional to a current (e.g., a charging current or a discharging current) of the battery, using a voltage-to-frequency converter to convert the sense voltage to a frequency signal linearly proportional to the sense voltage, and counting a number of waves/pulses of the frequency signal to generate a count value. The count value can represent an amount of accumulation of electric charges passing in and out of the battery.

FIG. 1 illustrates a circuit diagram of a conventional voltage-to-frequency converter 100. As shown in FIG. 1, the voltage-to-frequency converter 100 includes a sense resistor R′_SEN, an integrator (e.g., a combined circuit of a resistor R′_INT, an integrating capacitor C′_INT, and an operational amplifier (OPA) 112; hereinafter, integrator (R′_INT, C′_INT, 112)), comparators 114 and 116, compensation circuitry (e.g., a combined circuit of capacitors CP₁ and CP₁, and switches M₁-M₈), and a control circuit 150.

The sense resistor R′_SEN senses a battery current I′_BAT of a battery 152 to generate a sense voltage V′_SEN. The integrator (R′_INT, C′_INT, 112) integrates the sense voltage V′_SEN to generate an integral result V′_INT. The integral result V′_INT represents an integral value of the sense voltage V′_SEN, and therefore represents an integral value of the battery current I′_BAT. The comparators 114 and 116 compare the integral result V′_INT with voltage references V′_H and V′_L (V′_L<V′_H) to generate a train of pulse signals CMP′_H and CMP′_L. The control circuit 150 controls the switches M₁-M₈ according to the pulse signals CMP′_H and CMP′_L. A frequency f′_CMP of the pulse signals CMP′_H or CMP′_L represents the battery current I′_BAT.

By way of example, if the battery 152 is in a charging mode, then the sense voltage V′_SEN labeled in FIG. 1 has a positive value, and the integral result V′_INT decreases because of integration of the positive sense voltage V′_SEN. When the integral result V′_INT decreases to the voltage reference V′_L, the comparator 116 generates a pulse signal CMP′_L, and the control circuit 150 turns on a first group of switches M₁, M₄, M₆ and M₇, and turns off a second group of switches M₂, M₃, M₅ and M₈. Hence, the capacitor CP₁ provides an amount of compensation charges Q′_REF to the integrating capacitor C′_INT through the inverting input terminal 154 of the OPA 112 to increase the integral result V′_INT, and the capacitor CP₂ is charged by the voltage V′_REF to store an amount of compensation charges Q′_REF. The increased integral result V′_INT continues to decrease because of the integration of the positive sense voltage V′_SEN. When the integral result V′_INT decreases to the voltage reference V′_L, the comparator 116 generates a next pulse signal CMP′_L, and the control circuit 150 turns off the first group of switches and turns on the second group of switches. Hence, the capacitor CP₂ provides the stored compensation charges Q′_REF to the integrating capacitor C′_INT through the terminal 154 to increase the integral result V′_INT, and the capacitor CP₁ is charged by the voltage V′_REF to store an amount of compensation charges Q′_REF. Thus, if the battery 152 is in a charging mode, then the control circuit 150 alternately turns on the first group of switches and the second group of switches, and the comparator 116 generates a train of pulse signals CMP′_L at a frequency f′_CMP that is linearly proportional to the charging current I′_BAT. Similarly, if the battery 152 is in a discharging mode, then the control circuit 150 alternately turns a third group of switches M₁, M₂, M₇ and M₈ and a fourth group of switches M₃, M₄, M₅ and M₆, and the comparator 114 can generate a train of pulse signals CMP′_H at a frequency f′_CMP that is linearly proportional to the discharging current I′_BAT. The frequency f′_CMP of the trigger signals CMP′_H/CMP′_L can be used for the abovementioned coulomb counting.

However, the voltage-to-frequency converter 100 has some shortcomings. For example, the frequency f′_CMP may have error caused by an input voltage offset V′_OS of the OPA 112. Thus, a count value obtained by counting a number of the pulse signals CMP′_H/CMP′_L may have error caused by the input voltage offset V′_OS.

Additionally, because the OPA 112 controls its inverting input terminal 154 and its non-inverting input terminal 156 to have the same voltage level, and the inverting input terminal 154 receives either a voltage level of V′_REF or a voltage level of −V′_REF from the capacitors CP₁ and CP₁, a voltage level at the non-inverting input terminal 156, which is also a terminal 156 of the sense resistor R′_SEN, should be neither relatively high nor relatively low compared with the voltage levels V′_REF and −V′_REF. Because the voltage V′_REF is relatively small compared with a voltage level at a positive terminal of the battery 152, the voltage level at the terminal 156 of the sense resistor R′_SEN should be relatively small. Thus, the sense resistor R′_SEN can be placed at the negative terminal of the battery 152 and cannot be placed at the positive terminal of the battery 152. In some situations, it would be beneficial to place the sense resistor R′_SEN at the positive terminal of the battery 154. For example, there may be a thermistor (not shown) connected to the negative terminal of the battery 152 to measure temperature of the battery 152, and a sense resistor R′_SEN placed at the negative terminal of the battery 152 may cause error in the measurement of the temperature. Placing the sense resistor R′_SEN at the positive terminal of the battery 152 can avoid this error.

Moreover, the compensation circuitry in the voltage-to-frequency converter 100 uses two capacitors CP₁ and CP₁ to provide compensation charges to the integrator (R′_INT, C′_INT, 112). It would be beneficial to use one capacitor instead of two capacitors to provide compensation charges, so as to reduce the cost and size of the compensation circuitry.

Furthermore, when the capacitor CP₁ provides compensation charges to the integrator (R′_INT, C′_INT, 112), the capacitor CP₁ attempts to apply a voltage level V′_REF or −V′_REF to the inverting input terminal 154 of the OPA 112 that is different from a voltage level, e.g., zero volts, at the non-inverting input terminal 156 of the OPA 112. Because the OPA 112 controls the terminals 154 and 156 to have the same voltage level, a relatively big current may be generated by the OPA 112 to flow through the capacitors C′_INT and CP₁ to discharge the capacitor CP₁, so as to reduce the voltage level V′_REF or −V′_REF of the capacitor CP₁ to zero volts relatively quickly. This requires that the OPA 112 has a relatively high sensibility, and is able to generate and sustain a relatively big current. Such an OPA is relatively expensive and consumes relatively high power.

FIG. 2 illustrates a circuit diagram of a conventional coulomb counter 200. The coulomb counter 200 includes a sense resistor R′_SEN, a counter 262, and a voltage-to-frequency converter. The voltage-to-frequency converter includes switches S₁-S₄, an integrator (e.g., a combined circuit of a resistor R′_INT, a capacitor C′_INT, and an operational amplifier OPA 212; hereinafter, integrator (R′_INT, C′_INT, 212)), an RC filter (including a resistor R_FLT and a capacitor R_FLT), comparators 214 and 216, and a control logic & polarity detection module 260.

The sense resistor R′_SEN generates a sense voltage V′_SEN1 indicative of a battery current I′_BAT. The switches S₁-S₄ receive the sense voltage V′_SEN1 and provide a voltage V′_SEN2 to the integrator (R′_INT, C′_INT, 212). The integrator (R′_INT, C′_INT, 112) integrates the voltage V′_SEN2 to generate an integral result V′_INT. The integral result V′_INT ramps up and down alternately, under control of the switches S₁-S₄. The comparators 214 and 216 compare the integral result V′_INT with voltage references V′_H and V′_L (V′_L<V′_H) to generate trigger signals CMP′_H and CMP′_L, alternately. The module 260 controls the switches S₁-S₄ according to the trigger signals CMP′_H and CMP′_L such that the voltage V′_SEN2 alternates between a voltage level of V′_SEN1 and a voltage level of −V′_SEN1. A frequency f′_CMP at which the trigger signals CMP′_H and CMP′_L alternate represents the battery current I′_BAT. The counter 262 counts a number of the trigger signals CMP′_H and CMP′_L to generate a count value representing an amount of accumulation of electric charges in the battery current I′_BAT.

By way of example, in a situation when the battery current I′_BAT flows through the sense resistor R′_SEN from the terminal labeled “CS+” to the terminal labeled “CS−,” the sense voltage V′_SEN1 labeled in FIG. 2 has a positive value. The module 260 can turn on the switches S₁ and S₄ and turn off the switches S₂ and S₃, and therefore the integrator (R′_INT, C′_INT, 212) integrates the voltage level of V′_SEN1 to decrease the integral result V′_INT. When the integral result V′_INT decreases to the voltage reference V′_L, the comparator 216 generates a trigger signal CMP′_L, and the module 260 turns off the switches S₁ and S₄ and turns on the switches S₂ and S₃ in response to the trigger signal CMP′_L. Hence, the integrator (R′_INT, C′_INT, 212) integrates the voltage level of −V′_SEN1 to increase the integral result V′_INT. When the integral result V′_INT increases to the voltage reference V′_H, the comparator 214 generates a trigger signal CMP′_H, and the module 260 turns on the switches S₁ and S₄ and turns off the switches S₂ and S₃ in response to the trigger signal CMP′_H. Thus, by alternately turning on the pair of switches S₁ and S₄ and the pair of switches S₂ and S₃ according to the trigger signals CMP′_H and CMP′_L, the coulomb counter 200 alternately generates the trigger signals CMP′_H and CMP′_L at an alternation frequency f′_CMP that is linearly proportional to the battery current I′_BAT. Similarly, when the battery current I′_BAT flows through the sense resistor R′_SEN from the terminal labeled “CS−” to the terminal labeled “CS+,” the coulomb counter 200 can also alternately generate trigger signals CMP′_H and CMP′_L at an alternation frequency f′_CMP that is linearly proportional to the battery current I′_BAT. The alternation frequency f′_CMP can be used for coulomb counting.

However, the coulomb counter 200 has some shortcomings. For example, in a first time interval, the integrator (R′_INT, C′_INT, 212) can integrate a voltage level of V′_SEN1 so that the integral result V′_INT decreases from the voltage reference V′_H to the voltage reference V′_L; and in a second time interval, the integrator (R′_INT, C′_INT, 212) can integrate a voltage level of −V′_SEN1 so that the integral result V′_INT increases from the voltage reference V′_L to the voltage reference V′_H. The OPA 212 may have an input voltage offset V′_OS, which causes a time difference between the first and second time intervals. Consequently, the alternation frequency f′_CMP of the trigger signals CMP′_H and CMP′_L may have error caused by an integration of the voltage offset V′_OS based on the time difference. A count value obtained by counting a number of the trigger signals CMP′_H/CMP′_L may have error caused by the input voltage offset V′_OS.

Additionally, due to non-ideality of the comparators 214 and 216, there may be time delays in generation of the trigger signals CMP′_H and CMP′_L. This may result in error, e.g., a decrement, in the alternation frequency f′_CMP of the trigger signals CMP′_H and CMP′_L. Comparators with relatively fast response speed may be used to reduce the error in the alternation frequency f′_CMP. However, such comparators may be relatively expensive and consume relatively high power.

A coulomb counter that addresses the abovementioned shortcomings would be beneficial.

SUMMARY

In one embodiment, an analog-to-frequency converting circuit includes a set of switches, an integral comparing circuit, a compensation circuit, and a control circuit. The switches receive a first sense signal indicative of a current and provide a second sense signal that alternates between an original version of the first sense signal and a reversed version of the first sense signal, under control of a switching signal. The integral comparing circuit integrates the second sense signal to generate an integral value and generates a train of trigger signals at a frequency indicative of the current. Each trigger signal of the trigger signals is generated when the integral value reaches a preset reference. The compensation circuit compensates for the integral value with a predetermined value in response to each trigger signal of the trigger signals. The control circuit generates the switching signal such that a first time interval during which the second sense signal is the original version and a second time interval during which the second sense signal is the reversed version are substantially the same.

BRIEF DESCRIPTION OF THE DRAWINGS

Features and advantages of embodiments of the claimed subject matter will become apparent as the following detailed description proceeds, and upon reference to the drawings, wherein like numerals depict like parts, and in which:

FIG. 1 illustrates a circuit diagram of a conventional voltage-to-frequency converter.

FIG. 2 illustrates a circuit diagram of a conventional coulomb counter.

FIG. 3 illustrates a circuit diagram of an example of a coulomb counter, in an embodiment according to the present invention.

FIG. 4 illustrates a circuit diagram of an example of a coulomb counter, in an embodiment according to the present invention.

FIG. 5A illustrates examples of waveforms of signals associated with a coulomb counter, in an embodiment according to the present invention.

FIG. 5B illustrates examples of waveforms of signals associated with a coulomb counter, in an embodiment according to the present invention.

FIG. 6A illustrates a circuit diagram of an example of a control circuit, in an embodiment according to the present invention.

FIG. 6B illustrates a circuit diagram of an example of a control circuit, in an embodiment according to the present invention.

FIG. 7 illustrates examples of waveforms of signals associated with a coulomb counter, in an embodiment according to the present invention.

FIG. 8 illustrates a flowchart of examples of operations performed by a coulomb counter, in an embodiment according to the present invention.

DETAILED DESCRIPTION

Reference will now be made in detail to the embodiments of the present invention. While the invention will be described in conjunction with these embodiments, it will be understood that they are not intended to limit the invention to these embodiments. On the contrary, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the appended claims.

Furthermore, in the following detailed description of the present invention, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, it will be recognized by one of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well known methods, procedures, components, and circuits have not been described in detail as not to unnecessarily obscure aspects of the present invention.

Embodiments according to the present invention provide solutions to convert an analog signal, e.g., a current signal or a voltage signal, to a frequency signal indicative of, e.g., linearly proportional to, the analog signal. The frequency signal can be used for coulomb counting.

FIG. 3 illustrates a circuit diagram of an example of a coulomb counter 300, in an embodiment according to the present invention. In one embodiment, the coulomb counter 300 includes an analog-to-frequency converting circuit, a logic circuit 304, and a counter 306. The analog-to-frequency converter can include a sense resistor R_SEN, a set of switches 302, an integral comparing circuit 310, a control circuit 320, and a compensation circuit 330.

In one embodiment, the analog-to-frequency converting circuit converts an analog signal (e.g., a current I_SEN flowing through the sense resistor R_SEN, a first sense voltage V_SEN1 across the sense resistor R_SEN, or a second sense voltage V_SEN2 input to the integral comparing circuit 310) to a frequency signal such as a train of trigger signals S_FREQ (e.g., a series of rising edges or a series of falling edges) at a frequency f_CMP indicative of (e.g., linearly proportional to) the analog signal. The counter 306 can count a number of the trigger signals S_FREQ (e.g., rising edges or falling edges) to generate a count value N_COUNT. The count value N_COUNT can represent a result of integrating the analog signal. In one embodiment, the analog signal includes a current I_SEN of a battery 344. In one such embodiment, the count value N_COUNT obtained by counting the trigger signals S_FREQ can represent an amount of accumulation of electric charges in the current I_SEN, e.g., an amount of accumulation of electric charges passing in and out of the battery 344.

More specifically, in the analog-to-frequency converting circuit in one embodiment, the sense resistor R_SEN senses a current I_SEN flowing through the sense resistor R_SEN, and generates a first sense signal V_SEN1, e.g., a voltage signal, indicative of the current I_SEN. The switches 302 receive the sense signal V_SEN1 and provide a second sense signal V_SEN2, e.g., a voltage signal, that alternates between an original version of the sense signal V_SEN1 (e.g., V_SEN2=V_SEN1) and a reversed version of the sense signal V_SEN1 (e.g., V_SEN2=−V_SEN1), under control of a switching signal F_CHOP. The integral comparing circuit 310 integrates the sense signal V_SEN2 to generate an integral value V_INT of the sense signal V_SEN2. The integral value V_INT may increase or decrease depending on whether the sense signal V_SEN2 is the original version of V_SEN1 or the reversed version of V_SEN1. The integral comparing circuit 310 also compares the integral value V_INT with preset references V_H and V_L (V_L<V_H) and generates a train of trigger signals CMP_H and CMP_L at a frequency f_CMP indicative of the current I_SEN. In one embodiment, the abovementioned trigger signals S_FREQ include the trigger signals CMP_H and CMP_L. Each trigger signal of the trigger signals CMP_H and CMP_L is generated when the integral value V_INT of the sense signal V_SEN2 reaches a preset reference V_H or V_L. In one embodiment, a trigger signal CMP_H represents that the integral value V_INT has increased to the preset reference V_H, and a trigger signal CMP_L represents that the integral value V_INT has decreased to the preset reference V_L. The compensation circuit 330 can compensate, e.g., decrease, the integral value V_INT with a predetermined value in response to each trigger signal CMP_H such that a train of trigger signals CMP_H are generated. Similarly, the compensation circuit 330 can compensate, e.g., increase, the integral value V_INT with a predetermined value in response to each trigger signal CMP_L such that a train of trigger signals CMP_L are generated. The control circuit 320 can generate the switching signal F_CHOP such that a first time interval T_A during which the sense signal V_SEN2 is in the original version of the sense signal V_SEN1 (e.g., V_SEN2=V_SEN1) and a second time interval T_B during which the sense signal V_SEN2 is in the reversed version of the sense signal V_SEN1 (e.g., V_SEN2=−V_SEN1) are substantially the same.

By way of example, the switches 302 include switches S_A1, S_A2, S_B1 and S_B2. The switching signal F_CHOP can turn off the switches S_B1 and S_B2 and turn on the switches S_A1 and S_A2 so that the sense signal V_SEN2 is in an original version of the sense signal V_SEN1, e.g., V_SEN2=V_SEN1. The switching signal F_CHOP can also turn off the switches S_A1 and S_A2 and turn on the switches S_B1 and S_B2 so that the sense signal V_SEN2 is in a reversed version of the sense signal V_SEN1, e.g., V_SEN2=−V_SEN1.

The integral comparing circuit 310 can include an integrator (e.g., a combined circuit of an integrating resistor R_INT, an integrating capacitor C_INT, and an operational amplifier (OPA) 312; hereinafter, integrator (R_INT, C_INT, 312)) and comparator circuitry (e.g., including comparators 314 and 316; hereinafter, comparator circuitry 314-316). The integrator (R_INT, C_INT, 312) integrates the sense signal V_SEN2 and generates an integral value V_INT of the sense signal V_SEN2. The integral value V_INT can be given by:

$V_{\mathrm{INT}}=\frac{1}{R_{\mathrm{INT}}C_{\mathrm{INT}}}\int \left(V_{\mathrm{OS}}-V_{\mathrm{SEN}2}\right)t+V_R^{\prime },$

where, R_INT represents the resistance of the integrating resistor R_INT, C_INT represents the capacitance of the integrating capacitor C_INT, V_OS represents an input offset of the OPA 312, V_SEN2 represents a voltage level of the sense signal V_SEN2, and V′_R represents a voltage level at the non-inverting input terminal of the OPA 312. Hence, the integral value V_INT decreases if the sense signal V_SEN2 is positive, and increases if the sense signal V_SEN2 is negative. The comparator circuitry 314-316 compares the integral value V_INT with a high-side reference V_H and a low-side reference V_L that is less than the high-side reference V_H, generates a trigger signal CMP_H if the integral value V_INT is greater than the high-side reference V_H, and generates a trigger signal CMP_L if the integral value V_INT is less than the low-side reference V_L. In the example of FIG. 3, the trigger signals CMP_H and CMP_L are logic-low signals (e.g., falling edges of signals). However, the invention is not so limited. In another embodiment, the comparator circuitry is in another structure, and the trigger signals CMP_H and CMP_L can be logic-high signals (e.g., rising edges of signals) or a combination of a logic-high signal and a logic-low signal.

The control circuit 320 can receive the trigger signals CMP_H and CMP_L and generate one or more control signals S_CTRL to control the compensation circuit 330 according to the trigger signals CMP_H and CMP_L, so as to control a frequency f_CMP of the trigger signals CMP_H/CMP_L to be indicative of (e.g., linearly proportional to) the sense signal V_SEN2. More specifically, in one embodiment, the compensation circuit 330 can provide compensation charges to the integral comparing circuit 310 such that positive charges pass through the integrating capacitor C_INT from the inverting input terminal of the OPA 312 to the output terminal of the OPA 312, and therefore the integral value V_INT decreases. Such compensation charges can be referred to as “positive compensation charges.” The compensation circuit 330 can also provide compensation charges to the integral comparing circuit 310 such that positive charges pass through the integrating capacitor C_INT from the output terminal of the OPA 312 to the inverting input terminal of the OPA 312, and therefore the integral value V_INT increases. Such compensation charges can be referred to as “negative compensation charges.” In one embodiment, if a trigger signal CMP_H is generated from the comparator 314, e.g., indicating that the integral value V_INT increases to the high-side reference V_H, then the control circuit 320 controls the compensation circuit 330 to provide positive compensation charges in a predetermined amount Q_REF to the integrating capacitor C_INT, so as to decrease the integral value V_INT by a predetermined value ΔV_INT. The predetermined value ΔV_INT can be determined by the value of Q_REF/C_INT. After it is decreased, the decreased integral value V_INT may continue to increase because of the integration of the sense signal V_SEN2. When the integral value V_INT increases to the high-side reference V_H, the compensation circuit 330 provides positive compensation charges Q_REF to the integrating capacitor C_INT again, so as to again decease the integral value V_INT by the predetermined value ΔV_INT. By compensating the integrating capacitor C_INT with the same amount of electric charges Q_REF each time the integral value V_INT increases to the high-side reference V_H, the integral value V_INT can ramp up and down alternately, and the integral comparing circuit 310 can generate a train of trigger signals CMP_H at a frequency f_CMP linearly proportional to the sense signal V_SEN2, e.g., linearly proportional to the current I_SEN.

Similarly, if a trigger signal CMP_L is generated from the comparator 316, e.g., indicating that the integral value V_INT decreases to the low-side reference V_L, then the control circuit 320 controls the compensation circuit 330 to provide negative compensation charges in the predetermined amount Q_REF to the integrating capacitor C_INT, so as to increase the integral value V_INT by a predetermined value ΔV_INT. The predetermined value ΔV_INT can be determined by the value of Q_REF/C_INT. The compensation circuit 330 can compensate the integrating capacitor C_INT with the same amount of electric charges Q_REF to increase the integral value V_INT each time the integral value V_INT decreases to the low-side reference V_L. Hence, the integral comparing circuit 310 can generate a train of trigger signals CMP_L at a frequency f_CMP linearly proportional to the sense signal V_SEN2, e.g., linearly proportional to the current I_SEN.

FIG. 4 illustrates a circuit diagram of an example of a compensation circuit 330 in a coulomb counter 400, in an embodiment according to the present invention. In one embodiment, the compensation circuit 330 described in relation to FIG. 3 has the circuit structure of the compensation circuit 330 disclosed in FIG. 4.

As shown in FIG. 4, the compensation circuit 330 includes a capacitive component C_COM, e.g., a compensation capacitor, a set of switches 434, e.g., including switches S_L1, S_L2, S_H1, S_H2, S_C1 and S_C2, and a voltage source 432. The capacitive component C_COM can be used to store and provide the abovementioned compensation charges Q_REF. Under control of the control circuit 320, the switches 434 can selectively connect the voltage source 432 to the capacitive component C_COM to charge the capacitive component C_COM so that the capacitive component C_COM has a voltage level V_REF of the voltage source 432. The switches 434 can also selectively connect the capacitive component C_COM to the integral comparing circuit 310 so that the compensation circuit 330 provides a compensation signal V_COM to compensate, e.g., increase or decrease, the integral value V_INT. By way of example, the voltage source 432 provides a reference voltage V_REF. The control circuit 320 can generate control signals S_CTRL1 and S_CTRL2 to turn on the switches S_C1 and S_C2 and turn off the switches S_L1, S_L2, S_H1 and S_H1, so that the capacitive component C_COM is charged to have a voltage level of V_REF. The capacitive component C_COM can store an amount of compensation charges Q_REF that is given by: Q_REF=V_REF*C_COM. The control circuit 320 can also generate control signals S_CTRL1 and S_CTRL2 to turn off the switches S_C1 and S_C2 and turn on the switches S_H1 and S_H2 or the switches S_L1 and S_L2, and therefore the two terminals of the capacitive component C_COM are coupled to the two input terminals of the OPA 312 respectively. In one embodiment, because the OPA 312 controls its two input terminals to have the same voltage level, which results in controlling a voltage across the capacitive component C_COM to be zero volts, the capacitive component C_COM can discharge compensation charges Q_REF to the integral comparing circuit 310. As shown in FIG. 4, when the switches S_H1 and S_H2 are on and the switches S_L and S_L2 are off, the compensation signal V_COM provided to the integral comparing circuit 310 is at a voltage level of V_REF, e.g., a positive level, and therefore the compensation circuit 330 provides positive compensation charges Q_REF to the integrating capacitor C_INT to decrease the integral value V_INT. When the switches S_L1 and S_L2 are on and the switches S_H1 and S_H2 are off, the compensation signal V_COM provided to the integral comparing circuit 310 is at a voltage level of −V_REF, e.g., a negative level, and therefore the compensation circuit 330 provides negative compensation charges Q_REF to the integrating capacitor C_INT to increase the integral value V_INT.

In one embodiment, the control circuit 320 controls the switches 434 according to the trigger signals CMP_H and CMP_L such that the compensation signal V_COM is selectively at the voltage level V_REF and a reversed level −V_REF of the voltage level V_REF. For example, as mentioned above, the capacitive component C_COM can be charged to have the voltage level V_REF when the switches S_C1 and S_C2 are on and the switches S_L1, S_L2, S_H1 and S_H2 are off. If the control circuit 320 detects that a trigger signal CMP_H is generated, e.g., indicating that the integral value V_INT increases to the high-side reference V_H, then the control circuit 320 turns off the switches S_C1 and S_C2, turns on the switches S_H1 and S_H2, and maintain the switches S_L1 and S_L2 off. Hence, the compensation circuit 330 provides a compensation signal V_COM at the voltage level V_REF to the integral comparing circuit 310 to decrease the integral value V_INT. For another example, the capacitive component C_COM can be charged to have a voltage level of V_REF when the switches S_C1 and S_C2 are on and the switches S_L1, S_L2, S_H1 and S_H2 are off. If the control circuit 320 detects that a trigger signal CMP_L is generated, e.g., indicating that the integral value V_INT decreases to the low-side reference V_L, then the control circuit 320 turns off the switches S_C1 and S_C2, turns on the switches S_L1 and S_L2, and maintain the switches S_H1 and S_H2 off. Hence, the compensation circuit 330 provides a compensation signal V_COM at the voltage level −V_REF to the integral comparing circuit 310 to increase the integral value V_INT.

In one embodiment, the compensation circuit 330 also include a filter circuit 436 such as a low-pass filter. The filter circuit 436 can be implemented in many different circuit structures, and FIG. 4 shows an example thereof. In the example of FIG. 4, the filter circuit 436 includes resistors R₁ and R₂ and a capacitor C_LPF. The filter circuit 436 can pass the compensation charges Q_REF from the compensation circuit 330 to the integral comparing circuit 310 to change the integral value V_INT. The filter circuit 436 can also buff, e.g., slow down, the flowing of the compensation charges Q_REF from the capacitive component C_COM to the integral comparing circuit 310, so as to reduce/smoothen the change in the integral value V_INT caused by the compensation charges Q_REF. More details are described in combination with FIG. 5A and FIG. 5B.

FIG. 5A illustrates examples of waveforms of the sense signal V_SEN2, the integral value V_INT, the trigger signal CMP_H, and a charge capacity Q_COM of the capacitive component C_COM, in an embodiment according to the present invention. FIG. 5A is described in combination with FIG. 3 and FIG. 4.

In the example of FIG. 5A, the sense signal V_SEN2 has a negative voltage level, and the integral value V_INT increases due to integration of the negative sense signal V_SEN2. At time t_N1, the integral value V_INT increases to the high-side reference V_H, and a trigger signal CMP_H (e.g., a falling edge of a signal) is generated. Hence, the control circuit 320 turns on the switches S_H1 and S_H2 and turns off the switches S_L1, S_L2, S_C1 and S_C2 so that the capacitive component C_COM discharges an amount of the compensation charges Q_REF to the integrating capacitor C_INT to decrease the integral value V_INT by a decrement ΔV_INT. In one embodiment, the decrement ΔV_INT can be considered to be Q_REF/C_INT, where Q_REF represents the amount of compensation charges provided to the integrating capacitor C_INT, and C_INT represents the capacitance of the integrating capacitor C_INT. In other words, as shown in FIG. 5A, at time t_N1, the integral value V_INT can be considered to decrease to the voltage level of V_H-Q_REF/C_INT, and then linearly increases from the voltage level of V_H-Q_REF/C_INT toward the high-side reference V_H. In one embodiment, because the filter circuit 436 such as a low-pass filter can reduce/smoothen the change in the integral value V_INT caused by the compensation charges Q_REF, the integral value V_INT does not decrease to the voltage level of V_H-Q_REF/C_INT in practice. However, because the change in the charge capacity of the integrating capacitor C_INT caused by the compensation charges Q_REF is equal to Q_REF, the change in a voltage across the integrating capacitor C_INT, e.g., the change in the integral value V_INT, caused by the compensation charges Q_REF can be considered to be equal to Q_REF/C_INT.

Between time t_N1 and time t_N2, the charge capacity Q_COM of the capacitive component C_COM decreases to zero because the two terminals of the capacitive component C_COM, coupled to the two input terminals of the OPA 312 respectively, are controlled to have the same voltage level by the OPA 312. At time t_N2, the control circuit 320 turns on the switches S_C1 and S_C2 and turns off the switches S_H1, S_H2, S_L1 and S_L2 so that the capacitive component C_COM is charged by the voltage source 432 and the charge capacity Q_COM of the capacitive component C_COM increases to Q_REF, e.g., Q_REF=V_REF*C_COM.

As shown in FIG. 5A, a trigger signal CMP_H is generated each time the integral value V_INT increases to the high-side reference V_H. In response to each trigger signal CMP_H, the capacitive component C_COM discharges to provide an amount of compensation charges Q_REF to the integrating capacitor C_INT to decrease the integral value V_INT by a decrement ΔV_INT, e.g., ΔV_INT=Q_REF/C_INT. The charge capacity Q_COM of the capacitive component C_COM can decrease to zero. After a preset time interval Δt from the generation of each trigger signal CMP_H, e.g., Δt=t_N2−t_N1=t_N4−t_N3=t_N6−t_N5 . . . , the capacitive component C_COM is charged by the voltage source 432. The charge capacity Q_COM of the capacitive component C_COM can increase to Q_REF. Thus, in one embodiment, each time the integral value V_INT increases to the high-side reference V_H, the compensation circuit 330 provides the same amount of compensation charges Q_REF to the integrating capacitor C_INT to decrease the integral value V_INT, e.g., by the same decrement Q_REF/C_INT. In one embodiment, an integration time T_INT, e.g., t_N3−t_N1, t_N5−t_N3, etc., for integrating the sense signal V_SEN2 so that the integral value V_INT increases from the voltage level V_H−Q_REF/C_INT to the voltage level V_H can be given by: T_INT=Q_REF*R_INT/V_SEN2. Thus, the integration time T_INT is inversely proportional to the sense signal V_SEN2. In other words, the trigger signals CMP_H are generated at a frequency f_CMP (e.g., 1/T_INT) that is linearly proportional to the sense signal V_SEN2.

FIG. 5B illustrates examples of waveforms of the sense signal V_SEN2, the integral value V_INT, the trigger signal CMP_L, and the charge capacity Q_COM of the capacitive component C_COM, in an embodiment according to the present invention. FIG. 5B is described in combination with FIG. 3, FIG. 4, and FIG. 5A. In the example of FIG. 5B, the sense signal V_SEN2 has a positive voltage level, and the integral value V_INT decreases due to integration of the positive sense signal V_SEN2.

Similar to the operations described in relation to FIG. 5A, as shown in FIG. 5B, a trigger signal CMP_L is generated each time the integral value V_INT decreases to the low-side reference V_L. In response to each trigger signal CMP_L, the capacitive component C_COM discharges to provide an amount of compensation charges Q_REF to the integrating capacitor C_INT to increase the integral value V_INT by an increment ΔV_INT, e.g., ΔV_INT=Q_REF/C_INT. The charge capacity Q_COM of the capacitive component C_COM can decrease to zero. After a preset time interval Δt from the generation of each trigger signal CMP_L, e.g., Δt=t_P2−t_P1=t_P4−t_N3=t_P6−t_P5 . . . , the capacitive component C_COM is charged by the voltage source 432. The charge capacity Q_COM of the capacitive component C_COM can increase to Q_REF. Thus, in one embodiment, each time the integral value V_INT decreases to the low-side reference V_L, the compensation circuit 330 provides the same amount of compensation charges Q_REF to the integrating capacitor C_INT to increase the integral value V_INT, e.g., by the same increment Q_REF/C_INT. In one embodiment, an integration time T_INT, e.g., t_P3−t_P1, t_P5−t_P3, etc., for integrating the sense signal V_SEN2 so that the integral value V_INT decreases from the voltage level V_L+Q_REF/C_INT to the voltage level V_L can be given by: T_INT=Q_REF*R_INT/V_SEN2. Thus, the integration time T_INT is inversely proportional to the sense signal V_SEN2. In other words, the trigger signals CMP_L are generated at a frequency f_CMP (e.g., 1/T_INT) that is linearly proportional to the sense signal V_SEN2.

Accordingly, in one embodiment, the analog-to-frequency converting circuit, e.g., including the sense resistor R_SEN, the switches 302, the integral comparing circuit 310, the control circuit 320, and the compensation circuit 330, can sense a current I_SEN flowing through the sense resistor R_SEN and generate a train of trigger signals CMP_H/CMP_L (e.g., a series of rising edges or a series of a falling edges) at a frequency f_CMP indicative of (e.g., linearly proportional to) the current I_SEN.

Returning to FIG. 4, the capacitive component C_COM includes a first terminal 450 and a second terminal 452 selectively coupled to a first terminal 454 of the OPA 312 and a second terminal 456 of the OPA 312 through the switches S_L1, S_L2, S_H1 and S_H1. The control circuit 320 controls the switches S_L1, S_L2, S_H1 and S_H1 such that if the first terminal 450 of the capacitive component C_COM is coupled to the first terminal 454 of the OPA 312, then the second terminal 452 of the capacitive component C_COM is coupled to the second terminal 456 of the OPA 312, and if the first terminal 450 of the capacitive component C_COM is coupled to the second terminal 456 of the OPA 312, then the second terminal 452 of the capacitive component C_COM is coupled to the first terminal 454 of the OPA 312. In other words, when the compensation circuit 330 provides compensation charges to the integral comparing circuit 310, the two terminals of the capacitive component C_COM are coupled to the two input terminals of the OPA 312 respectively. This capacitive component C_COM can be referred to as a “flying” or “floating” capacitor. Thus, terminals 340 and 342 of the sense resistor R_SEN, different from the terminal 156 of the sense resistor R′_SEN in FIG. 1, can have voltage levels independent of the compensation signal V_COM, e.g., V_REF or −V_REF, from the capacitive component C_COM. Advantageously, the sense resistor R_SEN can be placed at a negative terminal of a battery or a positive terminal of a battery.

Although FIG. 3 and FIG. 4 disclose that the positive terminal of the battery 344 is coupled to the terminal 342 of the sense resistor R_SEN, the invention is not so limited. In another embodiment, the positive terminal of the battery 344 can be coupled to the other terminal 340 of the sense resistor R_SEN. In yet another embodiment, the negative terminal of the battery 344 can be coupled to the terminal 340 or the terminal 342 of the sense resistor R_SEN.

Additionally, in one embodiment, the compensation circuit 330 can, but not necessarily, use one capacitor C_COM to provide compensation charges. Thus, compared with the compensation circuitry in FIG. 1 that includes two capacitors CP₁ and CP₁, the compensation circuit 330 can cost less and have a smaller size.

Moreover, in one embodiment, due to non-ideality of the comparators 314 and 316, there may be delays, e.g., Δt, in the generation of the trigger signals CMP_H and CMP_L. With reference to FIG. 5A, in one embodiment, the trigger signals CMP_H may be generated at times t_N1+Δt, t_N3+Δt, t_N5+Δt, etc., instead of times t_N1, t_N3, t_N5, etc., due to non-ideality of the comparator 314. In this embodiment, the delay Δt exists each time a trigger signal CMP_H is generated. Thus, the delays in generation of the trigger signals CMP_H may not result in error, e.g., a decrement, in the frequency f_CMP of the trigger signals CMP_H. Similarly, delays in generation of trigger signals CMP_L may not result in error, e.g., a decrement, in the frequency f_CMP of the trigger signals CMP_L.

Furthermore, as mentioned above, the filter circuit 436 such as a low-pass filter can buff, e.g., slow down, the flowing of the compensation charges Q_REF from the capacitive component C_COM to the integral comparing circuit 310, so as to reduce/smoothen the change in the integral value V_INT caused by the compensation charges Q_REF. Thus, a current generated by the OPA 312 and through the integrating capacitor C_INT and the capacitive component C_COM to discharge the capacitive component C_COM can be reduced, compared with the current generated to discharge the capacitor CP₁ described in relation to FIG. 1.

Returning to FIG. 3, the coulomb counter 300 also includes a logic circuit 304 and a counter 306. The counter 306 can count a number of the trigger signals S_FREQ, e.g., including signals CMP_H and CMP_L, to generate a count value N_COUNT. The count value N_COUNT can represent a result of integrating a current I_SEN flowing through the sense resistor R_SEN. In one embodiment, the current I_SEN can be a charging current or a discharging current of a battery 344. In one such embodiment, the count value N_COUNT can represent an amount of accumulation of electric charges in the battery current I_SEN, e.g., an amount of accumulation of electric charges passing in and out of the battery 344.

In one embodiment, the OPA 312 may include an input offset V_OS, e.g., an input voltage offset, which may cause error in the frequency f_CMP of the trigger signals S_FREQ, e.g., including signals CMP_H and CMP_L. Advantageously, as mentioned above, the control circuit 320 can generate a switching signal F_CHOP to control the switches 302 such that the sense signal V_SEN2 alternates between an original version of the sense signal V_SEN1 (e.g., V_SEN2=V_SEN1) and a reversed version of the sense signal V_SEN1 (e.g., V_SEN2=−V_SEN1), and that a first time interval T_A during which the sense signal V_SEN2 is in the original version V_SEN1 (e.g., V_SEN2=V_SEN1) and a second time interval T_B during which the sense signal V_SEN2 is in the reversed version −V_SEN1 (e.g., V_SEN2=−V_SEN1) are substantially the same. As used herein, “substantially the same” means that the time intervals T_A and T_B are controlled to be the same but a negligibly small difference may exist between the time intervals T_A and T_B due to non-ideality of the control circuit 320 and/or the switches 302. The counter 306 can count a number ΔN_SUM of the trigger signals S_FREQ for the first and second time interval to generate a count value, e.g., ΔN_SUM=ΔN₁+ΔN₂ where ΔN₁ represents the number of the trigger signals S_FREQ generated in the first time interval, and ΔN₂ represents the number of the trigger signals S_FREQ generated in the second time interval. For example, the counter 306 can increase the count value N_COUNT by an increment of ΔN₁+ΔN₂ if the current I_SEN is a charging current of the battery 344, or decrease the count value N_COUNT by a decrement of ΔN₁+ΔN₂ if the current I_SEN is a discharging current of the battery 344. As a result, the control circuit 320 can counteract error, caused by the input offset V_OS of the OPA 312, in the count value N_COUNT by controlling the first and second time intervals T_A and T_B to be substantially the same. More details will be described in combination with FIG. 7.

In one embodiment, the logic circuit 304 cooperates with the control circuit 320 to generate a count-direction signal S_D/U. The count-direction signal S_D/U can represent a flowing direction of the current I_SEN, e.g., indicate whether the current I_SEN is a charging current or a discharging current of the battery 344, and can control the counter 306 to increase or decrease the count value N_COUNT when a trigger signal S_FREQ is generated. More specifically, in one embodiment, the logic circuit 304 includes an XOR gate, and the control circuit 320 includes a circuit structure disclosed in FIG. 6A.

FIG. 6A illustrates a circuit diagram of an example of the control circuit 320, e.g., labeled 320A, in an embodiment according to the present invention. FIG. 6A is described in combination with FIG. 3 and FIG. 4. As shown in FIG. 6A, the control circuit 320A includes a switch control circuit 622 and a logic circuit 624A. The switch control circuit 622 can generate a switching signal F_CHOP to control the switches 302, and generate one or more control signals S_CTRL to control the compensation circuit 330, e.g., generate control signals S_CTRL1 and S_CTRL2 to control the switches 434 in FIG. 4. In one embodiment, when the switching signal F_CHOP is in a first status, e.g., logic high (or logic low), the switches 302 can provide a sense signal V_SEN2 at the voltage level of V_SEN1 to the integral comparing circuit 310; and when the switching signal F_CHOP is in a second status, e.g., logic low (or logic high; the opposite of the first status), the switches 302 can provide a sense signal V_SEN2 at the voltage level of −V_SEN1 to the integral comparing circuit 310. The switch control circuit 622 can control the switching signal F_CHOP to have a 50% duty cycle such that the abovementioned first time interval T_A and second time interval T_B are the same.

In one embodiment, the logic circuit 624A generates a trigger signal S_FREQ on detection of each trigger signal of the signals CMP_H and CMP_L. By way of example, the logic circuit 624A includes an AND gate 626A coupled to the comparators 314 and 316 and operable for receiving trigger signals CMP_H and CMP_L from the comparators 314 and 316. In the example of FIG. 6A, the comparator 314 receives the integral value V_INT at its inverting input terminal and receives the high-side reference V_H at its non-inverting input terminal. Thus, the trigger signal CMP_H, representing that the integral value V_INT has increased to the high-side reference V_H, is a logic-low signal, e.g., a falling edge of a signal. Similarly, the comparator 316 receives the integral value V_INT at its non-inverting input terminal and receives the low-side reference V_L at its inverting input terminal. Thus, the trigger signal CMP_L, representing that the integral value V_INT has decreased to the low-side reference V_L, is a logic-low signal, e.g., a falling edge of a signal. The AND gate 626A can output a trigger signal S_FREQ at logic-low when detecting a trigger signal CMP_H or CMP_L, e.g., a logic-low signal, at its input terminals. Consequently, the trigger signals S_FREQ represent a combination of the trigger signal CMP_H and CMP_L and can be considered to include the trigger signal CMP_H or CMP_L.

In one embodiment, the logic circuit 624A cooperates with the switch control circuit 622 and the logic circuit 304 to generate a count-direction signal S_D/U.

The count-direction signal S_D/U determines whether to increase or decrease the abovementioned count value N_COUNT when a trigger signal S_FREQ is generated. More specifically, in one embodiment, the logic circuit 624A generates a polarity signal S_POL, sets the polarity signal S_POL to a first logic level on detection of the trigger signal CMP_H, and sets the polarity signal S_POL to a second logic level on detection of the trigger signal CMP_L. The first and second logic levels are different. As used herein, two logic levels are “different” if the two logic levels are inverted relative to each other. For example, if one of the logic levels is logic high, then the other one of the logic levels is logic low; and if one of the logic levels is logic low, then the other one of the logic levels is logic high. In the example of FIG. 6A, a set-reset NAND latch 628A in the logic circuit 624A can set the polarity signal S_POL to be logic high on detection of a trigger signal CMP_H at its set input terminal labeled “S”, and set the polarity signal S_POL to be logic low on detection of a trigger signal CMP_L at its reset input terminal labeled “R”. In one embodiment, as mentioned above, a trigger signal CMP_H can be generated during a situation where the sense signal V_SEN2 is positive, and a trigger signal CMP_L can be generated during a situation where the sense signal V_SEN2 is negative. Thus, the polarity signal S_POL can represent whether the second sense signal V_SEN2 is positive or negative.

Additionally, the logic circuit 304 such as an XOR gate can set the count-direction signal S_D/U to a third logic level if a logic level of the polarity signal S_POL and a logic level of the switching signal F_CHOP are the same, and set the count-direction signal S_D/U to a fourth logic level if a logic level of the polarity signal S_POL and a logic level of the switching signal F_CHOP are the different. The third and fourth logic levels are different, similar to the abovementioned first and second logic levels. As used herein, two logic levels are “the same” if the two logic levels are either logic high at the same time or logic low at the same time. In the example of FIG. 6A, the logic circuit 304 can set the count-direction signal S_D/U to be logic low if the polarity signal S_POL and the switching signal F_CHOP both are logic high or logic low. The logic circuit 304 can also set the count-direction signal S_D/U to be logic high if the polarity signal S_POL is logic high and the switching signal F_CHOP is logic low, or if the polarity signal S_POL is logic low and the switching signal F_CHOP is logic high. In one embodiment, whether the polarity signal S_POL and the switching signal F_CHOP have the same logic level is determined by the direction the current I_SEN is flowing through the sense resistor R_SEN (shown in FIG. 3). Thus, the logic level of the count-direction signal S_D/U can represent whether the current I_SEN is a charging current or a discharging current of the battery 344, and therefore can determine whether to increase or decrease the abovementioned count value N_COUNT when a trigger signal S_FREQ is generated. More details are described in combination with FIG. 7.

FIG. 7 illustrates examples of waveforms of the first sense signal V_SEN1, the second sense voltage V_SEN2, the trigger signals S_FREQ, the polarity signal S_POL, the switching signal F_CHOP, the count-direction signal S_D/U, and the count value N_COUNT, in an embodiment according to the present invention. FIG. 7 is described in combination with FIG. 3, FIG. 4, and FIG. 6A. In the example of FIG. 7, during time to and time t₄, the battery 344 is in a charging mode, the first sense signal V_SEN1 is positive, and the count value N_COUNT increases in response to each trigger signal S_FREQ; and during time t₄ to time t₈, the battery 344 is in a discharging mode, the first sense signal V_SEN1 is negative, and the count value N_COUNT decreases in response to each trigger signal S_FREQ.

In the example of FIG. 7, the control circuit 320 can turn off the switches S_A1 and S_A2 and turn on the switches S_B1 and S_B2 by setting the switching signal F_CHOP to be logic high, and can turn off the switches S_B1 and S_B2 and turn on the switches S_A1 and S_A2 by setting the switching signal F_CHOP to be logic low. Thus, as shown in FIG. 7, when the battery 344 is in a charging mode, e.g., from time t₀ to time t₄, the second sense voltage V_SEN2 is negative if the switching signal F_CHOP is logic high, and is positive if the switching signal F_CHOP is logic low. When the battery 344 is in a discharging mode, e.g., from time t₄ to time t₈, the second sense voltage V_SEN2 is positive if the switching signal F_CHOP is logic high, and is negative if the switching signal F_CHOP is logic low.

From time t₀ to time t₁, the switching signal F_CHOP is logic high, and the second sense voltage V_SEN2 is negative. Hence, a set of trigger signals CMP_H are generated, and the latch 628A sets the polarity signal S_POL to be logic high accordingly. Because the polarity signal S_POL and the switching signal F_CHOP have the same logic level, e.g., logic high, the logic circuit 304 sets the count-direction signal S_D/U to be logic low. Similarly, from time t₁ to time t₂, the switching signal F_CHOP is logic low, and the second sense voltage V_SEN2 is positive. Hence, a set of trigger signals CMP_L are generated, and the latch 628A sets the polarity signal S_POL to be logic low accordingly. Because the polarity signal S_POL and the switching signal F_CHOP have the same logic level, e.g., logic low, the logic circuit 304 sets the count-direction signal S_D/U to be logic low. In addition, from time t₄ to time t₅, the switching signal F_CHOP is logic high, and the second sense voltage V_SEN2 is positive. Hence, a set of trigger signals CMP_L are generated, and the latch 628A sets the polarity signal S_POL to be logic low accordingly. Because the polarity signal S_POL and the switching signal F_CHOP have different logic levels, the logic circuit 304 sets the count-direction signal S_D/U to be logic high. Similarly, from time t₅ to time t₆, the switching signal F_CHOP is logic low, and the second sense voltage V_SEN2 is negative. Hence, a set of trigger signals CMP_H are generated, and the latch 628A sets the polarity signal S_POL to be logic high accordingly. Because the polarity signal S_POL and the switching signal F_CHOP have different logic levels, the logic circuit 304 sets the count-direction signal S_D/U to be logic high. Thus, in this example of FIG. 7, the count-direction signal S_D/U is logic low if the battery 344 is in a charging mode, and is logic high if the battery 344 is in a discharging mode. If the counter 306 detects that the count-direction signal S_D/U is logic low, then the counter 306 increases the count value N_COUNT in response to each trigger signal S_FREQ; and if the counter 306 detects that the count-direction signal S_D/U is logic high, then the counter 306 decreases the count value N_COUNT in response to each trigger signal S_FREQ.

FIG. 7 shows examples of the signals associated with the coulomb counter 300 (or 400) in one embodiment, but the invention is not limited to these examples. In another embodiment, the control circuit 320 can turn off the switches S_A1 and S_A2 and turn on the switches S_B1 and S_B2 by setting the switching signal F_CHOP to be logic low, and can turn off the switches S_B1 and S_B2 and turn on the switches S_A1 and S_A2 by setting the switching signal F_CHOP to be logic high. In one such embodiment, if the count-direction signal S_D/U is logic high, then it indicates that the battery 344 is in a charging mode, and the counter 306 increases the count value N_COUNT in response to each trigger signal S_FREQ. If the count-direction signal S_D/U is logic low, then it indicates that the battery 344 is in a discharging mode, and the counter 306 decreases the count value N_COUNT in response to each trigger signal S_FREQ. In yet other embodiments, the battery 344 may be coupled to the sense resistor R_SEN in other manners, and the signals V_SEN1, V_SEN2, S_FREQ, S_POL, F_CHOP, and S_D/U can have different waveforms accordingly.

Additionally, as mentioned above, the control circuit 320 can counteract error, caused by an input offset V_OS of the OPA 312, in the count value N_COUNT by controlling the first and second time intervals T_A and T_B to be substantially the same. Taking FIG. 7 as an example, the reference line labeled “V_R” represents a voltage level at the terminal 342 of the sense resistor R_SEN, and the reference line labeled “V′_R” represents a voltage level at the non-inverting input terminal of the OPA 312. A voltage difference, e.g., labeled “V_OS” in FIG. 7, between the voltage levels V_R and V′_R can represent an input offset of the OPA 312. Thus, from time t₀ to time t₁, the integral comparing circuit 310 integrates a voltage level of −(V_SEN1+V_OS) and generates a number ΔN₁ of the trigger signals CMP_H. The number ΔN₁ represents a result of integrating an absolute value |V_SEN1+V_OS| for the time duration between to and t₁ (e.g., referred to as “second time interval T_B”). From time t₁ to time t₂, the integral comparing circuit 310 integrates a voltage level of V_SEN1−V_OS and generates a number ΔN₂ of the trigger signals CMP_L. The number ΔN₂ represents a result of integrating an absolute value |V_SEN1−V_OS| for the time duration between t₁ and t₂ (e.g., referred to as “first time interval T_A”). In one embodiment, the control circuit 320 controls the switching signal F_CHOP to have a 50% duty cycle such that the first time interval T_A and the second time interval T_B are the same. Consequently, the error caused by integrating the input offset V_OS for the first time interval T_A and the error caused by integrating the input offset V_OS for the second time interval T_B can be counteracted by each other. The sum ΔN_SUM of the numbers ΔN₁ and ΔN₂ can represent a result of integrating an absolute value |V_SEN2|.

FIG. 6B illustrates a circuit diagram of another example of the control circuit 320, e.g., labeled 320B, in an embodiment according to the present invention.

FIG. 6B is described in combination with FIG. 3, FIG. 4, and FIG. 6A. The example of FIG. 6B is similar to the example of FIG. 6A except that the comparator 314 in FIG. 6B receives the integral value V_INT at its non-inverting input terminal and receives the high-side reference V_H at its inverting input terminal, the comparator 316 receives the integral value V_INT at its inverting input terminal and receives the low-side reference V_L at its non-inverting input terminal, and the control circuit 320B includes a logic circuit 624B. In the example of FIG. 6B, trigger signals CMP_H and CMP_L generated by the comparators 314 and 316 are logic-high signals, e.g., rising edges. The logic circuit 624B includes an OR gate 626B and a set-reset latch 628B. The OR gate 626B can detect the trigger signals CMP_H and CMP_L, and output a logic-high trigger signal S_FREQ if detecting that a trigger signal CMP_H or CMP_L is generated. The latch 628B, similar to the latch 628A, can set the polarity signal S_POL to a first logic level, e.g., logic high, on detection of a trigger signal CMP_H, and sets the polarity signal S_POL to a second logic level, e.g., logic low, on detection of a trigger signal CMP_L. Similar to the example of FIG. 6A, the logic circuit 624B can cooperate with the switch control circuit 622 and the logic circuit 304 to generate a count-direction signal S_D/U, and the count-direction signal S_D/U determines whether to increase or decrease the count value N_COUNT when a trigger signal S_FREQ is generated.

Although, as disclosed in FIG. 6A (and similarly disclosed in FIG. 6B), the output terminal of the compactor 314 is coupled to the set terminal of the latch 628A (or 628B), and the output terminal of the compactor 316 is coupled to the reset terminal of the latch 628A (or 628B), the invention is not so limited. In another embodiment, the output terminal of the compactor 314 can be coupled to the reset terminal of the latch 628A (or 628B), and the output terminal of the compactor 316 is coupled to the set terminal of the latch 628A (or 628B).

Returning to FIG. 3, in one embodiment, in operation, the control circuit 320 generates a switching signal F_CHOP to control the switches 302 such that the second sense voltage V_SEN2 alternates between the voltage levels of V_SEN1 and −V_SEN. Based on the integrating of the second sense voltage V_SEN2, the comparing between the integral value V_INT and the references V_H and V_L, and the compensating with electric charges from the compensation circuit 330, the integral comparing circuit 310 can alternately generate a set of trigger signals CMP_H and a set of trigger signals CMP_L. The control circuit 320 can generate a set of trigger signals S_FREQ that present a combination of the trigger signals CMP_H and CMP_L. The control circuit 320 can also set a polarity signal S_POL to a first logic level when receiving a trigger signal CMP_H, and set the polarity signal S_POL to a second logic level when receiving a trigger signal CMP_L. The logic circuit 304 can output a count-direction signal S_D/U according to the polarity signal S_POL and the switching signal F_CHOP. The counter 306 can count the number of the trigger signals S_FREQ to generate a count value N_COUNT, and increase or decrease the count value N_COUNT according to the count-direction signal S_D/U.

FIG. 8 illustrates a flowchart 800 of examples of operations performed by a coulomb counter, e.g., 300 or 400, in an embodiment according to the present invention. FIG. 8 is described in combination with FIG. 3, FIG. 4, FIG. 5A, FIG. 5B, FIG. 6A, FIG. 6B, and FIG. 7.

In block 802, the sense resistor R_SEN generates a first sense signal V_SEN1 indicative of a current I_SEN flowing through the sense resistor R_SEN.

In block 804, the switches 302 provide a second sense signal V_SEN2 that alternates between an original version of the first sense signal V_SEN1 (e.g., V_SEN2=V_SEN1) and a reversed version of the first sense signal V_SEN1 (e.g., V_SEN2=−V_SEN1).

In block 806, the integral comparing circuit 310 integrates the second sense signal V_SEN2 to generate an integral value V_INT.

In block 808, the integral comparing circuit 310 generates a train of trigger signals S_FREQ, e.g., including trigger signals CMP_H and CMP_L. Each trigger signal of the trigger signals S_FREQ is generated when the integral value V_INT reaches a preset reference V_H or V_L.

In block 810, the compensation circuit 330 compensates for the integral value V_INT with a predetermined value, e.g., Q_REF/C_INT, in response to each trigger signal of the trigger signals S_FREQ.

In block 812, the control circuit 320 controls a first time interval T_A, during which the second sense signal V_SEN2 is the original version of V_SEN1, and a second time interval T_B, during which the second sense signal V_SEN2 is the reversed version of V_SEN1, to be substantially the same.

In summary, embodiments according to the present invention provide analog-to-frequency converting circuits to convert an analog signal, e.g., a current of a battery, to a train of trigger signals at a frequency. By using an integral comparing circuit, compensation circuit, and control circuit in the analog-to-frequency converting circuit, in one embodiment according to the present invention, to generate the trigger signals, the frequency of the trigger signals can be linearly proportional to the current of the battery. A coulomb counter can count the trigger signals to generate a count value, and the count value can represent an accumulation of charges passing in and out of the battery. In one embodiment, by controlling the abovementioned first time interval T_A and second time interval T_B to be substantially the same, error in the count value caused by an input offset of the integral comparing circuit can be eliminated. The analog-to-frequency converting circuit can be used for coulomb counting in many applications such as battery monitoring and battery management in various portable electronic devices, e.g., mobile phones, cameras, laptop computers, tablet computers, GPS navigation devices, etc. The analog-to-frequency converting circuit can also be used in other situations in which an analog signal is needed to be converted to a frequency signal to indicate the analog signal.

While the foregoing description and drawings represent embodiments of the present invention, it will be understood that various additions, modifications and substitutions may be made therein without departing from the spirit and scope of the principles of the present invention as defined in the accompanying claims. One skilled in the art will appreciate that the invention may be used with many modifications of form, structure, arrangement, proportions, materials, elements, and components and otherwise, used in the practice of the invention, which are particularly adapted to specific environments and operative requirements without departing from the principles of the present invention. The presently disclosed embodiments are therefore to be considered in all respects as illustrative and not restrictive, the scope of the invention being indicated by the appended claims and their legal equivalents, and not limited to the foregoing description.

Claims

1. An analog-to-frequency converting circuit comprising:

a first plurality of switches operable for receiving a first sense signal indicative of a current and providing a second sense signal that alternates between an original version of said first sense signal and a reversed version of said first sense signal, under control of a switching signal;

an integral comparing circuit, coupled to said first plurality of switches, operable for integrating said second sense signal to generate an integral value, and operable for generating a train of trigger signals at a frequency indicative of said current, wherein each trigger signal of said trigger signals is generated when said integral value reaches a preset reference;

a compensation circuit, coupled to said integral comparing circuit, operable for compensating for said integral value with a predetermined value in response to each trigger signal of said trigger signals; and

a control circuit, coupled to said first plurality of switches, operable for generating said switching signal such that a first time interval during which said second sense signal is said original version and a second time interval during which said second sense signal is said reversed version are substantially the same.

2. The analog-to-frequency converting circuit of claim 1, wherein said control circuit counteracts error, caused by an input offset of said integral comparing circuit, in a count value by controlling said first and second time intervals to be substantially the same, and wherein said count value is obtained by counting a number of said trigger signals and represents an amount of accumulation of electric charges in said current.

3. The analog-to-frequency converting circuit of claim 1, wherein said integral comparing circuit comprises:

comparator circuitry operable for comparing said integral value with a first reference and a second reference that is less than said first reference, generating a first trigger signal if said integral value is greater than said first reference, and generating a second trigger signal if said integral value is less than said second reference, wherein said preset reference is one of said first and second references.

4. The analog-to-frequency converting circuit of claim 3, wherein if said first trigger signal is generated, then said compensation circuit provides compensation charges in a predetermined amount to said integral comparing circuit to decrease said integral value by a predetermined value, and wherein if said second trigger signal is generated, then said compensation circuit provides compensation charges in said predetermined amount to said integral comparing circuit to increase said integral value by a predetermined value.

5. The analog-to-frequency converting circuit of claim 1, wherein said compensation circuit comprises:

a capacitive component; and

a second plurality of switches, coupled to said capacitive component, operable for selectively connecting a voltage source to said capacitive component to charge said capacitive component so that said capacitive component has a voltage level of said voltage source, and selectively connecting said capacitive component to said integral comparing circuit so that said compensation circuit provides a compensation signal to compensate for said integral value,

wherein said control circuit controls said second plurality of switches according to said trigger signals such that said compensation signal is selectively at said voltage level and a reversed level of said voltage level.

6. The analog-to-frequency converting circuit of claim 5, wherein said capacitive component comprises a first terminal and a second terminal selectively coupled to a first terminal of said integral comparing circuit and a second terminal of said integral comparing circuit through said second plurality of switches, and wherein said control circuit controls said second plurality of switches such that if said first terminal of said capacitive component is coupled to said first terminal of said integral comparing circuit, then said second terminal of said capacitive component is coupled to said second terminal of said integral comparing circuit, and such that if said first terminal of said capacitive component is coupled to said second terminal of said integral comparing circuit, then said second terminal of said capacitive component is coupled to said first terminal of said integral comparing circuit.

7. The analog-to-frequency converting circuit of claim 1, wherein said compensation circuit comprises a filter circuit operable for passing compensation charges from said compensation circuit to said integral comparing circuit to change said integral value and reducing a change in said integral value caused by said compensation charges.

8. A coulomb counter comprising:

an analog-to-frequency converting circuit operable for converting a current to a train of trigger signals at a frequency indicative of said current, said analog-to-frequency converting circuit comprising: a first plurality of switches operable for receiving a first sense signal indicative of said current and providing a second sense signal that alternates between an original version of said first sense signal and a reversed version of said first sense signal, under control of a switching signal; an integral comparing circuit, coupled to said first plurality of switches, operable for integrating said second sense signal to generate an integral value, and operable for generating a trigger signal of said trigger signals when said integral value reaches a preset reference; a compensation circuit, coupled to said integral comparing circuit, operable for compensating for said integral value with a predetermined value in response to each trigger signal of said trigger signals; and a control circuit, coupled to said first plurality of switches, operable for generating said switching signal such that a first time interval during which said second sense signal is said original version and a second time interval during which said second sense signal is said reversed version are substantially the same; and

a counter, coupled to said analog-to-frequency converting circuit, operable for counting a number of said trigger signals to generate a count value representing an amount of accumulation of electric charges in said current.

9. The coulomb counter of claim 8, wherein said control circuit counteracts error, caused by an input offset of said integral comparing circuit, in said count value by controlling said first and second time intervals to be substantially the same.

10. The coulomb counter of claim 8, wherein said integral comparing circuit comprises:

comparator circuitry operable for comparing said integral value with a first reference and a second reference that is less than said first reference, generating a first trigger signal if said integral value is greater than said first reference, and generating a second trigger signal if said integral value is less than said second reference, wherein said preset reference is one of said first and second references.

11. The coulomb counter of claim 10, further comprising:

a first logic circuit, coupled to said comparator circuitry, operable for generating a polarity signal, setting said polarity signal to a first logic level on detection of said first trigger signal, and setting said polarity signal to a second logic level on detection of said second trigger signal, wherein said first and second logic levels are different; and

a second logic circuit, coupled to said first logic circuit and said control circuit, operable for generating a count-direction signal, setting said count-direction signal to a third logic level if a logic level of said polarity signal and a logic level of said switching signal are the same, and setting said count-direction signal to a fourth logic level if a logic level of said polarity signal and a logic level of said switching signal are different, wherein said third and fourth logic levels are different, and wherein said count-direction signal controls said counter to increase or decrease said count value when a trigger signal of said trigger signals is generated.

12. The coulomb counter of claim 10, wherein if said first trigger signal is generated, then said compensation circuit provides compensation charges in a predetermined amount to said integral comparing circuit to decrease said integral value by a predetermined value, and wherein if said second trigger signal is generated, then said compensation circuit provides compensation charges in said predetermined amount to said integral comparing circuit to increase said integral value by a predetermined value.

13. The coulomb counter of claim 8, wherein said compensation circuit comprises:

a capacitive component; and

a second plurality of switches, coupled to said capacitive component, operable for selectively connecting a voltage source to said capacitive component to charge said capacitive component so that said capacitive component has a voltage level of said voltage source, and selectively connecting said capacitive component to said integral comparing circuit so that said compensation circuit provides a compensation signal to compensate for said integral value,

wherein said control circuit controls said second plurality of switches according to said trigger signals such that said compensation signal is selectively at said voltage level and a reversed level of said voltage level.

14. A method comprising:

generating a first sense signal indicative of a current;

providing a second sense signal that alternates between an original version of said first sense signal and a reversed version of said first sense signal;

integrating said second sense signal to generate an integral value;

generating a train of trigger signals, wherein each trigger signal of said trigger signals is generated when said integral value reaches a preset reference;

compensating for said integral value with a predetermined value in response to each trigger signal of said trigger signals; and

controlling a first time interval during which said second sense signal is said original version and a second time interval during which said second sense signal is said reversed version to be substantially the same.

15. The method of claim 14, further comprising:

generating said trigger signals at a frequency indicative of said current.

16. The method of claim 14, further comprising:

counting a number of said trigger signals to generate a count value representing an amount of accumulation of electric charges in said current.

17. The method of claim 14, further comprising:

comparing said integral value with a first reference and a second reference that is less than said first reference;

generating a first trigger signal if said integral value is greater than said first reference; and

generating a second trigger signal if said integral value is less than said second reference,

wherein said preset reference is one of said first and second references.

18. The method of claim 17, further comprising:

setting a polarity signal to a first logic level on detection of said first trigger signal;

setting said polarity signal to a second logic level on detection of said second trigger signal, wherein said first and second logic levels are different;

setting a count-direction signal to a third logic level if a logic level of said polarity signal and a logic level of said switching signal are the same;

setting said count-direction signal to a fourth logic level if a logic level of said polarity signal and a logic level of said switching signal are different, wherein said third and fourth logic levels are different; and

controlling said counter to increase or decrease said count value when a trigger signal of said trigger signals is generated, according to said count-direction signal.

19. The method of claim 17, wherein said compensating comprises:

providing compensation charges in a predetermined amount to an integral comparing circuit that performs said integrating, to decrease said integral value by a predetermined value if said first trigger signal is generated; and

providing compensation charges in said predetermined amount to said integral comparing circuit to increase said integral value by a predetermined value if said second trigger signal is generated.

20. The method of claim 14, wherein said compensating comprises:

selectively connecting a voltage source to a capacitive component in a compensation circuit that performs said compensating, to charge said capacitive component so that said capacitive component has a voltage level of said voltage source;

selectively connecting said capacitive component to an integral comparing circuit that performs said integrating, so that said compensation circuit provides a compensation signal to compensate for said integral value; and

controlling said compensation signal to be selectively at said voltage level and a reversed level of said voltage level according to said trigger signals.