Constant-peak-current minimum-off-time pulse frequency modulator for switching regulators

- Monolithic Power Systems

Methods and apparatus are disclosed for controlling switching regulators to automatically switch from a traditional pulse-width modulation (PWM) mode, when the load current is higher than a predetermined value, to a pulse-frequency modulation (PFM) mode, when the load current drops to a point lower than the predetermine value. Switching modes increases the efficiency of the regulator when the load current is low without provoking a large output voltage transient. In some embodiments, a single switching cycle regulates the output in the PFM mode, which reduces the switching losses and enhances the efficiency of the regulator.

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Description
CROSS REFERENCE TO RELATED APPLICATION(S)

This application is a continuation of U.S. patent application Ser. No. 11/261,874 filed Oct. 28, 2005 to which the benefit of priority is claimed.

TECHNICAL FIELD

The embodiments described below relate, in general, to switching regulators and, in particular, to controlling schemes for switching regulators during high and low load currents.

BACKGROUND

When higher efficiency, smaller size, lighter weight, and prolonging battery lifetime are required, switching regulators are used for power management as replacements for simpler linear regulators. Furthermore, while the linear regulators can only step down, the switching regulators can be configured to step up (boost), step down (buck), or invert the output voltage with respect to the input voltage. A switching regulator is basically a circuit for regulating the voltage across a load. It transfers energy from input to output in discrete packets and a feedback circuitry regulates the energy transfer to maintain a constant voltage within the load limits of the circuit. Portable devices often operate in low-power stand-by mode because increasing the efficiency significantly increases the battery life.

The popular control scheme for switching regulators is the pulse-width modulation (PWM). The control uses a constant switching frequency but varies the duty cycle as the load current varies. This scheme achieves good regulation, low noise spectrum, and high efficiency; however, when the load current is low the PWM mode is not efficient due to the switching loss and the high quiescent current.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a schematic diagram of a 5-pin boost converter to exemplify disclosed automatically adaptable switching regulators, in accordance with an embodiment of the invention.

FIG. 2 shows an application of the 5-pin boost converter along with external components and arrangements.

FIG. 3 shows some key waveforms during a transition from PWM to PFM mode.

FIG. 4 shows some key waveforms during a transition from PFM to PWM mode.

FIGS. 5 and 6 illustrate simulation results of a transition from PWM to PFM mode and a transition from PFM to PWM mode, respectively.

FIG. 7 shows transient effects of a load current changing from 5 mA to 200 mA and back to 5 mA.

 DETAILED DESCRIPTION

Control of the switching regulators by PWM, when the load current is low (light load), is not efficient due to the switching loss and the high quiescent current. In such conditions, the pulse frequency modulation (PFM) mode is preferable. The disclosed embodiments describe methods and apparatus that automatically switch between the PWM mode and the PFM mode without provoking a large output voltage transient.

Switching regulators generally experience two kinds of losses: switching losses and conduction losses. While the output of a switching regulator feeds a light load, the regulator's efficiency can be enhanced by reducing the switching losses. In PFM mode, a switching regulator switches as many cycles as necessary to keep the output regulated. Once it achieves regulation, the switching stops until the output needs regulation again, at which time the switching cycles start again. In the existing art, a switching regulator has to switch several cycles before it stops switching. However, in the embodiments disclosed here, a single switching cycle can regulate the output. By having only a single pulse in PFM mode, the switching losses are reduced, which enhances the efficiency of the regulator.

Various embodiments of the invention will now be described. The following description provides specific details for a thorough understanding and enabling description of these embodiments. One skilled in the art will understand, however, that the invention may be practiced without many of these details. Additionally, some well-known structures or functions may not be shown or described in detail, so as to avoid unnecessarily obscuring the relevant description of the various embodiments.

The terminology used in the description presented below is intended to be interpreted in its broadest reasonable manner, even though it is being used in conjunction with a detailed description of certain specific embodiments of the invention. Certain terms may even be emphasized below; however, any terminology intended to be interpreted in any restricted manner will be overtly and specifically defined as such in this Detailed Description section.

The description of the embodiments of the invention and their applications as set forth herein is illustrative and is not intended to limit the scope of the invention. Variations and modifications of the embodiments are possible and practical alternatives to, or equivalents of the various elements of, the embodiments disclosed herein and are known to those of ordinary skill in the art. Such variations and modifications of the disclosed embodiments may be made without departing from the scope and spirit of the invention.

In the disclosed embodiments, when the load current is high, a traditional peak-current constant-frequency scheme is used in a PWM mode. When the load current drops to or below a predetermine value, the controller enters constant-peak-current minimum-off-time PFM mode, where the switching frequency is lower, resulting in higher efficiency and lower switching loss. Also, the quiescent current consumption is significantly reduced for better standby performance. Another advantage of this scheme is that there is no pulse group at PFM that leads to lower output ripple.

FIG. 1 shows a simplified schematic diagram of an embodiment of the invention. A 5-pin boost converter 100 is employed in this embodiment for illustration purposes. The controller part comprises a reference block U12, an error amplifier U1, a loop compensation network R1, C1, and C2, a PWM comparator U4, a current limit comparator U3, a light load comparator U5, a current sense amplifier U6, a feedback comparator U2, a timer U8, an oscillator U7, a driver U11, and an N-channel power MOSFET, SW. FIG. 2 shows an exemplary application of the 5-pin boost converter along with external components and connections.

PWM Mode

Operating in PWM mode, the error amplifier U1 regulates the output voltage by monitoring the FB pin voltage, which receives some specified portion of the output voltage VOUT, and compares it to an internal reference voltage VREF to generate the control voltage VCONTROL by transistor MN2. At the beginning of each clock cycle, RS-flip-flop X18 is set and the N-channel switch SW turns on, forcing the inductor L1 current to rise. The current at the sense resistor RSEN is amplified by current sense amplifier U6 and converted to voltage VIPK by MP5 and R6. VIPK is then compared to VCONTROL by PWM comparator U4. When inductor L1 current rises sufficiently, which makes VIPK equal to VCONTROL, the PWM comparator U4 goes high to reset RS-flip-flop X18 and turns off the N-channel switch SW, forcing the inductor L1 current to the output capacitor though an external rectified diode D1.

The control voltage VCONTROL rises up or falls down to control the peak inductor L1 current, which in turn is controlled by the output voltage VOUT. Thus, the output voltage VOUT controls the inductor L1 current to satisfy the load. MP6 generates slope compensation current when the duty cycle is greater than 40%. MP4 and R5 set the IMAX voltage VIMAX threshold for the current limit comparator U3 to limit the peak current.

PFM Mode

Operating in PFM mode, the error amplifier U1, the PWM comparator U4, the current limit comparator U3, and the oscillator U7 are disabled to lower the quiescent current. The controller runs in discontinuous conduction mode (DCM) with constant peak current and variable off-time. Feedback comparator U2 compares output voltage VOUT via the FB pin to internal reference voltage VREF. When VOUT<VREF, the RS-flip-flop X14 is set and the N-channel SW turns on, forcing the inductor L1 current to rise.

The current at the sense resistor RSEN is amplified by current sense amplifier U6 and converted to voltage VIPK by MP5 and R6. VIPK is compared to VIMIN at the connection of MN3 and R3 by light load comparator U5. When the inductor L1 current rises to a predetermined value (normally set to 10-20% of maximum peak current), which makes VIPK equal to VIMIN, the light load comparator U5 goes high to reset the RS-flip-flop X14 and turns off the N-channel switch SW, forcing inductor L1 current to charge up the output voltage VOUT through an external rectified diode.

The control keeps idling until the FB voltage VFB drops to VREF. The switching frequency is proportional to the load current, thus the switching loss is reduced. During each switching cycle, the controller only generates one “on” pulse to send a fixed amount of inductor L1 energy to the output. Therefore, the output voltage VOUT ripple is much smaller compared to the burst mode architecture.

Mode Transition

Disclosed embodiments provide a transition between modes that minimizes the output voltage VOUT transient. FIG. 3 shows some of the key waveforms during a transition from PWM mode to PFM. At the end of every “on” cycle, the controller monitors the peak current through the light load comparator U5. If the peak current is lower than the preset value (VIPK<VIMIN) for two consecutive cycles, and the output voltage VOUT is higher than a target value, which is decided by feedback comparator U2, then an active high of CLK will set the RS-flip-flop X12. This results in MODE=1 and in switching to PFM mode.

Two processes take place when MODE=1. First, as MN4 turns off, VIMIN is switched from VIMIN1 to a higher voltage VIMIN2, which results in a higher peak current than the controller is demanding and a reduction of the switching frequency in PFM mode. Second, the error amplifier U1 output VEAO is tri-stated and is clamped to MN2 gate voltage, which has gate and drain connected to the current source MP2. Note that when MODE=1, MP7 is “off” and MN7 is “on”. In this situation, because MP2=MP3, MN2=MN3, and R2=R3+R4, then VCNTRL=VIMIN=VIMIN2. Therefore, when the controller switches back to PWM mode, the error amplifier U1 output VEAO is close to the controlled value.

FIG. 4 shows a transition from PFM to PWM mode. At the end of every “on” cycle, the timer U8 starts and ends with T1=1 to indicate a minimum “off” time. If the output voltage VOUT is still less than the target value, which is determined by the feedback comparator U2 when T1=1, then output Q of D-flip-flop X4 goes high to reset the RS-flip-flop X12, which results in MODE=0 and in switching back to the PWM mode.

FIGS. 5 and 6 show simulations of the transitions from PWM to PFM mode and from PFM to PWM mode, respectively. Top trace is the MODE, below the MODE is the load current, and the third trace is the inductor L1 current. The output voltage is depicted by the bottom trace. When the load current drops to 9 mA, for example, the controller switches to PFM mode and switches back to PWM mode when the load current crosses 17 mA.

FIG. 7 shows some signal transitions as the load current changes from 5 mA to 200 mA and back to 5 mA. In this Figure, the top trace is the MODE, the middle trace is the inductor L1 current, and the bottom trace is the output voltage. The MODE trace starts with the PFM mode and transitions into PWM mode before returning to PFM mode.

CONCLUSION

Unless the context clearly requires otherwise, throughout the description and the claims, the words “comprise,” “comprising,” and the like are to be construed in an inclusive sense, as opposed to an exclusive or exhaustive sense; that is to say, in the sense of “including, but not limited to.” As used herein, the terms “connected,” “coupled,” or any variant thereof, means any connection or coupling, either direct or indirect, between two or more elements; the coupling of connection between the elements can be physical, logical, or a combination thereof.

Additionally, the words “herein,” “above,” “below,” and words of similar import, when used in this application, shall refer to this application as a whole and not to any particular portions of this application. Where the context permits, words in the above Detailed Description using the singular or plural number may also include the plural or singular number respectively. The word “or,” in reference to a list of two or more items, covers all of the following interpretations of the word: any of the items in the list, all of the items in the list, and any combination of the items in the list.

The above detailed description of embodiments of the invention is not intended to be exhaustive or to limit the invention to the precise form disclosed above. While specific embodiments of, and examples for, the invention are described above for illustrative purposes, various equivalent modifications are possible within the scope of the invention, as those skilled in the relevant art will recognize.

The teachings of the invention provided herein can be applied to other systems, not necessarily the system described above. The elements and acts of the various embodiments described above can be combined to provide further embodiments.

Changes can be made to the invention in light of the above Detailed Description. While the above description describes certain embodiments of the invention, and describes the best mode contemplated, no matter how detailed the above appears in text, the invention can be practiced in many ways. Details of the compensation system described above may vary considerably in its implementation details, while still being encompassed by the invention disclosed herein.

As noted above, particular terminology used when describing certain features or aspects of the invention should not be taken to imply that the terminology is being redefined herein to be restricted to any specific characteristics, features, or aspects of the invention with which that terminology is associated. In general, the terms used in the following claims should not be construed to limit the invention to the specific embodiments disclosed in the specification, unless the above Detailed Description section explicitly defines such terms. Accordingly, the actual scope of the invention encompasses not only the disclosed embodiments, but also all equivalent ways of practicing or implementing the invention under the claims.

While certain aspects of the invention are presented below in certain claim forms, the inventors contemplate the various aspects of the invention in any number of claim forms. Accordingly, the inventors reserve the right to add additional claims after filing the application to pursue such additional claim forms for other aspects of the invention.

Claims

1. A switching-mode voltage regulator integrated circuit, comprising:

a switch circuit; and
a controller, responsive to a feedback signal from the output of said switch circuit, operable to generate a control signal that toggles said switch circuit on and off so as to generate a pulse signal, wherein said controller compares a load current with a reference current, when said load current is strictly less than said reference current for a consecutive predetermined number of clock cycles, said controller toggles said switch circuit by varying the frequency of said control signal; otherwise, said controller toggles said switch circuit by varying the pulse width of said control signal.

2. The switching-mode voltage regulator integrated circuit of claim 1 wherein said controller varies the frequency of said control signal by changing the off-time and maintaining a constant on-time of said control signal.

3. The switching-mode voltage regulator integrated circuit of claim 1 further comprising:

an LC filter circuit further comprising an inductor electrically coupled in series to a capacitor, said LC filter electrically coupled to receive said pulse signal from said switch circuit so as to generate a regulated output signal; and
a diode electrically coupled to said LC filter and to an electrical ground.

4. The switching regulator integrated circuit of claim 1 wherein said switch circuit further comprises a Metal Oxide Semiconductor (MOS) transistor.

5. The switching-mode voltage regulator integrated circuit of claim 1 wherein said controller further comprises:

a gate driver circuit electrically coupled to generate said control signal that drives said switch circuit;
a pulse frequency modulation (PFM) circuit, electrically coupled to said gate driver circuit, operable to generate said control signal that has a varying frequency; and
a pulse width modulation (PWM) circuit, electrically coupled to said gate driver circuit, operable generate said control signal that has a varying pulse width.

6. The switching-mode voltage regulator integrated circuit of claim 3 wherein said controller further comprises:

a mode selector circuit; and
a flip flop circuit, electrically coupled to said mode selector circuit, operable to generate an output signal to control said mode selector circuit, wherein when said output signal is a logic HIGH, said mode selector circuit selects said PFM circuit, when said output signal is a logic LOW, said mode selector circuit selects said PWM circuit.

7. The switching-mode voltage regulator integrated circuit of claim 6 wherein said controller further comprises:

an oscillator that generates a clock signal;
an N-bit counter, electrically coupled to said oscillator and to said mode selector circuit, operable to set said flip flop circuit to a logic HIGH output signal only when said load current is strictly less than said reference current for a predetermined number of clock cycles.

8. The switching-mode voltage regulator integrated circuit of claim 6 wherein said controller further comprises a timer circuit, electrically coupled to said flip-flop circuit, operable to set or reset said flip-flop circuit.

9. The switching-mode voltage regulator integrated circuit claim 6 wherein said mode selector circuit further comprises a single-pole double-throw (SPDT) switch.

10. The switch regulating integrated circuit of claim 6 wherein said mode selector circuit further comprises a multiplexer circuit.

11. The switching-mode voltage regulator integrated circuit of claim 6 wherein said controller circuit further comprises a comparator circuit, whose input terminals electrically coupled to said feedback signal and to said reference voltage, operable to control said timer circuit and said PFM circuit.

12. The switching-mode voltage regulator integrated circuit of claim 1 wherein said controller further comprises an error amplifier, whose input terminals are electrically coupled to said reference voltage and to said feedback signal, operable to control said PWM circuit.

13. A method of controlling a switching-mode voltage regulator integrated circuit having a controller electrically coupled to drive a Metal Oxide Semiconductor (MOS) transistor, and an LC output filer, comprising:

sensing a load current from the output of said MOS transistor switch;
comparing said load current with a reference current;
if said load current is strictly less than said reference current for a consecutive predetermined number of clock cycles, then, responsive to a feedback voltage received from the output of said LC output filter, driving said MOS transistor switch using a control signal that is varied in frequency; otherwise, driving said MOS transistor switch using said control signal having varying pulse width.

14. The method of claim 13 wherein said sensing load current further comprises sensing a peak value of said load current by coupling a sense resistor to a Metal Oxide Semiconductor (MOS) transistor switch.

15. The method of claim 13 wherein said comparing said load current with a reference current further comprises:

converting an output current from said Metal Oxide Semiconductor (MOS) transistor switch into a voltage signal; and
comparing said voltage signal with a reference voltage.

16. The method of claim 13 wherein said driving said MOS transistor switch that is varied in frequency further comprises:

changing the off-time of said control signal; and
maintaining a constant pulse width of said control signal.

17. The method of claim 13 wherein said driving said MOS transistor switch that is varied in frequency further comprises:

when said feedback voltage is greater than a reference voltage, stop changing the frequency of said control signal and start varying the pulse width of said control signal.

18. The method of claim 13 wherein said driving said MOS transistor switch that is varied in frequency further comprises reducing a quiescent current in said switch-mode voltage regulator when a feedback voltage decreases to a reference voltage.

19. The method of claim 13 said driving said MOS transistor switch that is varied in frequency further comprises varying the off-time of said control signal in accordance with a peak value of said load current.

20. The method of claim 13 wherein said driving said MOS transistor switch using a control signal that is varied in pulse width further comprises when a duty cycle is greater than a predetermined percentage number, slope compensating said load current by adding a ramp signal into said load current.

21. The method of claim 13 wherein said regulating said input voltage using a PWM mode further comprises changing the pulse width of said control signal in accordance with the strength of said feedback signal.

Referenced Cited
U.S. Patent Documents
5568044 October 22, 1996 Bittner
5747977 May 5, 1998 Hwang
6713995 March 30, 2004 Chen
7030596 April 18, 2006 Salerno et al.
7061213 June 13, 2006 Yoshida et al.
7116089 October 3, 2006 Nguyen et al.
7173404 February 6, 2007 Wu
7245113 July 17, 2007 Chen et al.
7315153 January 1, 2008 Tazawa et al.
Patent History
Patent number: 7439720
Type: Grant
Filed: Aug 25, 2006
Date of Patent: Oct 21, 2008
Patent Publication Number: 20070096710
Assignee: Monolithic Power Systems (San Jose, CA)
Inventors: James H. Nguyen (San Jose, CA), Thomas T. Chiang (San Jose, CA), Christopher T. Falvey (Fremont, CA)
Primary Examiner: Gary L Laxton
Attorney: Perkins Coie LLP
Application Number: 11/509,908
Classifications
Current U.S. Class: With Threshold Detection (323/284); Using Choke And Switch Across Source (323/222)
International Classification: G05F 1/575 (20060101); G05F 1/62 (20060101);