PULSE-SHAPED ORTHOGONAL FREQUENCY DIVISION MULTIPLEXING
A method and apparatus for performing pulse shaping using different windowing functions for different sub-bands of a transmission is disclosed. A method for use in a wireless transmit/receive unit (WTRU) may include the WTRU receiving data symbols. The WTRU may assign the data symbols to a plurality of subcarriers in different sub-bands and map the data symbols on each of the plurality of subcarriers in the different sub-bands to a plurality of corresponding subcarriers of an inverse fast Fourier transform (IFFT) block. The WTRU may take an IFFT of the block for each sub-band and pad an output of the IFFT block with a prefix and a postfix for each sub-band. The WTRU may apply a windowing function to an output of the padding for each sub-band and form a composite signal for transmission by adding an output of the windowing of each sub-band. The WTRU may transmit the signal.
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This application is a continuation of U.S. patent application Ser. No. 14/766,040 filed on Aug. 5, 2015 which issued as U.S. Pat. No. 10,523,475 on Dec. 31, 2019, which is the U.S. National Stage, under 35 U.S.C. § 371, of International Patent Application No. PCT/US2014/014717, filed on Feb. 4, 2014, which claims the benefit of U.S. Provisional Application No. 61/871,461 filed on Aug. 29, 2013 and U.S. Provisional Application No. 61/760,938 filed on Feb. 5, 2013, the contents of which are hereby incorporated by reference herein.
BACKGROUNDMulticarrier modulation (MCM) is based on the idea of splitting a high-rate wideband signal into multiple lower-rate signals, where each signal occupies a narrower band. Orthogonal frequency division multiplexing (OFDM) has proved itself as one of the most popular MCM techniques and is currently used in many wireless communication systems such as 3rd Generation Partnership Project (3GPP) Long Term Evolution (LTE), 802.11, etc. OFDM offers many advantages such as robustness to multipath propagation, simple equalization, a simple transceiver architecture and efficient use of the available bandwidth through overlapping subchannels. On the other hand, OFDM has several disadvantages such as spectral leakage due to high sidelobes, and high peak-to-average power ratio (PAPR).
The demand for higher data rates has been increasing significantly. Several techniques have been studied and proposed to meet this demand, such as overlaying small cells over macro cells to allow spectral reuse, opening new bands to wireless communication, and utilizing the bandwidth more efficiently by spectrum sharing via cognitive radio. Since wireless systems are evolving towards a “network of networks” architecture where many networks are expected to share the spectrum, spectrally agile waveforms with small out-of-band leakage are important. To that end, the adjacent channel interference created by the spectral leakage of OFDM makes this waveform unsuitable for these networks.
As an alternative to OFDM, filter bank multicarrier (FBMC) modulation schemes, specifically OFDM-Offset quadrature amplitude modulation (QAM), have recently taken interest. OFDM-OQAM is another MCM technique where data on each sub-carrier is shaped with an appropriately designed pulse so that sidelobes are lower. A real data symbol is transmitted in each subchannel and on each OFDM-OQAM symbol. Consecutive OFDM-OQAM symbols are staggered. Adjacent subchannels overlap to maximize the spectral efficiency, creating inter-carrier interference (ICI); and consecutive OFDM-OQAM symbols interfere with each other due to the long pulse, creating intersymbol interference (ISI). In an ideal single path Additive White Gaussian Noise (AWGN) channel, perfect orthogonality may be achieved and ISI/intercarrier interference (ICI) may be cancelled. The OFDM-OQAM transmitter and receiver may be implemented in an efficient manner by using the polyphase filterbanks. Although OFDM-OQAM offers less spectral leakage, its implementation in practical systems poses several challenges due to its complexity, latency, and more complex channel estimation and equalization algorithms in doubly dispersive channels. Therefore, it is desirable to design an OFDM-like, but spectral contained waveform with improved out-of-band emission characteristics.
Therefore, there is a need for an advanced waveform for spectral agile systems that is capable of sharing opportunistically available and non-contiguous spectrum resources with other users. The characteristics of such a waveform should include low out-of-band emission (OOBE), low in-band distortion, low complexity, low latency, low PAPR, robustness to frequency and timing asynchronous, and robustness to power amplifier (PA) nonlinearity. The existing baseband waveforms in those systems possess very large OOBE, which may make it difficult for the existing baseband waveforms to be used in spectral agile systems.
SUMMARYMethods and apparatus for performing transmitter and receiver side pulse shaping using different windowing functions for different sub-bands of a transmission are disclosed. A method for use in a wireless transmit/receive unit (WTRU) for performing transmitter side pulse shaping may include the WTRU receiving data symbols. The WTRU may assign the data symbols to a plurality of subcarriers in the different sub-bands and map the data symbols on each of the plurality of subcarriers in the different sub-bands to a plurality of corresponding subcarriers of an inverse fast Fourier transform (IFFT) block. The WTRU may take an IFFT of the block for each sub-band and pad an output of the IFFT block with a cyclic prefix (CP) and a postfix for each sub-band. The WTRU may apply a windowing function to an output of the padding for each sub-band and form a composite signal for transmission by adding an output of the windowing of each sub-band. The WTRU may transmit the signal.
A method for use in a wireless transmit/receive unit (WTRU) for performing receiver side pulse shaping may include a WTRU receiving a signal comprising data symbols and assigning the data symbols to a plurality of subcarriers in the different sub-bands. The WTRU may apply a receive windowing function to each sub-band and map the data symbols on each of the plurality of subcarriers in the different sub-bands to a plurality of corresponding subcarriers of a fast Fourier transform (FFT) block. The WTRU may take an FFT of the block for each sub-band and apply further processing to an output of the FFT block for each sub-band.
Methods and apparatus for performing transmitter and receiver side pulse shaping with zero-padded OFDM instead of OFDM with cyclic prefix (CP) are also disclosed. Methods and apparatus for improving performance of receive windowing, including interference cancellation, CP overhead reduction, and the utilization of CP samples, are also disclosed.
A more detailed understanding may be had from the following description, given by way of example in conjunction with the accompanying drawings wherein:
As shown in
The communications systems 100 may also include a base station 114a and a base station 114b. Each of the base stations 114a, 114b may be any type of device configured to wirelessly interface with at least one of the WTRUs 102a, 102b, 102c, 102d to facilitate access to one or more communication networks, such as the core network 106, the Internet 110, and/or the other networks 112. By way of example, the base stations 114a, 114b may be a base transceiver station (BTS), a Node-B, an eNode B, a Home Node B, a Home eNode B, a site controller, an access point (AP), a wireless router, and the like. While the base stations 114a, 114b are each depicted as a single element, it will be appreciated that the base stations 114a, 114b may include any number of interconnected base stations and/or network elements.
The base station 114a may be part of the RAN 104, which may also include other base stations and/or network elements (not shown), such as a base station controller (BSC), a radio network controller (RNC), relay nodes, etc. The base station 114a and/or the base station 114b may be configured to transmit and/or receive wireless signals within a particular geographic region, which may be referred to as a cell (not shown). The cell may further be divided into cell sectors. For example, the cell associated with the base station 114a may be divided into three sectors. Thus, in one embodiment, the base station 114a may include three transceivers, i.e., one for each sector of the cell. In another embodiment, the base station 114a may employ multiple-input multiple-output (MIMO) technology and, therefore, may utilize multiple transceivers for each sector of the cell.
The base stations 114a, 114b may communicate with one or more of the WTRUs 102a, 102b, 102c, 102d over an air interface 116, which may be any suitable wireless communication link (e.g., radio frequency (RF), microwave, infrared (IR), ultraviolet (UV), visible light, etc.). The air interface 116 may be established using any suitable radio access technology (RAT).
More specifically, as noted above, the communications system 100 may be a multiple access system and may employ one or more channel access schemes, such as CDMA, TDMA, FDMA, OFDMA, SC-FDMA, and the like. For example, the base station 114a in the RAN 104 and the WTRUs 102a, 102b, 102c may implement a radio technology such as Universal Mobile Telecommunications System (UMTS) Terrestrial Radio Access (UTRA), which may establish the air interface 116 using wideband CDMA (WCDMA). WCDMA may include communication protocols such as High-Speed Packet Access (HSPA) and/or Evolved HSPA (HSPA+). HSPA may include High-Speed Downlink Packet Access (HSDPA) and/or High-Speed Uplink Packet Access (HSUPA).
In another embodiment, the base station 114a and the WTRUs 102a, 102b, 102c may implement a radio technology such as Evolved UMTS Terrestrial Radio Access (E-UTRA), which may establish the air interface 116 using Long Term Evolution (LTE) and/or LTE-Advanced (LTE-A).
In other embodiments, the base station 114a and the WTRUs 102a, 102b, 102c may implement radio technologies such as IEEE 802.16 (i.e., Worldwide Interoperability for Microwave Access (WiMAX)), CDMA2000, CDMA2000 1×, CDMA2000 EV-DO, Interim Standard 2000 (IS-2000), Interim Standard 95 (IS-95), Interim Standard 856 (IS-856), Global System for Mobile communications (GSM), Enhanced Data rates for GSM Evolution (EDGE), GSM EDGE (GERAN), and the like.
The base station 114b in
The RAN 104 may be in communication with the core network 106, which may be any type of network configured to provide voice, data, applications, and/or voice over internet protocol (VoIP) services to one or more of the WTRUs 102a, 102b, 102c, 102d. For example, the core network 106 may provide call control, billing services, mobile location-based services, pre-paid calling, Internet connectivity, video distribution, etc., and/or perform high-level security functions, such as user authentication. Although not shown in
The core network 106 may also serve as a gateway for the WTRUs 102a, 102b, 102c, 102d to access the PSTN 108, the Internet 110, and/or other networks 112. The PSTN 108 may include circuit-switched telephone networks that provide plain old telephone service (POTS). The Internet 110 may include a global system of interconnected computer networks and devices that use common communication protocols, such as the transmission control protocol (TCP), user datagram protocol (UDP) and the internet protocol (IP) in the TCP/IP internet protocol suite. The networks 112 may include wired or wireless communications networks owned and/or operated by other service providers. For example, the networks 112 may include another core network connected to one or more RANs, which may employ the same RAT as the RAN 104 or a different RAT.
Some or all of the WTRUs 102a, 102b, 102c, 102d in the communications system 100 may include multi-mode capabilities, i.e., the WTRUs 102a, 102b, 102c, 102d may include multiple transceivers for communicating with different wireless networks over different wireless links. For example, the WTRU 102c shown in
The processor 118 may be a general purpose processor, a special purpose processor, a conventional processor, a digital signal processor (DSP), a plurality of microprocessors, one or more microprocessors in association with a DSP core, a controller, a microcontroller, Application Specific Integrated Circuits (ASICs), Field Programmable Gate Array (FPGAs) circuits, any other type of integrated circuit (IC), a state machine, and the like. The processor 118 may perform signal coding, data processing, power control, input/output processing, and/or any other functionality that enables the WTRU 102 to operate in a wireless environment. The processor 118 may be coupled to the transceiver 120, which may be coupled to the transmit/receive element 122. While
The transmit/receive element 122 may be configured to transmit signals to, or receive signals from, a base station (e.g., the base station 114a) over the air interface 116. For example, in one embodiment, the transmit/receive element 122 may be an antenna configured to transmit and/or receive RF signals. In another embodiment, the transmit/receive element 122 may be an emitter/detector configured to transmit and/or receive IR, UV, or visible light signals, for example. In yet another embodiment, the transmit/receive element 122 may be configured to transmit and receive both RF and light signals. It will be appreciated that the transmit/receive element 122 may be configured to transmit and/or receive any combination of wireless signals.
In addition, although the transmit/receive element 122 is depicted in
The transceiver 120 may be configured to modulate the signals that are to be transmitted by the transmit/receive element 122 and to demodulate the signals that are received by the transmit/receive element 122. As noted above, the WTRU 102 may have multi-mode capabilities. Thus, the transceiver 120 may include multiple transceivers for enabling the WTRU 102 to communicate via multiple RATs, such as UTRA and IEEE 802.11, for example.
The processor 118 of the WTRU 102 may be coupled to, and may receive user input data from, the speaker/microphone 124, the keypad 126, and/or the display/touchpad 128 (e.g., a liquid crystal display (LCD) display unit or organic light-emitting diode (OLED) display unit). The processor 118 may also output user data to the speaker/microphone 124, the keypad 126, and/or the display/touchpad 128. In addition, the processor 118 may access information from, and store data in, any type of suitable memory, such as the non-removable memory 130 and/or the removable memory 132. The non-removable memory 130 may include random-access memory (RAM), read-only memory (ROM), a hard disk, or any other type of memory storage device. The removable memory 132 may include a subscriber identity module (SIM) card, a memory stick, a secure digital (SD) memory card, and the like. In other embodiments, the processor 118 may access information from, and store data in, memory that is not physically located on the WTRU 102, such as on a server or a home computer (not shown).
The processor 118 may receive power from the power source 134, and may be configured to distribute and/or control the power to the other components in the WTRU 102. The power source 134 may be any suitable device for powering the WTRU 102. For example, the power source 134 may include one or more dry cell batteries (e.g., nickel-cadmium (NiCd), nickel-zinc (NiZn), nickel metal hydride (NiMH), lithium-ion (Li-ion), etc.), solar cells, fuel cells, and the like.
The processor 118 may also be coupled to the GPS chipset 136, which may be configured to provide location information (e.g., longitude and latitude) regarding the current location of the WTRU 102. In addition to, or in lieu of, the information from the GPS chipset 136, the WTRU 102 may receive location information over the air interface 116 from a base station (e.g., base stations 114a, 114b) and/or determine its location based on the timing of the signals being received from two or more nearby base stations. It will be appreciated that the WTRU 102 may acquire location information by way of any suitable location-determination method while remaining consistent with an embodiment.
The processor 118 may further be coupled to other peripherals 138, which may include one or more software and/or hardware modules that provide additional features, functionality and/or wired or wireless connectivity. For example, the peripherals 138 may include an accelerometer, an e-compass, a satellite transceiver, a digital camera (for photographs or video), a universal serial bus (USB) port, a vibration device, a television transceiver, a hands free headset, a Bluetooth® module, a frequency modulated (FM) radio unit, a digital music player, a media player, a video game player module, an Internet browser, and the like.
The RAN 104 may include eNode-Bs 140a, 140b, 140c, though it will be appreciated that the RAN 104 may include any number of eNode-Bs while remaining consistent with an embodiment. The eNode-Bs 140a, 140b, 140c may each include one or more transceivers for communicating with the WTRUs 102a, 102b, 102c over the air interface 116. In one embodiment, the eNode-Bs 140a, 140b, 140c may implement MIMO technology. Thus, the eNode-B 140a, for example, may use multiple antennas to transmit wireless signals to, and receive wireless signals from, the WTRU 102a.
Each of the eNode-Bs 140a, 140b, 140c may be associated with a particular cell (not shown) and may be configured to handle radio resource management decisions, handover decisions, scheduling of users in the uplink and/or downlink, and the like. As shown in
The core network 106 shown in
The MME 142 may be connected to each of the eNode-Bs 140a, 140b, 140c in the RAN 104 via an Si interface and may serve as a control node. For example, the MME 142 may be responsible for authenticating users of the WTRUs 102a, 102b, 102c, bearer activation/deactivation, selecting a particular serving gateway during an initial attach of the WTRUs 102a, 102b, 102c, and the like. The MME 142 may also provide a control plane function for switching between the RAN 104 and other RANs (not shown) that employ other radio technologies, such as GSM or WCDMA.
The serving gateway 144 may be connected to each of the eNode Bs 140a, 140b, 140c in the RAN 104 via the Si interface. The serving gateway 144 may generally route and forward user data packets to/from the WTRUs 102a, 102b, 102c. The serving gateway 144 may also perform other functions, such as anchoring user planes during inter-eNode B handovers, triggering paging when downlink data is available for the WTRUs 102a, 102b, 102c, managing and storing contexts of the WTRUs 102a, 102b, 102c, and the like.
The serving gateway 144 may also be connected to the PDN gateway 146, which may provide the WTRUs 102a, 102b, 102c with access to packet-switched networks, such as the Internet 110, to facilitate communications between the WTRUs 102a, 102b, 102c and IP-enabled devices.
The core network 106 may facilitate communications with other networks. For example, the core network 106 may provide the WTRUs 102a, 102b, 102c with access to circuit-switched networks, such as the PSTN 108, to facilitate communications between the WTRUs 102a, 102b, 102c and traditional land-line communications devices. For example, the core network 106 may include, or may communicate with, an IP gateway (e.g., an IP multimedia subsystem (IMS) server) that serves as an interface between the core network 106 and the PSTN 108. In addition, the core network 106 may provide the WTRUs 102a, 102b, 102c with access to the networks 112, which may include other wired or wireless networks that are owned and/or operated by other service providers.
One way to improve the spectral containment of OFDM may be by filtering the time domain signal at the output of the OFDM modulator. In a fragmented spectrum where available sub-bands are not contiguous, filtering becomes challenging since a separate filter may need to be designed and used for each fragment.
Another method used to improve the spectral containment of OFDM is pulse shaping, also known as windowing. It should be noted that the terms pulse shaping and windowing may be used interchangeably throughout this description and are meant to have the same connotation. Pulse shaping is a method used to reduce the spectral leakage at the transmitter. Pulse shaping may also be used to reject adjacent channel interference at the receiver. In this technique, the rectangular pulse shape of the OFDM symbol is smoothed to prevent sharp transitions between consecutive OFDM symbols, resulting in lower sidelobes. A mechanism is deployed at the receiver to reject the adjacent channel interference leakage. This is because even if the interfering signal in the adjacent band has low out-of-band emission, the spectral leakage from the interfering signal increases after cyclic prefix (CP) removal if the received filter covers the whole accessible band. Therefore, before CP is removed, the received signal should be filtered for individual sub-bands.
Similar to transmitter filtering, receive filtering imposes challenges in fragmented spectrum. Receive windowing has been used to reduce the impact of ICI due to carrier frequency offset or Doppler and to suppress radio frequency interference (RFI) in discrete multitone (DMT) systems.
In current OFDM-based communications systems, e.g., LTE, a CP may be used in OFDM to mitigate ISI due to multipath channel or timing offset distortion. CP may be prepended at the output of the inverse fast Fourier transform (IFFT) at the transmitter side and discarded at the receiver side before the fast Fourier transform (FFT). The overhead due to the CP may be significant. Therefore, reducing this overhead while not degrading the system performance is beneficial. This is because some WTRUs may be in different locations in a cell and experience different delay spreads, and some WTRUs may need shorter CP than others. This is true for both downlink and uplink transmission.
In addition, CP in OFDM carries useful information since it is a replica of the time domain samples at the tail of an OFDM symbol. CP may also be used at single carrier systems and again consists of the time domain samples at the tail of the symbol. As noted previously, CP is discarded at the receiver since it is contaminated by ISI. However, most of the time, the channel delay spread is smaller than the length of the CP, resulting in some of the CP samples being free of ISI. These samples may be used at the receiver to improve performance.
A transceiver architecture based on transmit and receive pulse shaping to reduce spectral leakage and reject adjacent channel interference in multicarrier modulation systems; methods and apparatus for transmitter side implementation of pulse shaping on time domain samples of each symbol for OFDM based systems; methods and apparatus for receiver side implementation of windowing on time domain samples of each symbol for OFDM based systems; and methods for applying windowing over a plurality of received OFDM data block are described herein.
A general MCM scheme will now be described. For a general MCM scheme, the input data sequence to be transmitted on the k′th subcarrier and ′th symbol may be denoted as Sk[], where k denotes the subcarrier index and denotes the symbol index. Then, the input data symbols for the k′th subcarrier may be written as
xk(t)=Sk[]δ[t−T′], Equation (1)
where T′ is the symbol interval. The data on each subcarrier may be convolved by a filter p(t) that is modulated to the frequency of that subcarrier. The aggregate transmitted signal may be written as
x(t)=Σk=0M-1[{Sk[]δ[t−T′]}*{p(t)ej2πkF
where M is the total number of subcarriers and Fs is the spacing between the subcarriers.
is typically equal to or smaller than T′.
After expanding the convolution in Equation 2, the following may be obtained:
x(t)=Σk=0M-1Στ=−∞∞Sk[]δ(τ−T′)p(t−τ)ej2πkF
Since δ(τ−T′)=0, τ≠T′, the following may be obtained:
x(t)=Σk=0M-1Sk[]p(t−−T′). Equation (4)
Equation (4) may be viewed as the general multicarrier modulation scheme.
An example of the OFDM structure will now be described. With respect to OFDM, it may be assumed that the signal in Equation (4) is sampled at a sampling rate of Ts=T/N and that T′=λTs. Here, the discrete-time of Equation (4) may be written as:
where n denotes the time sample index.
For OFDM without CP with critical sampling, the parameters in Equation (5) are as follows:
With these parameters, Equation (5) may be written as
Since the consecutive symbols do not overlap, only a single ′th
OFDM symbol may be considered:
From Equation (7), the ′th OFDM symbol may be computed by taking the inverse fast Fourier transform (IFFT) of the input data symbols.
When a CP is appended, the pulse shape may be defined as:
where NG is the number of samples in the guard interval, e.g., cyclic prefix. Again, the consecutive OFDM symbols may not overlap, so it may be sufficient to consider a single OFDM symbol. The parameters in Equation (5) may be as follows:
Defining n=(N+NG)+m, where m=−NG, . . . , 0, 1, . . . N−1, then:
are equal. Therefore, Equation (8) may be implemented by taking the IFFT of the input data symbols, and padding the last NG samples of the IFFT output to the front of the IFFT output.
As discussed above, pulse shaping, also known as windowing, may be used at the transmitter side and receiver side to improve the spectral containment of OFDM.
Methods and apparatus for transmitter side windowing will now be described. As an example, the pulse shaping function for windowing in Equation (5) may be defined as
where NT=N+NG, and λ=(1+β)NT. Other pulse shaping functions are also possible. Pulse shaping functions should generally create a smooth transition at the boundary of two consecutive symbols. In this case, the new guard interval is generally larger than the cyclic prefix. N′G=NG+NEGI may be defined, where NEGI is the extended guard interval. However, from a signal processing point of view, is nothing but a longer cyclic prefix. It should be noted that windowing functions other than the one in Equation (10) are also possible, but the following approach will be similar.
Defining n=i(1+β)NT+m, where m=0, . . . , (1+β)NT−1, Equation (4) may be written as
which may be written as
xi′[m]=Ek=0M-1Sk[]p[(i−)(1−β)NT+m]ej2πkF
defining x′[m]=x[i(1+β)NT+m].
Since for the i′th block, only two symbols overlap due to the pulse shape design where the pulse shape should not be much longer than N, the terms corresponding to =i and =i−1 in the summation over are retained. Then,
which is equal to
In Equation (14), it can be seen that for the first term is non-zero for m=0, 1, . . . , βNT has non-zero values. The limits where the pulse shape is defined may be seen in Equation (10). Therefore, the implementation, as shown in
Referring to
A transmitter side windowing method and apparatus using zero-padding (ZP) OFDM instead of CP will now be described.
A method and apparatus for applying different window functions to separate groups of a data signal will now be described. Different windowing functions may be applied to different sub-bands of the transmission band. As an example, the sub-bands next to the edges may be shaped with longer windows to get better spectral containment of those sub-bands. On the other hand, the sub-bands away from the edges, or in the middle of the band, may be shaped with shorter windows. Since windowing may introduce distortion, the possibly larger distortion introduced by the longer windows will be limited to the sub-bands on the edges.
Referring to
Methods and apparatus for receiver side windowing will now be described. A mechanism may be used at the receiver to reject the adjacent channel interference leakage. Such a mechanism is used because even if the interfering signal in the adjacent band has low out-of-band emission, the spectral leakage from the interfering signal increases after the CP removal. Therefore, before the CP is removed, the received signal may be filtered. OFDM achieves this by using the rectangular windowing, which corresponds to a sinc-type filter with high tails, and is therefore not satisfactory for interference rejection capability.
Similar to the transmitter side filtering, receive filtering imposes challenges in a fragmented spectrum. An alternative method is to use windowing at the receiver. In general, if the transmitter attaches a prefix and postfix as illustrated in
One way the receive window may be defined is as follows:
In general, the receive window may be defined beyond NT. However, this may call for the next symbol to be used, causing a small delay. The following approach will hold regardless.
In the case of no transmit windowing, i.e., all 1's, and no overlapping between consecutive symbols, the transmitted symbol may be written as
where λ=N+Npre+Npost. For n=lλ−Npre, . . . , lλ+(N+Npost−1),
where the Npref may include the guard interval for the CP as well. Let n=lλ+m for l=−∞, . . . , 0, . . . , ∞, m=−Npre, . . . , N+Npost. Then,
The receiver windowing coefficients may be defined as: {w[m], m=−Npre, . . . , N+Npost}. Applying windowing and converting the received signal back to the frequency domain by FFT, i.e., frequency demodulation at frequency k/N,
If m′ is defined as follows, m′=N+m for m=Npre, . . . , −1, then, the terms in Equation 18 may be written as follows:
Expanding w[m] such that w[m]=0 for m=−N, . . . , −Nprep−1. Then the first term becomes:
Expanding w[m] such that w[m]=0 for m=N+Npost, . . . , 2N. Then,
Combining the terms, gives
To recover the transmitted symbols, make xl[m]=xl[m−N] for m=N−Npre, . . . , N−1 and xl[m]=xl[m+N] for m=0, . . . , Npost−1.
The transmitted signal may be considered as
where λ=N+Npost+Npre. Assuming that p[n]≠0 for n=−Npre, . . . , N+Npost, define m=n−lλ. Then, taking samples from x[n]:
For m=−Npre, . . . , N+Npost. Note that y[m]=y[m−N] for m=N−Npre, . . . , N−1 and y[m]=y[m+N] for m=0, . . . , Npost−1. At the receiver side, the transmitter window p[m] should be chosen such that xl[m]=p[m]y[m] also satisfies this condition.
Applying the transmitted signal at the receiver side,
or a constant, for m=0, . . . , N−1, the above expression is Sk′[l].
As an alternative method, the received signal on subcarrier {circumflex over (k)} after windowing and taking the FFT may be written as
where the receive window is denoted as w[n].
Defining n=iN+m, and ignoring p[n] because it is all 1's, Equation 29 is obtained using only i=−1, 0, 1 because of the extent of the window.
From Equation 29, it can be seen that, to ensure orthogonality, Σi=−11w[iN+m]=constant. Under this condition, orthogonality is preserved, and
[{circumflex over (k)}]=Σk=0N-1Σm=0N-1sk[]ej2π(k−{circumflex over (k)})m Equation (30)
Note that, similar to adding the CP in Equation 8, the extension of the OFDM signal over duration N is equal to taking the IFFT and adding the first samples as postfix.
From Equation (29),
The above window is the most general case. In practical systems, by way of example, the postfix of one symbol serves as the prefix of the next symbol. Therefore, applying windowing to the postfix introduces additional ISI from the following symbol and increases latency.
For this case,
For orthogonality, Σi=−10 w[iN+m]=constant.
With transmit windowing, it can be shown that to maintain orthogonality,
Σi=−10w[iN+m]p[m+iN]=constant. Equation (33)
Receive windowing may also be implemented by point-wise multiplying the input received block by the receiver windowing coefficients.
Equations (27) and (33) specify the conditions to maintain orthogonality at the receiver when both transmit and receive windowing are applied. Given a transmit window function, the receiver window function may be computed from these equations.
Intersymbol interference due to receive windowing will now be described. The following analysis assumes that the second type of receive windowing, as described in
Assume that the transmitted signal goes through a channel denoted as h[n]. The received signal, after windowing, may be written as
[{circumflex over (k)}]=Σi=−10Σm=0N-1w[iN+m]r[iN+m]e−j2π{circumflex over (k)}F
where r[n]=x[n]*h[n]=Σu=0L-1x(n−u)h(u) Then,
[{circumflex over (k)}]=Σi=−10Σm=0N-1w[iN+m]Σu=0L-1x(iN+m−u)h(u)e−j2π{circumflex over (k)}F
Note that, the prefix part of the data block may contain interference from the previous data block. This interference is multiplied by the window function and added to the desired signal as shown in Equation (23). The level of interference may depend on the channel delay spread and the length of the window function. If the zero-part of the window function is long enough to absorb the ISI, interference may not occur since the prefix is discarded. However, if the delay spread is long enough, then interference may occur. Assuming that the channel delay spread L<N, then ISI is due to only the previous data block. In this case, interference contributes from i=−1, due to data transmitted in i=−2. Then, the interference on the k′th subcarrier may be written as
[{circumflex over (k)}]=Σm=0N-1w[iN+m]Σu=0L-1x(iN+m−u)v(iN+m−u)h(u)e−j2π{circumflex over (k)}F
where v(iN+m−u)=1 for iN+m−u<−N, else 0.
From here, the interference power may be computed. Matrix notation may be used to see the ISI more clearly. The transmitted signal may be written as
where FH is the IFFT matrix and appends a prefix of β samples. The received signal after the channel is
where the first part of Equation (27) is the desired signal, the second part of Equation (27) is the ISI from the previous block, and the third part of Equation (27) is the noise. The channel matrices in Equation (27) may be written as
After applying the windowing to the received signal,
where W performs the windowing operation and is defined as follows:
takes the first β values of the processed data block and adds to the last β values. These two matrices together perform the receive windowing. Equation 29 may be rewritten as
After the removal of the guard band, the signal part preserves the orthogonality. However, ISI from the previous block is introduced. The amount of the ISI depends on L and β. The effect of ISI may be more clearly seen in the following:
The ISI introduced will be Fyisi, where
yisi=[0 0 0 0 . . . w−βg(0) . . . w−1g(β−1)]T. Equation (44)
The samples multiplied by the window's 0-coefficients will not contribute to the ISI. If w−β+α is the first non-zero sample of the window, then ISI contribution will be zero if α≥L.
Methods to improve performance of receive windowing, such as interference cancellation will now be described. Successive interference cancellation will now be described. The received signal may contain interference from the previous transmitted block due to the multi-path channel. Equation (44) characterizes the interference in terms of the channel, the previous block, and the receive windowing coefficients. After windowing, depending on the CP length and the windowing coefficients, some of this interference is added to the time samples of the current symbol. One method to improve the performance is to regenerate and cancel this interference. Assuming that the channel is known and the previous symbol has been demodulated, the interference may be regenerated and subtracted from the current symbol. The subtraction may be done either in the time domain or the frequency domain.
If the transmitted signal was also windowed, then even without multi-path channel, the receive windowing introduces ISI.
Referring to
Referring to
To understand the cases shown in
The first term of Equation (45) may be described as follows. The data block goes through IFFT. To perform transmit windowing, the last β symbols are copied as the prefix, and the first α symbols are copied as the postfix. Note that, since overlapping of the prefix and postfix of consecutive symbols will be performed, α and β are preferred to be equal. Then, the signal is multiplied by the window coefficients, denoted as the diagonal matrix P, whose diagonal elements are the window coefficients. This transmitted signal goes through the channel.
The second term of Equation (45) is similar to the first term of Equation (45) and contains the coefficients from the previous data block.
The third term of Equation (45) may be the intersymbol interference from the previous symbol. The received signal is processed by the receive windowing. Depending on the window coefficients, the interference may be estimated and canceled.
Methods and apparatus for applying different windowing functions to separate groups of a received signal will now be described. Different windowing functions may be applied to different sub-bands of the transmission band. As an example, the sub-bands next to the edges may be shaped with longer windows to better reject adjacent channel interference. On the other hand, the sub-bands away from the edges may be shaped with shorter windows. Since windowing may introduce distortion, the possibly larger distortion introduced by the longer windows may be limited to the sub-bands on the edges. An example of this method is shown in
The requirements of spectral leakage and interference may change, therefore, adaptation of the windowing function may be beneficial. As discussed previously, the window function at the transmitter shapes the CP and therefore may introduce interference if the unshaped portion of the CP is not long enough to compensate for the channel delay spread. Similarly, windowing at the receiver may introduce interference. There may, however, be a tradeoff between interference and spectral leakage reduction and adjacent channel interference rejection. For example, as the roll-off portion of a window gets longer, e.g., the window gets smoother, spectral leakage reduction at the transmitter and adjacent channel interference rejection at the receiver improves. However, self-created interference, due to ISI and/or ICI may increase.
Accordingly, it may be beneficial to adaptively change the window function depending on the requirements on the spectral leakage and adjacent channel interference rejection. If the requirements are tight, then a smoother window function may be used at the transmitter and/or receiver at the expense of more ISI/ICI. Otherwise, a less smooth window function may be preferred, resulting in less ISI/ICI. The selection may be done by the receiver or transmitter based on measurements, such as sensing, or the receiver may select the receive window function while the transmitter may select the transmit window function.
In transmit windowing, the window function may be chosen such that the samples corresponding to the data block (e.g., those that are produced by the IFFT before CP attachment) are multiplied by unity (i.e., the weights of the window function are 1). The samples that correspond to the CP and postfix may be multiplied by non-unity weights. If the lengths of the CP, data block, and postfix are k, n, k, respectively, then the total length of the window may be n+2k, with the middle samples indexed by k+1 to k+n being unity. Sometimes, a tight requirement on spectral leakage may necessitate a very smooth window function to be used at the transmitter. But since the size of the CP may be fixed, this may not be possible. One method to overcome this may be to allow some of the n weights of the window function to have non-unity values. For example, Equation (47) below may be used. The smoothness of the window function may be improved significantly without bit error rate (BER) degradation.
During receiver windowing, the overall performance of the receiver may be improved. In one example, the received signal-interference-to-noise-ratio (SINR) may be improved by adding the samples corresponding to the CP to the end of the received symbol. It should be noted that if windowing is not applied, this operation may be done independently.
For example, the transmitted signal may be written in matrix notation as
where FH is the inverse fast Fourier transform (IFFT) matrix and
appends a prefix of NG samples. The received signal after the channel may be written as
where the first part of Equation (49) is the desired signal, the second part is the inter-symbol interference (ISI) from the previous block, and the third part is the noise. The channel matrices in Equation (49) may be written as
where [h0, h1, . . . , hL-1] is the channel response of the multipath channel.
In the above example, if the channel order L=NG, then all samples of the cyclic prefix may be contaminated. However, when a wireless communication system is designed, the length of the cyclic prefix, NG, may be selected based on worst case scenarios. Therefore, very often, L<NG, so NG−L samples of the cyclic prefix may be ISI-free. The initial paths of the channel may have significant power, and the delayed paths may have much less power. Thus, many samples of the cyclic prefix may only be contaminated by low-power ISI.
If sample n of the received OFDM symbol (with the CP) is ISI free, the received OFDM symbol may be expressed as y[n]=x[n]+w1[n]. Due to how the cyclic prefix is formed, sample (n+N) may be written as y[n+N]=x[n+N]+w2[n+N], where x[n]=x[n+N].
The noise samples w1 and w2 may be independent and may have the same statistics. Assuming they are zero-mean and have variance δ2, if y[n] and y[n+N] are added, the result may be expressed as y[n]+y[n+N]=x[n]+x[n+N]+w1[n]+w2[n+N]=2x[n]+w1[n]+w2[n+N].
The SINR of the added symbols may be
where P0=E{xn2}. Accordingly, the power of one sample is doubled (in the time domain) due to the CP being a copy of the original samples. In the frequency domain, assuming the received signal is written as
the estimate of the transmitted signal on subcarrier k may be written as
From this, the SINR on the subcarrier k may be given as
By way of example, if it is assumed that the CP is 25% of the total symbol duration, and half of the samples are ISI-free, then the improvement in SINR per sample in the frequency domain is 0.3 dB. Assuming that there is a channel with L1 paths, where L1<NG, sample n may be written as y[n]=y[n+N]=Σi=0N
Even a small increase in SINR may result in a jump in the rate because the Modulation and Coding Scheme (MCS) works on discrete channel quality indicator (CQI) values. If a small increase results in the next MCS, the rate may be doubled in some scenarios. On the other hand, WTRUs closer to the transmitter may have most of the samples in the CP ISI-free. Further, even if a sample is not ISI-free, it may still be useful if the contribution of the desired signal outweighs the contribution of the ISI. For example, assuming that the received samples are y[n]=x[n]+w1[n]+z[n] and y[n+N]=x[n+N]+w2[n+N], where z[n] is the ISI, if y[n] and y[n+N] are added, y[n]+y[n+N]=x[n]+x[n+N]+w1[n]+w2[n+N]+z[n]=2x[n]+w1[n]+w2[n+N]+z[n]. The SINR of the added symbols in the time domain is
then it may be beneficial to use that sample from the CP.
When the samples from the CP are added to the samples at the tail of the OFDM symbol, each sample may be divided by 2 so that orthogonality is preserved. This may be due to the orthogonality condition in receive windowing.
The result in Equation (53) may be generalized to the case where a receive window is applied. After receive windowing of the received samples, the CP may be added to the end of the OFDM symbol. Assuming an additive white Gaussian noise (AWGN) channel is used (e.g., ISI has not contaminated the CP samples), the received signal may be:
where z[n] is the AWGN with zero mean and variance σ2. The estimate of the transmitted data symbol on subcarrier {circumflex over (k)} may be written, after windowing with v[n], as
From Equation (55), the SINR on the subcarrier k may be given as
Similarly, with a multi-path channel (at the absence of ISI), the SINR becomes
However, due to ISI, windowing may introduce interference, and the exact SINR may depend on the channel model.
With channel estimation, the WTRU may be able to determine the channel delay spread and distribution of the paths. This would allow the WTRU to determine if the CP contains samples that are ISI-free or are contaminated with small enough ISI. Since WTRUs closer to the transmitter may be most likely to benefit from this (i.e., CP length is larger than the delay spread), it may be assumed that they already have better a better signal-to-noise ratio (SNR) and channel estimation may be reliable.
As discussed previously, the length of the CP in current wireless systems using CP may be set to one value and may be used for all users. However, in a given transmission area, some users may be experiencing channels with smaller delay spread than others. If that is the case, it may be beneficial to use a shorter CP for those users experiencing channels with smaller delay spread.
Referring to
Use of a variable CP may cause, in general, ICI. As in the example described in
x(t)=Skpk(t)exp(j2πFkt)+Smpm(t)exp(j2πFmt). Equation (57)
At one of the receivers, the data symbol may be estimated as in Equation (58).
The first part of Equation (58) is the desired signal and the second part is the interference. Integration is performed over an interval of T, and the CP discarded. But, for the interfering signal, this T duration may cover an arbitrary portion of its CP and its data part. Assume that in this T interval, the interfering signal does not jump from one block to a new one, e.g., there is only one data symbol. As shown in Equation (59), there will not be any interference.
However, if the T interval covers two symbols, then 0.
The same may be true also for multi-path channels since each path introduces a multiplicative coefficient. If the interval T has only one data symbol of the interfering signal, then there will be no interference. So, in general, there may be interference most of the time. The symbols may drift because they have different lengths. For some symbols, orthogonality may be preserved.
A simulation is described to evaluate the interference power caused by using a variable CP. In this simulation, a transmitter similar to that described above and in
Pulse shaping, or windowing, as described heretofore is one technique that may be used to reduce the ICI at the transmitter when using variable CP lengths.
Referring to
Another technique that may be used to reduce the ICI when using variable CP lengths is filtering at the transmitter, at the receiver, or both.
Each per-RB multicarrier modulated signal 2660a, 2660b, . . . , 2660n only has a signal overlapping its adjacent RBs but not the RBs beyond its adjacent RBs. It is assumed that a per-RB transmit filter brings the signal leakage of a per-RB multicarrier modulated signal to its non-adjacent RBs to be negligible. The signal overlap between adjacent RBs may not create inter-subcarrier interference due to orthogonality between subcarriers in different RBs.
Since the transmitter will be able to use CPs of different lengths at a given time, the frame structure may need to be addressed.
One option is to keep the subframe length unchanged but to change the number of OFDM symbols depending on the CP length. This will result in a different amount of data blocks existing in different signals.
In an example, assume the transmitter is LTE-based and uses short and long CP simultaneously. The subframe length is 1 ms. Here, 14 OFDM data blocks may exist in the first signal, and 12 OFDM data blocks may exist in the second signal.
1. A method for performing pulse shaping at a transmitter, comprising:
performing an inverse fast Fourier transform (IFFT) over N symbols of a current block of data.
2. The method of embodiment 1, further comprising:
padding a prefix of the current block of data.
3. The method of embodiment 2, wherein the prefix has a predetermined length.
4. The method of any one of embodiments 1-3, further comprising: multiplying a result of the IFFT by a windowing function.
5. The method of embodiment 4, wherein the windowing function is defined by the equation:
p[m],m=0,1, . . . ,(β+1)NT−1.
6. The method of any one of embodiments 1-5, further comprising: perform a second IFFT on a previous block of data.
7. The method of embodiment 6, further comprising:
adding a prefix and a postfix to a result of the second IFFT.
8. The method of embodiment 7, further comprising:
multiplying a result of adding the prefix and the postfix to the result of the second IFFT by a second windowing function.
9. The method of embodiment 8, wherein the second windowing function is defined by the equation:
p[m],m=0,1, . . . ,(2β+1)NT−1.
10. The method of embodiment 9, further comprising:
taking a last βNT samples of the windowed signal; and
adding the last βNT samples to a first βNT samples of the output of the windowing function.
11. The method of embodiment 1, further comprising:
adding a prefix and a postfix to a result of the IFFT.
12. The method of embodiment 11, further comprising:
multiplying a result of adding the prefix and the postfix to the result of the IFFT by a windowing function.
13. The method of embodiment 12, wherein the windowing function is defined by the equation:
p[m],m=0,1, . . . ,(2β+1)NT−1.
14. The method of embodiment 13, further comprising:
retaining a last βNT samples of the windowed signal in a buffer.
15. The method of embodiment 14, further comprising:
adding the samples in the buffer to a first βNT samples of the windowed signal.
16. The method of embodiment 15, further comprising:
transmitting a first (1+β)NT samples.
17. A method for performing transmitter windowing of a signal, comprising:
mapping modulation symbols to corresponding elements of an inverse fast Fourier transform (IFFT) block.
18. The method of embodiment 17, further comprising:
taking an IFFT of the block.
19. The method of embodiment 18, further comprising:
taking a first M samples with indices; and
adding the samples to a tail of the block as a postfix.
20. The method of embodiment 19, further comprising:
adding K zeros to a head of the block as a prefix, wherein the prefix is a zero prefix.
21. The method of embodiment 20, further comprising:
point-wise multiplying the block with a windowing function.
22. The method of embodiment 21, further comprising:
discarding the zero prefix at a receiver side.
23. The method of embodiment 22, further comprising:
adding a postfix to a first head of the data block; and
discarding the postfix.
24. The method of embodiment 23, further comprising:
processing a result of the adding the postfix by a fast Fourier transform.
25. A method for performing transmitter windowing for non-contiguous sub-bands, comprising:
applying a different windowing function to different sub-bands.
26. The method of embodiment 25, wherein longer windows are applied to sub-bands adjacent to edges of a transmission band.
27. The method of embodiment 25, wherein shorter windows are applied to sub-bands distant from edges of a transmission band.
28. The method of any one of embodiments 25-27, wherein the sub-bands are non-overlapping, whereby subcarriers in a sub-band are different than subcarriers in other sub-bands.
29. The method of any one of embodiments 25-28, further comprising:
passing incoming modulated symbols through a serial to parallel processor.
30. The method of embodiment 29, further comprising:
mapping the modulated symbols to subcarriers in an inverse fast Fourier transform (IFFT) block corresponding to the subcarriers in those sub-bands.
31. The method of embodiment 30, further comprising:
padding an output of the IFFT with a prefix and a postfix for each sub-band.
32. The method of embodiment 31, further comprising:
point-wise multiplying an output of the padding with a windowing function.
33. The method of embodiment 32, further comprising:
adding an output of all branches to create a composite signal to be transmitted.
34. A method for performing receive windowing, comprising:
rejecting adjacent channel interference leakage.
35. The method of embodiment 34, wherein the rejecting includes
applying a filter to a received signal.
36. The method of embodiment 34, wherein the rejecting includes
applying a receive window to a received signal.
37. The method of embodiment 36, further comprising:
converting the received signal back to the frequency domain via a fast Fourier transform.
38. A method for performing successive interference cancellation, comprising:
regenerating and canceling interference from a received signal.
39. The method of embodiment 38, wherein the interference is regenerated from a previous symbol in the received signal and is subtracted from a current symbol in the received signal.
40. The method of embodiment 39, wherein the subtraction is performed in the time domain.
41. The method of embodiment 39, wherein the subtraction is performed in the frequency domain.
42. The method of any one of embodiments 38-41, further comprising:
performing receive windowing.
43. The method of embodiment 42, further comprising:
discarding a cyclic prefix.
44. The method of embodiments 42 or 43, further comprising:
discarding an extended guard interval.
45. The method of any one of embodiments 42-44, wherein the receive window is applied to the cyclic prefix and the extended guard interval.
46. The method of any one of embodiments 38-45, wherein different windowing functions are applied to different sub-bands.
47. The method of embodiment 46, wherein longer windows are applied to sub-bands adjacent to edges of a transmission band.
48. The method of embodiment 46, wherein shorter windows are applied to sub-bands distant from edges of a transmission band.
49. The method of any one of embodiments 38-48, further comprising:
passing modulated symbols of the received signal through a serial to parallel processor.
50. The method of embodiment 49, further comprising:
copying the received signal to M branches.
51. The method of embodiment 50, further comprising:
applying receive windowing to each of the M branches.
52. The method of embodiment 51, further comprising:
removing the prefix and the postfix from the received signal.
53. The method of embodiment 52, further comprising:
performing a fast Fourier transform on the received signal after the prefix and the postfix have been removed.
54. The method of embodiment 53, further comprising:
selecting samples corresponding to subcarriers of the sub-band in each of the M branches.
55. The method of embodiment 54, wherein the selected samples are further processed.
56. The method of embodiment 55, wherein the further processing includes any one of: equalization or demodulation.
57. A method of reducing cyclic prefix (CP) overhead, the method comprising:
dividing a bandwidth into at least two sub-bands;
generating a signal for each of the at least two sub-bands; and
attaching a CP to each of the signals for each of the at least two sub-bands, wherein a length of at least two of the CPs is different.
58. The method of embodiment 57, further comprising adding the signals for each of the at least two sub-bands with the CP attached to each of the signals to form a transmit signal.
59. The method of embodiment 58, further comprising transmitting the transmit signal.
60. The method of embodiments 58 or 59, wherein the length of at least one of the at least two of the CPs attached to each of the signals is smaller than a standard fixed length CP.
61. The method of any one of embodiments 58-60, wherein the transmit signal is one of an orthogonal frequency-division multiplexing (OFDM) signal, a discrete Fourier transform (DFT)-spread OFDM signal or a single carrier (SC) signal with CP.
62. The method of any one of embodiments 57-61, wherein the at least two sub-bands are at least one of non-contiguous subcarriers or resource blocks (RBs).
63. The method of any one of embodiments 57-62, wherein at least two sub-carriers between bands are not used.
64. The method of any one of embodiments 57-63, wherein each of the at least two sub-bands includes only contiguous sub-carriers.
65. The method of any one of embodiments 57-64, wherein each of the at least two sub-bands are divided into sub-units.
66. The method of embodiment 65, further comprising filtering each sub-unit separately.
67. The method of embodiment 66, further comprising adding signals from each sub-unit to form the transmit signal.
68. The method of any one of embodiments 65-67, wherein the sub-units are RBs.
69. The method of any one of embodiments 57-68, wherein a sub-frame length is fixed, and a number of OFDM symbols depends on the length of the CP.
70. The method of any one of embodiments 57-69, wherein different frame formats correspond to different signals.
71. The method of any one of embodiments 57-70, further comprising adaptively changing a window function depending on requirements of spectral leakage and adjacent channel interference rejection.
72. The method of any one of embodiments 57-72, wherein a window function is chosen such that at least one sample corresponding to a data block is multiplied by unity and at least one sample corresponding to a data block is multiplied by a non-unity value.
73. A method of reducing cyclic prefix (CP) overhead, the method comprising receiving a signal that includes a CP attached to each of a plurality of signals, corresponding to respective sub-bands, which were added together, wherein a length of at least two of the CPs is different.
74. The method of embodiment 73, further comprising recovering a plurality of symbols from the received signal.
75. The method of embodiment 73 or 74, further comprising sampling the received signal to provide a plurality of samples.
76. The method of embodiments 74 or 75, further comprising adding at least one sample corresponding to at least one of the CPs to the end of at least one corresponding one of the recovered plurality of symbols.
77. The method of embodiments 75 or 76, further comprising discarding at least one sample due to at least one of the CPs.
78. The method of any one of embodiments 75-77, further comprising dividing each of the plurality of samples by 2 to preserve orthogonality.
79. The method of any one of embodiments 76-78, wherein the samples corresponding to a CP are added to the end of at least one corresponding one of the recovered plurality of symbols on a condition that the SINR of the added symbols in the time domain is
80. A wireless transmit/receive unit configured to perform the method of any one of embodiments 1-79.
81. An access point configured to perform the method of any one of embodiments 1-79.
82. A Node B configured to perform the method of any one of embodiments 1-79.
83. An integrated circuit configured to perform the method of any one of embodiments 1-79.
Although features and elements are described above in particular combinations, one of ordinary skill in the art will appreciate that each feature or element can be used alone or in any combination with the other features and elements. In addition, the methods described herein may be implemented in a computer program, software, or firmware incorporated in a computer-readable medium for execution by a computer or processor. Examples of computer-readable media include electronic signals (transmitted over wired or wireless connections) and computer-readable storage media. Examples of computer-readable storage media include, but are not limited to, a read only memory (ROM), a random access memory (RAM), a register, cache memory, semiconductor memory devices, magnetic media such as internal hard disks and removable disks, magneto-optical media, and optical media such as CD-ROM disks, and digital versatile disks (DVDs). A processor in association with software may be used to implement a radio frequency transceiver for use in a WTRU, UE, terminal, base station, RNC, or any host computer.
Claims
1. (canceled)
2. A method implemented by an 802.11 device for transmitting a signal, the method comprising:
- mapping a plurality of data symbols on each of a plurality of subcarriers in a plurality of sub-bands;
- performing an IFFT on each of the plurality of sub-bands to generate an output for each of the plurality of sub-bands;
- padding the output with a prefix or a postfix for each sub-band;
- applying a windowing function to the padded output for each sub-band;
- forming a composite signal for transmission from each of the windowed padded output; and
- transmitting the composite signal.
3. The method of claim 2, wherein different windowing functions are applied to different sub-bands.
4. The method of claim 2, wherein the prefix or the postfix is a guard interval.
5. The method of claim 2, wherein the subcarriers in the sub-bands are non-contiguous.
6. The method of claim 2, wherein the prefix for each sub-band is of a different length.
7. An 802.11 device configured to transmit a signal, the WTRU comprising:
- a processor configured to: map a plurality of data symbols on each of a plurality of subcarriers in a plurality of sub-bands; perform an IFFT on each of the plurality of sub-bands to generate an output signal for each of the plurality of sub-bands; padding an output of the IFFT block with a prefix or a postfix for each sub-band; apply a windowing function to the padded output for each sub-band; form a composite signal for transmission from each of the windowed padded output; and
- a transmitter, operatively coupled to the processor, configured to transmit the composite signal.
8. The 802.11 device of claim 7, wherein different windowing functions are applied to different sub-bands.
9. The 802.11 device of claim 7, wherein the prefix or the postfix is a guard interval.
10. The 802.11 device of claim 7, wherein the subcarriers in the sub-bands are non-contiguous.
11. The 802.11 device of claim 7, wherein the prefix for each sub-band is of a different length.
Type: Application
Filed: Dec 31, 2019
Publication Date: Jul 2, 2020
Applicant: IDAC Holdings, Inc. (Wilmington, DE)
Inventors: Erdem BALA (East Meadow, NY), Rui YANG (Greenlawn, NY), Jialing LI (San Diego, CA), Daniel R. COHEN (Huntington, NY)
Application Number: 16/731,432