Detection of the conduction state of an RC-IGBT

- Infineon Technologies AG

A circuit arrangement includes: a reverse conducting IGBT configured to allow for conducting a load current in a forward direction and in a reverse direction, the IGBT having a load current path and a gate electrode; a gate control unit connected to the gate electrode and configured to activate or deactivate the IGBT by charging or, respectively, discharging the gate electrode in accordance with a gate control signal; a gate driver unit configured to detect whether the IGBT conducts current in the forward direction or the reverse direction by sensing a gate current caused by a change of a voltage drop across the load path due to a changing of the reverse conducting IGBT into its reverse conducting state, the gate control unit further configured to deactivate the IGBT or to prevent an activation of the IGBT via its gate electrode when the gate driver unit detects that the IGBT is in its reverse conducting state.

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Description
TECHNICAL FIELD

The present disclosure relates to a method for controlling a reverse conducting insulated gate bipolar transistor (RC-IGBT), particularly to a method for detecting the direction of a current passing through the RC-IGBT.

BACKGROUND

Conventional IGBTs are used, for example, in conventional inverter circuits to regulate inductive loads related to an electric machine. The conventional inverter circuits include conventional IGBTs, each connected in parallel to a free-wheeling diode to allow for a bi-directional current, which becomes necessary, if IGBTs are to be applied, for example, in connection with electric machinery implicating inductive load characteristics. The application of parallel free-wheeling diodes, however, involves numerous disadvantages, i.e.: The IGBT housing is to be designed so as to accommodate both, the IGBT body and the separate free-wheeling diode alike, and the bonding of the two components is a comparatively complex and costly matter.

To avoid these disadvantages, reverse conducting IGBTs (so called RC-IGBTs) have become popular, such as disclosed in the U.S. Patent Publication No. 2007/0231973 A1, in which an IGBT and a free-wheeling diode are monolithically formed as one simple semiconductor component. The p-doped collector zone is locally disrupted by incisions, where n-doped semiconductor material contacts the collector metallization, resulting in a so-called PIN-electrode structure between the emitter structure, the lightly doped drift zone and the p-doped material in the MOS-channel zone.

The conducting state of a conventional free-wheeling diode connected antiparallel to an IGBT does not depend on the conducting state of the IGBT, whereas the intrinsic free-wheeling diode of an RC-IGBT is influenced by the conduction state of the MOS-channel zone of the RC-IGBT. That is, if the RC-IGBT is triggered via its gate electrode during its reverse conduction state, the MOS-channel is conductive. Since the MOS-channel allows for a bi-directional current, electrons in the reverse conduction path will see an additional current path in case of a triggered gate electrode. Such, the forward voltage drop may be substantially increased in the PIN-diode structure as not all of the electrons contribute to the flooding of the PIN-diode, which is undesirable in most cases.

Publication DE 10 2009 001 029 A1 teaches a control method and a corresponding circuit structure, how to overcome the problem described hereinbefore, the control method involving a method for detecting the current direction in an RC-IGBT. The current direction is particularly detected by means of a commonly used VCEsat-detection circuitry, VCEsat representing the collector-emitter saturation voltage of the RC-IGBT. One disadvantage about the VCEsat-detection circuitry may be that at least one (or a plurality of serial) diode(s) of high blocking capability is necessary, these diodes being comparatively costly on the one hand, and, on the other hand, they have a large space within a power electronic arrangement due to the required electric creepage distances.

There is a need to overcome or at least to alleviate the problems discussed above. Further, there is a need to provide a circuit arrangement and a corresponding method to prevent the RC-IGBT from being switched on via its gate electrode while being in its reverse conducting state.

SUMMARY

A circuit arrangement is disclosed. In accordance with one embodiment, the circuit arrangement includes: a reverse conducting IGBT configured to allow for conducting load current in a forward direction and in a reverse direction, the IGBT having a load current path and a gate electrode; a gate control unit coupled to the gate electrode and configured to activate or deactivate the IGBT by charging or, respectively, discharging the gate electrode in accordance with a gate control signal; a monitoring unit configured to detect whether the IGBT conducts current in the reverse direction by sensing a gate current caused by a change of a voltage drop across the load path due to a changing of the reverse conducting IGBT into its reverse conducting state, wherein the gate control unit is further configured to deactivate the IGBT or to prevent an activation of the IGBT when the monitoring unit detects that the IGBT is in its reverse conducting state.

Further, a method is disclosed for detecting whether an IGBT is in a forward conducting state or in a reverse conducting state. In accordance with one embodiment, the method includes: providing a reverse conducting state having a load path; and sensing a gate current caused by a change of a voltage drop across the load path due to a changing of the reverse conducting IGBT into its reverse conducting state.

The circuit arrangement and the method according to the present invention may also be used for any field-effect transistor and is not limited to IGBTs.

Those skilled in the art will recognize additional features and advantages upon reading the following detailed description, and upon viewing the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The components in the figures are not necessarily to scale, instead emphasis being placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts. In the drawings:

FIG. 1 illustrates an IGBT half-bridge circuit connected to a load impedance including an inductor;

FIG. 2 illustrates the IGBT half-bridge of FIG. 1 including a gate control unit for each transistor;

FIG. 3 illustrates a table listing the conducting states of the half-bridge IGBTs of FIG. 2 in dependence on the load current direction of the corresponding IGBT;

FIG. 4 illustrates an IGBT with a gate control unit monitoring the conducting state of the corresponding IGBT and switching the transistor depending on its conducting state;

FIG. 5 illustrates the dominant parasitic input capacitances related to an RC-IGBT;

FIG. 6 illustrates a variety of options for gate current sensing using a gate driver circuit;

FIG. 7 illustrates one advantageous circuit arrangement for gate current sensing by means of a current mirror connected to a gate driver circuit;

FIG. 8 illustrates the schematic time response of control signals, currents, and voltages of the inverter half-bridge in FIG. 1 and the circuit arrangement in FIG. 7.

 DETAILED DESCRIPTION

Insulated gate field effect transistors (IGBTs) usually are operated having a free-wheeling diode connected anti-parallel to their load current paths. Dependent on the load current direction and the IGBT gate signal, such a parallel circuit of IGBTs and diode may be in a forward conducting state, in a reverse conducting state or in an in forward blocking state. In the present example, the load current direction is essentially determined by the state of an inductive load. The forward current flows through the load current path (collector-emitter-path) of the IGBT and the reverse current flows through the diode. During reverse conduction, the conducting state of the diode is not influenced by the IGBTs regardless of whether the gate of the IGBT is actively driven to a high level or a low level.

So-called reverse conducting IGBTs (RC-IGBTs) have an intrinsic reverse diode such that no separate free-wheeling diode is required. However, the conduction behaviour of the intrinsic reverse diode heavily depends on whether the gate of the RC-IGBT is actively driven to a high level or a low level.

In RC-IGBTs the reverse voltage drop across the intrinsic reverse diode increases to undesired high levels if the IGBT gate is driven to a high level. The reverse voltage can thereby be regarded as the anode-cathode voltage across the load path of the internal free-wheeling diode.

RC-IGBTs are commonly used in (power) converters, wherein an IGBT half-bridge is used to control each phase-current. FIG. 1 schematically illustrates such a half-bridge. The IGBT half-bridge may include two n-channel IGBTs T1TOP, T1BOT connected in series. An operating voltage VDC (intermediate voltage or DC-link voltage) is applied across the half bridge, i.e. between the collector of the high-side transistor T1TOP and the emitter of the low-side transistor T1BOT. The collector potential of the high-side transistor T1TOP is referred to herein as VCC, whereas the emitter potential of the bottom transistor T1BOT is referred to herein as VEE. The common circuit node of T1TOP and T1BOT is the output node of the half-bridge that provides the output current iOUT (also called phase-current) to a connected load impedance. In most cases the load impedance includes an inductive load. Commonly, the DC link voltage is symmetrical with respect to a reference potential GND, i.e. VCC=VDC/2 and VEE=−VDC/2. In the present case, the load is connected between the output of the half-bridge and the reference potential GND.

A gate control unit 10 is provided for each transistor T1TOP, T1BOT to control the output current iOUT (half-bridge output current). The gate control unit 10 may include a gate driver unit 102 as shown in FIG. 4 which is described later herein, which may be configured to achieve a certain time response of the current iOUT (i.e. edge-shaped current steps) and/or to prevent the half-bridge from being short-circuited. Both of the gate control units 10 are provided with a (binary) gate control signal S1TOP and S1BOT, each indicating the desired conduction state of the corresponding transistor T1TOP or T1BOT. A converter half-bridge including gate control units 10 is illustrated in FIG. 2.

FIG. 3 illustrates the four possible switching states of a half-bridge, e.g. the half-bridge illustrated in FIG. 2. Depending on the control signals S1TOP and S1BOT and the direction of the output current iOUT (load current), which may be determined by an inductive load, the output potential of the half-bridge output node takes either (approximately) the value VEE=−VDC/2 or VCC=VDC/2.

Referring to FIG. 3, in switching state I both RC-IGBT control signals S1TOP, S1BOT drive the corresponding transistor to its off state. Thus, the output current iOUT can only flow through the intrinsic reverse diode of either the transistor T1TOP or the transistor T1BOT in accordance with the direction of the output current iOUT. That is, the output current iOUT either flows through the intrinsic free-wheeling diode of the top transistor T1TOP or through the free-wheeling diode of bottom transistor. Consequently, the output potential is either VEE (in case of a negative output current iOUT) or VCC (in case of a positive output phase current iOUT), this being a mere approximation as the forward voltage drop of the intrinsic diode is neglected in the above consideration.

Again referring to FIG. 3, in switching state II only the transistor T1BOT is driven so as to assume its conducting state (i.e. S1BOT=1, S1TOP=0), also called the on-state. Thus, independent of the direction of the output current iOUT, the output potential is approximately equal to VEE. In case of a positive output current iOUT (a positive current flows in the direction as indicated in FIG. 2 by the arrow referring to iOUT) the transistor T1BOT is in its forward conducting state, whereas, in case of a negative output current, T1BOT is in its reverse conducting state. Switching state III, is complementary to state II.

State IV refers to a state in which both transistors T1BOT and T1TOP are driven into their forward conducting states (i.e. S1TOP=S1BOT=1). However, in such a state the DC link voltage VDC would be short-circuited, which may result in malfunctions or even in the destruction of the half-bridge. The gate driver circuit should therefore be configured to prevent such a half-bridge shoot-through.

In switching states II and III the above-mentioned problem may occur, when a transistor is activated (i.e. S1TOP=1 or, respectively, S1BOT=1) while being in the reverse conducting state. In this case the voltage drop across its load-path would considerably increase, which would inevitably result in increased losses and consequently in significant heat generation. More precisely, the problem may occur in state II in case of a negative output current iOUT, and in state III in case of a positive output current iOUT.

The problem might be solved or at least alleviated by a circuit arrangement built in accordance to one example of the present invention. FIG. 4 illustrates a circuit arrangement that includes a reverse conducting IGBT T1 capable of conducting via its load path, a load current in forward direction (via a MOS channel) as well as in reverse direction (via the intrinsic anti-parallel reverse diode).

Transistor T1 includes a gate electrode G connected to a gate control unit 10, that drives the forward current path of the IGBT on and off in accordance with a corresponding control signal S1. The gate control unit 10 includes a gate driver unit 102 coupled to the gate electrode G, the gate control unit 10 being configured to supply appropriate gate driver signals to the gate G in accordance with the control signal S1, thus switching the forward current path of the transistor into an on-state or an off-state.

The gate driver signal supplied to the gate electrode G may be a suitable gate-emitter voltage VGE or a gate current iGATE, depending on whether the transistor T1 is to be current-driven or voltage-driven.

There are numerous alternatives as to how the signal S1 may be blocked or blanked so as to prevent a switch-on of the transistor T1 even when S1 is at a high-level. For example, the signal S1 is passed through an AND-gate 103 before being admitted to the gate driver unit 102 which generates the gate control signal. A current direction signal SVR is generated by the gate driver unit 102 and further components described later herein and applied to an input of the AND-gate 103. Assuming that SVR=0, if the transistor T1 is in its reverse conducting state, the signal S1 will be blanked before reaching the gate driver unit 102. Thus, the signal S1 is prevented from activating the transistor T1 while the transistor is in its reverse conducting state. Further assuming that SVR=1, if the transistor is not in its reverse conducting state (on-state or blocked), the signal S1 is not blanked and is passed on to the gate driver unit 102, which may then generate an appropriate gate driver signal switching the transistor to its on-state.

Parasitic capacitances are treated as inherent to every electronic switch, thus also inherent to an RC-IGBT as used in the example introduced in FIGS. 1 to 4. The gate-collector capacitance CGC and gate-emitter capacitance CGE are illustrated in the circuit diagram of FIG. 5. However, as compared to the gate-collector capacitance CGC the gate-emitter capacitance CGE can be neglected in the following considerations. To charge and discharge the mentioned capacitances, in particular the gate-collector capacitance CGC, a gate current iGATE has to flow into the gate electrode G or, respectively, originate therefrom. The capacitance CGC is charged or discharged upon a change of the direction of the voltage drop across the load path (collector-emitter path) of the RC-IGBT. The change of the direction of the voltage drop is caused by a change of the load current of the respective IGBT. The charging of the capacitance CGC implies a displacement current resulting in a corresponding gate current iG which the present invention may utilize to determine the conduction state of the corresponding RC-IGBT.

In order to attain the conducting state signal SVR, the gate driver unit 102 can sense the gate current flowing to and from the gate electrode G of the RC-IGBT. FIG. 6 shows an embodiment in accordance with the present invention. Current-measuring devices, for example appropriate ampere meters, are arranged in the first supply line of the gate driver (device A1), and/or in the second supply line (device A2), and/or in the gate driver output line (device A3), the gate driver output line being sometimes referred to as ‘gate wire’ in common literature. Conceivably, also two or more of the three current-measuring devices may be arranged in the circuitry according to the invention. Alternatively, though, only one of the three current measuring devices may be sufficient in many applications.

In accordance with the example illustrated in FIG. 6, a shunt resistance RS may be connected as current measuring device between the gate driver unit 102 and the gate electrode of the connected transistor T1 (i.e. the gate resistor is inserted in the gate driver output line). In this case, a parallel voltage-measuring device V1 may serve to sense the gate current in accordance with one example of the present invention. It is possible to combine one or more than one of the current-measuring devices or methods for gate current sensing described herein. In accordance with another example of the present invention, a shunt resistance is arranged in the first (high side) and/or in the second (low side) supply line supplying the gate driver unit 102. Again, the voltage drop across the shunt resistor may be measured, thus obtaining a measurement value representative of the gate current.

It is to be noted that a resistance is commonly arranged between the gate G and the output of the gate driver 102. This resistance, usually referred to as gate resistance RG (not shown), could also be used as shunt resistor for current sensing instead of an additional shunt resistance RS. It is noted that any combination of current-measuring devices and shunt (or gate) resistances may be employed in accordance with examples of the present invention, even if not explicitly referred to herein.

Still referring to FIG. 6, the gate driver unit 102 is shown to be supplied by a voltage source (V+, V−). A current sensing device A1 or A2 may be inserted in the supply lines providing current to the gate driver unit 102. A signal representative of the gate current might be sensed at various positions in gate control circuit 10. Further, current sensing may be performed at more than one position in the same circuitry.

FIG. 7 illustrates one particular implementation of the current-measuring device A1 of FIG. 6. Further, some details of the gate driver unit 102 are illustrated in FIG. 7. The gate control signal S1 (if not blanked) is provided to the gate electrodes of an emitter follower stage across a series resistance RV. The emitter follower stage (output stage), for example, can be formed by bipolar transistors D1 (which may be an npn transistor) and bipolar transistor D2 (which may be a pnp transistor). Instead of bipolar transistors, any transistor type (i.e. MOSFETs) may be applied for implementation of the emitter follower stage. The emitter follower stage provides the gate current iGATE to the RC-IGBT T1 across the gate driver output line (or gate wire). For the purpose of switching speed control (e.g. “edge shaping”), in most cases a gate resistance RG is placed in the gate wire.

As shown in FIG. 7, the current measuring device may comprise a current mirror which is formed by the transistors M1 and M2. A capacitor CG and an electronic switch E1 (i.e. another transistor) connected parallel thereto are coupled to the current mirror such that the mirror current may charge the capacitor CG. The current mirror provides a first current path across M1 and a second current path across M2. The first current path preferably carries the gate current iGATE, whereas the second current path carries a mirror current iGATE*=n·iGATE (where n is usually lower or equal than 1) which is equal or directly proportional to the gate current iGATE. Both the gate current iGATE and the mirror current iGATE* may be provided by the voltage sources V+ and V−.

As described hereinbefore, it is the nature of an RC-IGBT T1 that, upon a sudden change in the voltage drop across its collector-emitter path (e.g. a VCE voltage dip), the intrinsic parasitic capacitance CGC (and CGE) will be charged or discharged, resulting in a corresponding gate current iGATE. A discharging of the parasitic gate-collector capacitance CGC is particularly caused by a collector-emitter voltage (VCE) dip of a transistor presently passing from a forward blocking into its reverse conducting state. The voltage dip results in a displacement current and respective gate current which can be measured in the gate wire (gate supply line). If a gate current iGATE passes through the transistor M1, it is mirrored by the current mirror such that the sense capacitance CG will “see” a change of its charge state resulting from the mirror current iGATE*. The switch E1 may be used to short-circuit and discharge the capacitance CG. The charge QG stored in the capacitance CG is monitored appropriately using the relation QG=CG·VCG, i.e. by measuring the voltage drop VCG across CG. The value of CG is chosen appropriately according to the relation between the currents iGATE and iGATE*. Thus, the charge stored in capacitor CG is a measure for the gate current iGATE, the gate current again being a measure for the conducting state of the RC-IGBT. A circuit arrangement, particularly a gate driver circuit, as depicted in FIG. 7 for an RC-IGBT T1, is provided for each RC-IGBT of an inverter half-bridge (as shown in FIG. 1). A detailed description of how the conducting state of the RC-IGBT may be derived from the gate current iGATE, that is from the state of charge of the capacitance CG, is given next.

Considering switch E1, the capacitor CG is charged by iGATE* as long as E1 is open. If E1 is closed, capacitor CG will be discharged and a measurement cycle thus initialized. According to embodiment, the switch E1 may be switched synchronously with the gate control signal S1 of the corresponding RC-IGBT. The switch E1 resets the charge state (i.e. initializing step) of the sense capacitance CG prior to the initiation of the sense period, the sense period being the time interval while the RC-IGBT conducting state is determined.

In FIG. 8 is a timing diagram depicting the time response of signals, voltages, and currents in an inverter half-bridge such as illustrated in FIG. 1 with a gate control unit 10, each half-bridge RC-IGBT (T1BOT, T1TOP) comprising a gate driver circuit arrangement according to FIG. 7, where the transistor T1 may be a high side transistor (T1TOP) or a low side transistor (T1BOT).

The time responses of the control signals S1BOT and S1TOP are depicted in FIG. 8, where a high-signal means “turn-on transistor” and a low-signal means “turn-off transistor”. Assuming a positive load current iOUT (referring to the load current direction indicated in FIG. 1), the top transistor T1TOP will be in its reverse conducting state, as soon as the bottom transistor T1BOT is turned off by its control signal S1BOT. For a negative load current direction iOUT (again referring the direction indicated in FIG. 1), the same coherences are valid, but the designator “top” and “bottom” is to be exchanged.

It can be seen from FIG. 8 that a turn-off command of the bottom transistor T1BOT (i.e. at time instants t2 or t6) initiates the reverse conducting state of the top transistor T1TOP. In practice a dead time or delay time tDELAY between, for example, the time instants t2 and t3 or t6 and t7, is inevitable, meaning that neither control signal S1TOP or S1BOT is high and that neither transistor T1TOP or T1BOT is turned on via its gate electrode G, so as to avoid shoot-throughs. At the instants t3 or t7 the dead time tDELAY has elapsed and the control signal S1TOP of the reverse conducting top transistor T1TOP switches to a high level. As explained hereinbefore, the low-high transition of the control signal S1 (S1TOP or S1BOT) of a currently reverse conducting RC-IGBT T1 (T1TOP or T1BOT) should be prevented from switching on the transistor. Considering the inverter half-bridge in FIG. 1, this means that conducting state information concerning the conducting state of the corresponding RC-IGBT should be made available. Dependent on the conducting state information, a decision can be made whether to allow the gate control signal S1 to switch on its corresponding transistor T1, or whether to blank the control signal S1.

It can further be seen from FIG. 8 that in case of a currently reverse conducting RC-IGBT (in the present case the top transistor T1TOP) and an imminent low-high transition of the corresponding control signal (i.e. S1TOP), its collector-emitter-voltage VCE is zero. In this case a VCE voltage transition from VDC (which is an idealized consideration) to zero, this transition is also be referred to as VCE voltage dip, happened a dead time period earlier (i.e. tDELAY=t3−t2=t7−t6). The VCE voltage dip may be detected and used for generating a conducting state signal SVR. As already indicated, the VCE voltage dip, in the present case across the collector-emitter path of the reverse conducting top transistor T1TOP, causes a displacement current resulting in a respective gate current in the gate wire, as the charge changes in the parasitic capacitance CGC. The dead time period tDELAY lies between the event of the VCE voltage dip and the very time instant the conducting state signal SVR must be available, whereby the dead time period commonly is long enough (i.e. tDELAY≧1 μs) for electing the VCE voltage dip event and deriving a robust conducting state signal SVR.

Considering FIG. 8, E1TOP is closed until the instant t0 as S1TOP is high. Upon reaching the instant t0, E1TOP is opened and CGTOP carries no charge. Upon the dead time (i.e. tDELAY=t1−t0) having elapsed at the instant t1, the bottom IGBT T1BOT is switched on by its gate electrode (S1BOT undergoes a low-high transition at t=t1). Switching on the bottom transistor T1BOT is enabled in this case as no reverse conducting state of T1BOT has been detected. Further, switching on the bottom transistor T1BOT causes the load current iOUT to commutate from the currently reverse conducting top transistor T1TOP into the now forward conducting bottom transistor T1BOT. Subsequently, the collector-emitter voltage VCETOP across the top transistor increases from ideal zero to almost VDC (the DC-link voltage). Initiated by the increase of VCETOP, the gate-collector capacitance of the top transistor CGCTOP is charged, the (displacement) current iGATE in this case passing through the gate resistance RG and the transistor D2 e.g. as shown in FIG. 7. As the gate current does not pass transistor M1, no charge is transferred to the sense capacitor CG, and thus the voltage VCG across the sense capacitance CG remains zero.

Arriving at the time instant t2, the gate control signal of the bottom transistor T1BOT switches to its low state and forces the gate electrode to switch off. Consequently, the load current iOUT instantaneously commutates into the top transistor T1TOP. Prior to the top transistor passing into its reverse conducting state, the top collector-emitter voltage VCETOP drops from ideal VDC down to zero, causing the gate-collector capacitance CGCTOP of the top transistor to discharge resulting in gate current iGATE via the gate wire, the gate resistance RG, the transistor D1, and the current mirror transistor M1. As such, the discharge current iGATE is mirrored by the current mirror transistor M2. Depending on the current mirror ratio n, the charge injected by the mirror current iGATE*=n·iGATE into the sense capacitance CG is proportional to the charge originating from the gate-collector capacitance CGCTOP of the top transistor, as switch E1 is now open. Burdening the sense capacitance CG with a charge causes a voltage drop VGTOP which remains constant upon the switching transients having diminished. This means that if the collector-emitter voltage suddenly drops while the corresponding transistor passes into its reverse conducting state or has already passed into the reverse conducting state, the voltage VGTOP across the sense capacitance will almost immediately be increased by a displacement gate current caused by the collector-emitter voltage dip.

Progressing to time instant t3, the gate control signal of the top transistor S1TOP switches to its high state. An appropriate (and commonly used) control circuit (which is not explicitly depicted in the accompanying Figures) is now used to combine the low-high transition of the gate control signal of the top transistor S1TOP with the voltage drop information across the sense capacitance CG. If the voltage drop VG exceeds a certain threshold value, the gate control signal S1BOT is prevented from advancing to the gate electrode of the RC-IGBT in question (e.g. by means of the AND-gate in FIG. 4 and/or a particular embodiment of the gate driver unit 102).

The example provided hereinabove explains how the reverse conducting state of the top transistor T1TOP may be determined according to the present invention. The same principle is, of course, also valid vice versa, that is, if the conducting state of the bottom transistor of the inverter half-bridge T1BOT is to be determined.

Spatially relative terms such as “under”, “below”, “lower”, “over”, “upper” and the like, are used for ease of description to explain the positioning of one element relative to a second element. These terms are intended to encompass different orientations of the device in addition to different orientations than those depicted in the figures. Further, terms such as “first”, “second”, and the like, are also used to describe various elements, regions, sections, etc. and are also not intended to be limiting. Like terms refer to like elements throughout the description.

As used herein, the terms “having”, “containing”, “including”, “comprising” and the like are open ended terms that indicate the presence of stated elements or features, but do not preclude additional elements or features. The articles “a”, “an” and “the” are intended to include the plural as well as the singular, unless the context clearly indicates otherwise.

With the above range of variations and applications in mind, it should be understood that the present invention is not limited by the foregoing description, nor is it limited by the accompanying drawings. Instead, the present invention is limited only by the following claims and their legal equivalents.

Claims

1. A circuit arrangement comprising:

a reverse conducting IGBT configured to allow for conducting a load current in a forward direction and in a reverse direction, the IGBT having a load current path and a gate electrode;
a gate control unit connected to the gate electrode and configured to activate or deactivate the IGBT by charging or, respectively, discharging the gate electrode in accordance with a gate control signal;
a gate driver unit included in the gate control unit and configured to detect whether the IGBT conducts current in the forward direction or the reverse direction by sensing a gate current caused by a change of a voltage drop across the load path due to a changing of the reverse conducting IGBT into its reverse conducting state;
wherein the gate control unit is further configured to deactivate the IGBT or to prevent an activation of the IGBT via its gate electrode when the gate driver unit detects that the IGBT is in its reverse conducting state.

2. The circuit arrangement according to claim 1, wherein the gate driver unit is configured to generate a conduction state signal from the sensed gate current, the conduction state signal indicating the conduction state of the reverse conducting IGBT.

3. The circuit arrangement according to claim 1, wherein the gate control unit is configured to process a current direction signal such that an activation of the reverse conducting IGBT via its gate electrode is prevented if the current direction signal indicates the reverse conducting IGBT is in its reverse conducting state.

4. The circuit arrangement according to claim 1, wherein the gate driver unit is supplied by a voltage source via a first and a second supply line, the gate driver unit comprises an output and an output line, and the gate driver unit output is configured to inject or receive a gate current into or from the gate electrode of the reverse conducting IGBT via the output line, the gate driver output current being the gate current.

5. The circuit arrangement according to claim 4, wherein a current-measuring device is arranged in the first supply line of the gate driver unit, or in the second supply line, or in the output line, or wherein a plurality of current-measuring devices are arranged in the first supply line of the gate driver, and/or in the second supply line, and/or in the output line.

6. The circuit arrangement according to claim 5, wherein the current-measuring devices comprise a shunt resistance and a parallel voltage-measuring device.

7. The circuit arrangement according to claim 4, wherein a voltage-measuring device is connected parallel to a gate resistance, the gate resistance being arranged in the gate driver output line.

8. The circuit arrangement according to claim 4, wherein a shunt resistance with a parallel voltage-measuring device is arranged in the output line.

9. The circuit arrangement according to claim 5, wherein the current-measuring device in the first and/or the second supply line comprises a current mirror having a first and a second current path.

10. The circuit arrangement according to claim 9, wherein the current mirror is arranged such that the current in its first current path is equal to the gate current, and the current mirror is configured such that the current in its second current path is equal or directly proportional to the current in its first current path.

11. The circuit arrangement according to claim 9, wherein a sense capacitance is arranged in the second current path of the current mirror to be charged by the current flowing in the second current path of the current mirror, and a charge quantity carried by the sense capacitance is a measure for the gate current.

12. A method for the detection of a forward or a reverse conducting state of a reverse conducting IGBT, the method comprising:

providing a reverse conducting IGBT having a load path; and
sensing a gate current caused by a change of a voltage drop across the load path due to a changing of the reverse conducting IGBT into its reverse conducting state.

13. The method according to claim 12, comprising:

providing a conduction state signal indicating the conduction state of the reverse conducting IGBT;
blocking a gate control signal for the reverse conducting IGBT if the conduction state signal indicates that the reverse conducting IGBT is in its reverse conduction state.

14. The method according to claim 12, comprising:

charging or discharging a sense capacitance by means of a current proportional or equal to the gate current;
sensing a charge quantity carried by a capacitance, said charge quantity being representative of the gate current.

15. A circuit arrangement comprising:

a field-effect transistor configured to allow for conducting a load current in a forward direction and in a reverse direction, the field-effect transistor having a load current path and a gate electrode;
a gate control unit coupled to the gate electrode and configured to activate or deactivate the field-effect transistor by charging or, respectively, discharging the gate electrode in accordance with a gate control signal; and
a monitoring unit configured to detect whether the field-effect transistor conducts current in the forward direction or the reverse direction by sensing a gate current caused by a change of direction of a voltage drop across the load path corresponding to a change of the direction of the load current.

16. A method for the detection of a forward or a reverse conducting state of a field-effect transistor, the method comprising:

providing a field-effect transistor having a load path and configured to allow for conducting a load current in a forward direction and in a reverse direction; and
sensing a gate current caused by a change of direction of a voltage drop across the load path corresponding to a change of the direction of the load current.

17. A circuit arrangement comprising:

a half-bridge switching circuit comprising two reverse conducting insulated gate bipolar transistors (RC-IGBTs) configured to allow for conducting a load current in a forward direction and in a reverse direction, the RC-IGBTs having a load current path and a gate electrode and a common circuit node of the two RC-IGBTs being a half-bridge output configured to provide an output current;
a gate control unit for each RC-IGBT that is connected to the respective gate electrode and configured to activate or deactivate the respective RC-IGBT by charging or, respectively, discharging the gate electrode in accordance with a gate control signal received by the gate control unit;
at least one of the gate control units including a gate driver unit configured to detect whether the RC-IGBT conducts current in the reverse direction by sensing a gate current caused by a change of a voltage drop across the load path of the respective RC-IGBT due to a changing of the RC-IGBT into its reverse conducting state.
Referenced Cited
U.S. Patent Documents
4841166 June 20, 1989 Harnden
20030179021 September 25, 2003 Braun et al.
20100079192 April 1, 2010 Strzalkowski
Foreign Patent Documents
10212869 September 2003 DE
102006025374 December 2007 DE
102009001029 August 2010 DE
2007/137569 December 2007 WO
Patent History
Patent number: 8729914
Type: Grant
Filed: Nov 10, 2010
Date of Patent: May 20, 2014
Patent Publication Number: 20120112775
Assignee: Infineon Technologies AG (Neubiberg)
Inventor: Daniel Domes (Ruethen)
Primary Examiner: Tung X Nguyen
Application Number: 12/943,079
Classifications
Current U.S. Class: With A Capacitive Sensing Means (324/686)
International Classification: G01R 19/14 (20060101);