Communication system
At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into low and high frequency band components which are designated as first and second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.
Latest Panasonic Patents:
This application is a continuation-in-part of application Ser. No. 07/857,627, filed Mar. 25, 1992. This is a reissue application of U.S. Pat. No. 5,600,672, issued Feb. 4, 1997, and a divisional application of reissue application No. 09/680,176, filed Oct. 5, 2000 now abandoned, which is a divisional application of reissue application No. 09/244,037, filed Feb. 4, 1999, which is also a reissue application of U.S. Pat. No. 5,600,672, issued Feb. 4, 1997 which is a Continuation-In-Part of application Ser. No. 07/857,627, filed Mar. 25, 1992 now abandoned. Further reissue divisional applications have been filed, all of which are reissues of Pat. No. 5,600,672 and divisional applications of reissue application 09/244,037. These further applications are: 09/677,421, filed Oct. 5, 2000; 09/678,014, filed Oct. 5, 2000; 09/677,420, filed Oct. 5, 2000; 09/680,177, filed Oct. 5, 2000; 09/686,467, filed Oct. 12, 2000; 09/686,463, filed Oct. 12, 2000; 09/686,466, filed Oct. 12, 2000; 09/688,028, filed Oct. 12, 2000; 09/686,464, filed Oct. 12, 2000; 09/686,465, filed Oct. 12, 2000; 09/666,012, filed Sep. 19, 2000; 09/667,525, filed Sep. 21, 2000; 09/667,438, filed Sep. 21, 2000; 09/668,068, filed Sep. 25, 2000; 09/662,695, filed Sep. 15, 2000; 09/669,916, filed Sep. 25, 2000; 09/672,948, filed Sep. 29, 2000; 09/672,946, filed Sep. 29, 2000; 09/672,947, filed Sep. 29, 2000; 10/133,347, filed Apr. 29, 2002; and 10/133,364, filed Apr. 29, 2002.
BACKGROUND OF THE INVENTION1. Field of the Invention
The present invention relates to a communication system for transmission and reception of a digital signal through modulation of its carrier wave and demodulation of the modulated signal.
2. Description of the Prior Art
Digital signal communication systems have been used in various fields. Particularly, digital video signal transmission techniques have been improved remarkably.
Among them is a digital TV signal transmission method. So far, such digital TV signal transmission systems are in particular use for transmission between TV stations. They will soon be utilized for terrestrial and/or satellite broadcast service in every country of the world.
The TV broadcast systems including HDTV, PCM music, FAX, and other information services are now demanded to increase desired data in quantity and quality for satisfying millions of sophisticated viewers. In particular, the data has to be increased in a given bandwidth of frequency allocated for TV broadcast service. The data to be transmitted is always abundant and provided as much as handled with up-to-date techniques of the time. It is ideal to modify or change the existing signal transmission system corresponding to an increase in the data amount with time.
However, the TV broadcast service is a public business and cannot go further without considering the interests and benefits of viewers. It is essential to have any new service compatible with existing TV receivers and displays. More particularly, the compatibility of a system is much desired for providing both old and new services simultaneously or one new service which can be intercepted by both the existing and advanced receivers.
It is understood that any new digital TV broadcast system to be introduced has to be arranged for data extension in order to respond to future demands and technological advantages and also, for compatibility to allow the existing receivers to receive transmissions.
The expansion capability and compatible performance of the prior art digital TV system will be explained.
A digital satellite TV system is known in which NTSC TV signals compressed to an about 6 Mbps are muitiplexed multiplexed by time division modulation of 4 PSK and transmitted on 4 to 20 channels while HDTV signals are carried on a signal channel. Another digital HDTV system is provided in which HDTV video data compressed to as small as 15 Mbps are transmitted on a 16 or 32 QAM signal through ground stations.
Such a known satellite system permits HDTV signals to be carried on the channel by a conventional manner, thus occupying a band of frequencies equivalent to the same channels of NTSC signals. This causes the corresponding NTSC channels to be unavailable during the transmission of the HDTV signal. Also, the compatibility between NTSC and HDTV receivers or displays is hardly concerned and data expansion capability needed for matching a future advanced mode is utterly disregarded.
Such a common terrestrial HDTV system offers an HDTV service on conventional 16 or 32 QAM signals without any modification. In any analogue TV broadcast service, there are developed a lot of signal attenuating or shadow regions within its service area due to structural obstacles, geographical inconveniences, or signal interference from a neighbor station. When the TV signal is an analogue from form, it can be intercepted more or less at such signal attenuating regions although its reproduced picture is low in quality. If the TV signal is a digital form, it can rarely be reproduced at an acceptable level within the regions. This disadvantage is critically hostile to the development of any digital TV system.
SUMMARY OF THE INVENTIONIt is an object of the present invention, for solving the foregoing disadvantages, to provide a communication system arranged for compatible use for both the existing NTSC and newly introduced HDTV broadcast services, particularly via satellite and also, for minimizing signal attenuating or shadow region of its service area on the grounds ground.
A communication system according to the present invention intentionally varies signal points, which used to be disposed at uniform intervals, to perform the signal transmission and reception. For example, if applied to a QAM signal, the communication system comprises two major sections: a transmitter having a signal input circuit, a modulator circuit for producing m numbers of signal points, in a signal vector field through modulation of a plurality of out-of-phase carrier waves using an input signal supplied from the input circuit, and a transmitter circuit for transmitting a resultant modulated signal; and a receiver having an input circuit for receiving the modulated signal, a demodulator circuit for demodulating one-bit signal points of a QAM carrier wave, and an output circuit.
In operation, the input signal containing a first data stream of n values and a second data stream is fed to the modulator circuit of the transmitter where a modified m-bit QAM carrier wave is produced representing m signal points in a vector field. The m signal points are divided into n signal point groups to which the n values of the first data stream are assigned respectively. Also, data of the second data stream are assigned to m/n signal points or sub groups of each signal point group. Then, a resultant transmission signal is transmitted from the transmitter circuit. Similarly, a third data stream can be propagated.
At the p-bit demodulator circuit, p>m, of the receiver, the first data stream of the transmission signal if is first demodulated through dividing p signal points in a signal space diagram into n signal point groups. Then, the second data stream is demodulated through assigning p/n values to p/n signal points of each corresponding signal point group for reconstruction of both the first and second data streams. If the receiver is at P=n, the n signal point groups are reclaimed and assigned the n values for demodulation and reconstruction of the first data stream.
Upon receiving the same transmission signal from the transmitter, a receiver equipped with a large sized antenna and capable of large-data modulation can reproduce both the first and second data streams. A receiver equipped with a small sized antenna and capable of small-data modulation can reproduce the first data stream only. Accordingly, the compatibility of the signal transmission system will be ensured. When the first data stream is an NTSC TV signal or low frequency band component of an HDTV signal and the second data stream is a high frequency band component of the HDTV signal, the small-data modulation receiver can reconstruct the NTSC TV signal and the large-data modulation receiver can reconstruct the HDTV signal. As understood, a digital NTSC/HDTV simultaneous broadcast service will be feasible using the compatibility of the signal transmission system of the present invention.
More specifically, the communication system of the present invention comprises: a transmitter having a signal input circuit, a modulator circuit for producing m signal point, points in a signal vector field through modulation of a plurality of out-of-phase carrier waves using an input signal supplied from the input, and a transmitter circuit for transmitting a resultant modulated signal, in which the main procedure includes receiving an input signal containing a first data stream of n values and a second data stream, dividing the m signal points of the signal into n signal point groups, assigning the n values of the first data stream to the n signal point groups respectively, assigning data of the second data stream to signal points of each signal point group respectively, and transmitting the resultant modulated signal; and a receiver having an input circuit for receiving the modulated signal, a demodulator circuit for demodulating p signal points of a QAM carrier wave, and an output circuit, in which the main procedure includes dividing the p signal points into n signal point groups, demodulating the first data stream of which n values are assigned to the n signal point groups respectively, and demodulating the second data stream of which p/n values are assigned to p/n signal points of each signal point group respectively. For example, a transmitter produces a modified m-bit QAM signal of which first, second, and third data streams, each carrying n values, are assigned to relevant signal point groups with a modulator. The signal can be intercepted and the first data stream only reproduced by a first receiver, both the first and second data streams can be reproduced by a second receiver, and all the first, second, and third streams can be reproduced by a third receiver.
More particularly, a receiver capable of demodulation of n-bit data can reproduce n bits from a multiple-bit modulated carrier wave carrying m-bit data where m>n, thus allowing the communication system to have compatibility and capability of future extension. Also, a multi-level signal transmission will be possible by shifting the signal points of QAM so that a nearest signal point to the origin point of I-axis and Q-axis coordinates is spaced nf from the origin where f is the distance of the nearest point from each axis and n is more than 1.
Accordingly, a compatible digital satellite broadcast service for both the NTSC and HDTV systems will be feasible when the first data stream carries an NTSC signal and the second data stream carries a difference signal between NTSC and HDTV. Hence, the capability of corresponding to an increase in the data amount to be transmitted will be ensured. Also, on the ground, the service area will be increased while signal attenuating areas are decreased.
FIG. 25(a) and 25(b) are vector diagrams respectively showing an 8 and a 16 QAM signal of the first embodiment;
FIGS. 59(a) and 59(b) are diagrams showing assignment of signal points of the modified 4 ASK signal of the fifth embodiment and FIGS. 59(c) and 59(d) are diagrams respectively showing the slice levels of the modulated 4 ASK signal in subchannels 1 and 2;
FIG. 62(a) is a wave spectrum diagram of the ASK signal, i.e., a multi-value VSB signal before filtering, in the fifth embodiment of the invention and FIG. 62(b) is a wave spectrum diagram showing the characteristics of the filtered VSB signal;
FIG. 68(a) is an 8-level VSB constellation map in the fifth and sixth embodiments of the invention;
FIG. 68(b) is an 8-level VSB constellation map in the fifth and sixth embodiments of the invention;
FIG. 68(c) is an 8-level VSB signal-time waveform diagram in the fifth and sixth embodiments of the invention;
FIG. 74(a) is a block diagram of a video decoder of the fifth embodiment;
FIG. 74(b) is a diagram showing another time assignment of data components of the transmission signal according to the fifth embodiment;
FIG. 108(a) is a diagram showing a frequency distribution profile of a conventional TV signal;
FIG. 108(b) is a diagram showing a frequency distribution profile of a conventional two-layer TV signal;
FIG. 108(c) is a diagram showing threshold values of the third embodiment;
FIG. 108(d) is a diagram showing a frequency distribution profile of two-layer OFDM carriers of the ninth embodiment, and FIG. 108(e) is a diagram showing threshold values for three-layer OFDM of the ninth embodiment;
FIG. 119(a) is a diagram showing a time slot assignment of a conventional system;
FIG. 119(b) is a diagram showing a time slot assignment according to the eighth embodiment;
FIG. 120(a) is a diagram showing a time slot assignment of a conventional TDMA system;
FIG. 120(b) is a diagram showing a time slot assignment according to a TDMA system of the eighth embodiment;
FIG. 125(a) is a view showing a frequency assignment of a modulation signal of a conventional system;
FIG. 125(b) is a view showing a frequency assignment of a modulation signal according to the ninth embodiment;
FIG. 126(a) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein no weighting is applied;
FIG. 126(b) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein two channels of two-layer OFDM are weighted by transmission electric power;
FIG. 126(c) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein carrier intervals are doubled by weighting;
FIG. 126(d) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein carrier intervals are not weighted;
FIG. 128(a) is a block diagram of a trellis encoder (ratio 1/2) in embodiments 2, 4, and 5,
FIG. 128(b) is a block diagram of a trellis encoder (ratio 2/3) in embodiments 2, 4, and 5,
FIG. 128(c) is a block diagram of a trellis encoder (ratio 3/4) in embodiments 2, 4, and 5,
FIG. 128(d) is a block diagram of a trellis decoder (ratio 1/2) in embodiments 2, 4, and 5,
FIG. 128(e) is a block diagram of a trellis decoder (ratio 2/3) in embodiments 2, 4, and 5,
FIG. 128(f) is a block diagram of a trellis decoder (ratio 3/4) in embodiments 2, 4, and 5;
FIG. 145(a) is a diagram showing the waveform of the guard time and the symbol time in the multi-level OFDM according to the ninth embodiment, wherein multipath is short;
FIG. 145(b) is a diagram showing the waveform of the guard time and the symbol time in the multi-level OFDM according to the ninth embodiment, wherein multipath is long;
FIG. 149(a) is a diagram showing time slots of respective layers according to the ninth embodiment;
FIG. 149(b) is a diagram showing time distribution of guard times of respective layers according to the ninth embodiment;
FIG. 149(c) is a diagram showing time distribution of guard times of respective layers according to the ninth embodiment;
FIG. 159(a) is a signal point positioning diagram in 16-level VSB in the fifth embodiment of the invention;
FIG. 159(b) is a signal point positioning (8-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;
FIG. 159(c) is a signal point positioning (4-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;
FIG. 159(d) is a signal point positioning (16-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;
FIG. 160(a) is a block diagram of an ECC encoder in the fifth and sixth embodiments of the invention;
FIG. 160(b) is a block diagram of an ECC decoder in the fifth and sixth embodiments of the invention;
FIG. 165(a) is a block diagram of the Reed-Solomon encoder in the second, fourth, and fifth embodiments of the invention;
FIG. 165(b) is a block diagram of the Reed-Solomon decoder in the second, fourth, and fifth embodiments of the invention;
FIG. 168(a) is an interleave/deinterleave table for the second, third, fourth, and fifth embodiments of the invention;
FIG. 168(b) shows the interleave distance in the second, third, fourth, and fifth embodiments of the invention;
One embodiment of the present invention will be described referring to the relevant drawings.
In the preferred embodiment of the invention both the transmission apparatus, which comprises a transmitter for transmitting a digital HDTV signal or other digital signal and a receiver for receiving the transmitted signal, and the recording and reproducing apparatus, which records the digital HDTV signal or other digital signal on a magnetic tape or other recording medium and reproduces the recorded signal from said medium, are described.
It should be noted, however, that the configuration, operation, and principle of the digital modulator and demodulator, error correction encoder and decoder, and the encoder and decoder for image coding the HDTV signal are common to the transmission apparatus and the recording and reproducing apparatus, and apply essentially the same technologies. Therefore, to more concisely describe each embodiment, the block diagrams for either the transmission apparatus or the recording and reproducing apparatus are referenced in the description of each embodiment. In addition, the configuration of each embodiment of the invention can be achieved by means of any multi-value digital modulation method, e.g., QAM, ASK and PSK, positioning signal points in a constellation, and for brevity the embodiments of the present invention are described using only one modulation method.
The transmission signal is then sent down through three downlinks 21, 32 31, and 41 to a first 23, a second 33, and a third receiver 43 respectively. In the first receiver 23, the signal intercepted by an antenna 22 is fed through an input unit 24 to a demodulator 25 where its first data stream only is demodulated, while the second and third data streams are not recovered, before being transmitted further from an output unit 26.
Similarly, the second receiver 33 allows the first and second data streams of the signal intercepted by an antenna 32 and fed from an input unit 34 to be demodulated by a demodulator 35 and then, combined by a mixer 37 into a single data stream which is then transmitted further from an output unit 36.
The third receiver 43 allows all of the first, second, and third data streams of the signal intercepted by an antenna 42 and fed from an input unit 44 to be demodulated by a demodulator 45 and then, combined by a mixer 47 into a single data stream which is then transmitted further from an output unit 46.
As understood, the three discrete receivers 23, 33, and 43 have their respective demodulators of different characteristics such that their outputs demodulated from the same frequency band signal of the transmitter 1 contain data of different sizes. More particularly, three different but compatible data can simultaneously be carried on a given frequency band signal to their respective receivers. For example, each of three, existing NTSC, HDTV, and super HDTV, digital signals is divided into low, high, and super high frequency band components which represent the first, the second, and the third data stream respectively. Accordingly, the three different TV signals can be transmitted on a one-channel frequency band carrier for simultaneous reproduction of medium, high, and super high resolution TV images respectively.
The NTSC TV signal is intercepted by a receiver accompanied by a small antenna for demodulation of small-sized data; the HDTV signal is intercepted by a receiver accompanied by a medium antenna for demodulation of medium-sized data, and the super HDTV signal is intercepted by a receiver accompanied by a large antenna for demodulation of large-sized data. Also, as illustrated in
The first receiver 23 demodulates with its demodulator 25 the modulated digital signal supplied from the digital transmitter 51 into the original first data stream signal. Similarly, the same modulated digital signal can be intercepted and demodulated by the second receiver 33 or third receiver 43 into the first data stream or NTSC TV signal. In summary, the three discrete receivers 23, 33, and 43 all can intercept and process a digital signal of the existing TV system for reproduction.
The arrangement of the signal transmission system will be described in more detail.
Assuming that the input signal is a video signal, its low frequency band component is assigned to the first data stream, its high frequency band component to the second data stream, its super-high frequency band component to the third data stream. The three different frequency band signals are fed to a modulator input 61 of the modulator 4. Here, a signal point shifting circuit 67 shifts the positions of the signal points according to an externally given signal. The modulator 4 is arranged for amplitude modulation on two 90°-out-of phase carriers respectively which are then combined into a multiple QAM signal. More specifically, the signal from the modulator input 61 is fed to both a first AM modulator 64 62 and a second AM modulator 63. Also, a carrier wave of cos(2πfct) produced by a carrier generator 64 is directly fed to the first AM modulator 64 62 and also, to a π/2 phase shifter 66 where it is 90° shifted in phase to a sin(2πfct) form prior to being transmitted to the second AM modulator 63. The two amplitude modulated signals from the first and second AM modulators 64 62, 63 are combined by a summer 65 into a transmission signal which is then transferred to the transmitter unit 5 for output. The procedure is well known and will not be further explained.
The QAM signal will now be described in a common 4×4 or 16 state constellation referring to the first quadrant of a space diagram in FIG. 3. The output signal of the modulator 4 is expressed by a sum vector of two, Acos2πfct and Bcos Bsin 2πfct, vectors 81 and 82 which respectively represent the two 90°-out-of-phase carriers. When the distal point of a sum vector from the zero point represents a signal point, the 16 QAM signal has 16 signal points determined by a combination of four horizontal amplitude values a1, a2, a3, and a4 and four vertical amplitude values b1, b2, b3, and b4. The first quadrant in
c11 is a sum vector of a vector 0-a1 and a vector 0-b1 and thus, expressed as c11=a1cos2πfct−b1sin2πfct=Acos (2πfct+dπ/2).
It is now assumed that the distance between 0 and a, in the orthogonal coordinates of
As shown in
When the distance between two adjacent signal points is great, it will be identified by the receiver with much ease. Hence, it is desirable to space the signal points at greater intervals. If two particular signal points are allocated near to each other, they are rarely distinguished and the error rate will be increased. Therefore, it is most preferable to have the signal points spaced at equal intervals as shown in
The transmitter 1 of the embodiment is arranged to divide an input digital signal into a first, a second, and a third data or bit stream. The 16 signal points or groups of signal points are divided into four groups. Then, 4 two-bit patterns of the first data stream are assigned to the four signal point groups respectively, as shown in FIG. 6. More particularly, when the two-bit pattern of the first data stream is 11, one of four signal points of the first signal point group 91 in the first quadrant is selected depending on the content of the second data stream for transmission. Similarly, when 01, one signal point of the second signal point group 92 in the second quadrant is selected and transmitted. When 00, one signal point of the third signal point group 93 in the third quadrant is transmitted and when 10, one signal point of the fourth signal point group 94 in the fourth quadrant is transmitted. Also, 4 two-bit patterns in the second data stream of the 16 QAM signal, or e.g. 16 four-bit patterns in the second data stream of a 64-state QAM signal, are assigned to four signal points or sub signal point groups of each of the four signal point groups 91, 92, 93, and 94 respectively, as shown in FIG. 7. It should be understood that the assignment is symmetrical between any two quadrants. The assignment of the signal points to the four groups 91, 92, 93, and 94 is determined by priority to the two-bit data of the first data stream. As the result, two-bit data of the first data stream and two-bit data of the second data stream can be transmitted independently. Also, the first data stream will be demodulated by using a common 4 PSK receiver having a given antenna sensitivity. If the antenna sensitivity is higher, a modified type of the 16 QAM receiver of the present invention will intercept and demodulate both the first and second data stream streams with equal success.
When the low frequency band component of an HDTV video signal is assigned to the first data stream and the high frequency component to the second data stream, the 4 PSK receiver can produce an NTSC-level picture from the first data stream and the 16- or 64-state QAM receiver can produce an HDTV picture from a composite reproduction signal of the first and second data streams.
Since the signal points are allocated at equal intervals, there is developed in the 4 PSK receiver a threshold distance between the coordinate axes and the shaded area of the first quadrant, as shown in FIG. 9. If the threshold distance is ATO, a PSK signal having an amplitude of ATO will successfully be intercepted. However, the amplitude has to be increased to a three times greater value or 3ATO for transmission of a 16 QAM signal while the threshold distance ATO is maintained. More particularly, the energy needed for transmitting the 16 QAM signal is nine times greater than that for sending the 4 PSK signal. Also, when the 4 PSK signal is transmitted in a 16 QAM mode, energy waste will be high and reproduction of a carrier signal will be troublesome. Above all, the energy available for satellite transmitting is not abundant but strictly limited to minimum use. Hence, no large-energy-consuming signal transmitting system will be put into practice until more energy for satellite transmission is available. It is expected that a great number of the 4 PSK receivers will be introduced into the market as digital TV broadcasting is placed in service. After introduction to the market, the 4 PSK receivers will hardly be shifted to higher sensitivity models because a signal intercepting characteristic gap between the two, old and new, models is high. Therefore, the transmission of the 4 PSK signals must not be abandoned. In this respect, a new system is desperately needed for transmitting the signal point data of a quasi 4 PSK signal in the 16 QAM mode using less energy. Otherwise, the limited energy at a satellite station will degrade the entire transmission system.
The present invention resides in a multiple signal level arrangement in which the four signal point groups 91, 92, 93, and 94 are allocated at a greater distance from each other, as shown in
For clearing the relationship between the signal receiving sensitivity and the transmitting energy, the arrangement of the digital transmitter 51 and the first receiver 23 will be described in more detail referring to FIG. 1.
Both the digital transmitter 51 and the first receiver 23 23 are formed of known types for data transmission or video signal transmission e.g. in TV broadcasting service. As shown in
The resultant modulated signal is shown in the space diagram of FIG. 18.
It is known that the four signal points are allcated allocated at equal distances for achieving optimum energy utilization.
The first receiver 23 is arranged to receive at its small-diameter antenna 22 a desired or 4 PSK signal which is transmitted from the transmitter 1 or digital transmitter 51 respectively through the transponder 12 of the satellite 10 and demodulate it with the demodulator 24 25. In more detail, the first receiver 23 is substantially designed for interception of a digital TV or data communications signal of 4 PSK or 2 PSK mode.
The input signal to the first receiver 23 will now be explained in more detail referring to the vector diagram of FIG. 20. The 4 PSK signal received by the first receiver 23 from the digital transmitter 51 is expressed in an ideal form without transmission distortion and noise, using four signal points 151, 152, 153, and 154, as shown in FIG. 20.
In practice, the real four signal points appear in particular extended areas about the ideal signal positions 151, 152, 153, and 154 respectively due to noise, amplitude distortion, and phase error developed during transmission. If one signal point is unfavorably displaced from its original position, it will hardly be distinguished from its neighboring signal point and the error rate will thus be increased. As the error rate increases to a critical level, the reproduction of data becomes less accurate. For enabling the data reproduction at a maximum acceptable level of the error rate, the distance between any two signal points should be far enough to be distinguished from each other. If the distance is 1AR0, the signal point 151 of a 4 PSK signal close to a critical error level has to stay in a first discrimination area 155 denoted by the hatching of FIG. 20 and determined by |0−aR1|>AR0 and |0−bR1|>AR0. This allows the signal transmission system to reproduce carrier waves and thus, demodulate a wanted signal. When the minimum radius of the antenna 22 is set to r0, the transmission signal of more than a given level can be intercepted by any receiver of the system. The amplitude of a 4 PSK signal of the digital transmitter 51 shown in
Here, a case of receiving a QPSK signal will be considered. Similarly to the manner performed by the signal point setting circuit 67 in the transmitter shown in
The 16 PSK signal of the transmitter 1 will now be explained referring to the vector diagram of FIG. 9. When the horizontal vector distance A1 of the signal point 83 is greater than ATO of the minimum amplitude of the 4 PSK signal of the digital transmitter 51, the four signal points 83, 84, 85, and 86 in the first quadrant of
If the transponder of a satellite supplies an abundance of energy, the forgoing technique of 16 to 64-state QAM mode transmission will be feasible. However, the transponder of the satellite in any existing satellite transmission system is strictly limited in the power supply due to its compact size and the capability of solar batteries. If the transponder or satellite is increased in size and thus weight, its launching cost will soar. This disadvantage will rarely be eliminated by traditional techniques unless the cost of launching a satellite rocket is reduced by to a considerable level. In the existing system, a common communications satellite provides as low as 20 W of power and a common broadcast satellite offers 100 W to 200 W at best. For transmission of such a 4 PSK signal in the symmetrical 16-state QAM mode as shown in
It would be understood that the symmetrical signal state QAM technique is most effective when the receivers equipped with the same sized antennas are employed corresponding to a given transmitting power. Another novel technique will however be preferred for use with receivers equipped with different sized antennas.
In more detail, while the 4 PSK signal can be intercepted by a common low cost receiver system having a small antenna, the 16 QAM signal is intended to be received by a high cost, high quality, multiple-bit modulating receiver system with a medium or large sized antenna which is designed for providing highly valuable services, e.g. HDTV entertainment, to a particular person who invests more money. This allows both 4 PSK and 16 QAM signals, if desired, with a 64 DMA QAM to be transmitted simultaneously with the help of a small increase in the transmitting power.
For example, the transmitting power can be maintained low when the signal points are allocated at A1=A2 as shown in FIG. 10. The amplitude A(4) for transmission of 4 PSK data is expressed by a vector 96 equivalent to the square root of (A1+A2)2+(B1+B2)2. Then,
|A(4)|2=A12+B12=AT02+AT02=2AT02
|A(16)|2=(A1+A2)2+(B1+B2)2=4AT02+4AT02=8AT0
|A(16)/51 /51 A(4)|=2
Accordingly, the 16 QAM signal can be transmitted at a two times greater amplitude and a four times greater transmitting energy than those needed for the 4 PSK signal. A modified 16 QAM signal according to the present invention will not be demodulated by a common receiver designed for symmetrical, equally distanced signal point QAM. However, it can be demodulated with the second receiver 33 when two threshold values A1 and A2 are preset to appropriate values. In
In particular, n16 is expressed by ((A1+A2)/A1)2 which is the minimum energy for transmission of 4 PSK data. As the signal point distance suited for modified 16 QAM interception is A2, The the signal point distance for 4 PSK interception is 2A1, and the signal point distance ratio is A2/2A1, the antenna radius r2 is determined as shown in
Also, the point 102 indicates transmission of common 16 QAM at the equal distance signal state mode where the transmitting energy is nine times greater and thus will no more be practical. As apparent from the graph of
The transmitting energy at the satellite is limited to a small value and thus, n16 preferably stays not more than 5 times the value, as denoted by the hatching of FIG. 11. The point 104 within the hatching area 103 indicates, for example, that the antenna radius r2 of a two times greater value is matched with a 4× value of the transmitting energy. Also, the point 105 represents that the transmission energy should be doubled when r2 is about 5× greater. Those values are all within a feasible range.
The value of n16 not greater than 5× value is expressed using A1 and A2 as:
n16=((A1+A2)/A1)2<5
Hence, A2<1.23A1.
If the distance between any two signal point group segments shown in
The action of a modified 64 ASPK transmission will be described as the third receiver 43 can perform 64-state QAM demodulation.
r32={62/(n−1)}r12
This relationship between r3 r3 and n of a 64 QAM signal is also shown in the graphic representation of FIG. 13.
It is understood that the signal point assignment shown in
The compatibility between among the three discrete receivers can be implemented by a three-level grouping of signal points, as illustrated in
In particular, a first sub segment 181 in the first signal point group segment 91 is assigned the two-bit pattern 11 of the second data stream. Equally, a second 182, a third 183, and a fourth sub segment 184 are assigned 01, 00, and 10 of the same respectively. This assignment is identical to that shown in FIG. 7.
The signal point allocation of the third data stream will now be explained referring to the vector diagram of
As understood, the present invention permits not only transmission of six-bit data but also interception of three, two-bit, four-bit, and six-bit, different bit length data with their respective receivers while the signal compatibility remains between these levels.
The signal point allocation for providing compatibility between among the three levels will be described.
As shown in
It is necessary to space any two signal points from each other by such a distance that the sub segment signal points, e.g. 182, 183, 184, of the second data stream shown in
R32(12r1)2/(n−1) R32=(12r1)2/(n−1)
This relationship is also denoted by the curve 211 in FIG. 16. For example, if the transmitting energy is 6 or 9 times greater than that for 4 PSK transmission at the point 223 or 222, the antenna 32 having a radius of 8× or 6× value respectively can intercept the first, second, and third data streams for demodulation. As the signal point distance of the second data stream is close to 2/3A2, the relationship between r1 and r2 is expressed by:
R22=(3r1)2/(n−1) R22=(3R1)2/(n−1)
Therefore, the antenna 32 of the second receiver 33 has to be slightly increased in radius as denoted by the curve 223.
As understood, while the first and second data streams are transmitted through a traditional satellite which provides a small signal transmitting energy, the third data stream can also be transmitted through a future satellite which provides a greater signal transmitting energy without interrupting the action of the first and second receivers 23 or 33 or with no need of modification of the same and thus, both the compatibility and the advancement is ensured.
The signal receiving action of the second receiver 33 will first be described. As compared with the first receiver 23 arranged for interception with a small radius r1 antenna and demodulation of the 4 PSK modulated signal of the digital transmitter 51 or the first data stream of the signal of the transmitter 1, the second receiver 33 is adopted for perfectly demodulating the 16 signal state two-bit data, shown in
The block diagram of the second receiver 33 in
TH16=(A1+A2/2)/(A1+A2)
The various data for demodulation including A1 and A2 or TH16, and the value m for multiple-bit modulation are also transmitted from the transmitter 1 as carried in the first data stream. The demodulation controller 231 may be arranged for recovering such demodulation data through statistical process of the received signal.
A way of determining the shift factor A1/A2 will be described with reference to
FIGS. 25(a) and 25(b) are views showing signal point allocations for the C-CDM signal points, wherein signal points are added by shifting in the polar coordinate direction (r,θ). The previously described C-CDM is characterized in that the signal points are shifted in the rectangular coordinate direction, i.e. XY direction; therefore it is referred to as rectangular coordinate system C-CDM. Meanwhile, this C-CDM characterized by the shifting of signal points in the polar coordinate direction, i.e. r, θ direction, is referred to as polar coordinate system C-CDM.
FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein four signal points are added by shifting each of 4 QPSK signals in the radius r direction of the polar coordinate system. In this manner, the APSK of polar coordinate system C-CDM having 8 signal points is obtained from the QPSK as shown in FIG. 25(a). As the pole is shifted in the polar coordinate system to add signal points in this APSK, it is referred to as shifted pole-APSK, i.e. SP-APSK in the abbreviated form. In this case, coordinate values of the newly added four QPSK signals 85 are specified by using a shift factor S1 as shown in FIG. 139. Namely, 8PS-APSK signal points includes ordinary QPSK signal points 83 (r0, θ0) and a signal point ((S1+1)(r0, θ0) obtained by shifting the signal point 83 in the radius r direction by an amount of S1r0. Thus, a 1-bit subchannel 2 is obtained in addition to a 2-bit subchannel 1 identical with the QPSK.
Furthermore, as shown in the constellation diagram of
In FIG. 25(b), the signal points are allocated on the lines of θ=π/8. With this arrangement, the 16 PSK signal points are reduced or limited to 12 signal points, i.e. 3 signal points in each quadrant. With this limitation, these three signal points in each quadrant are roughly regarded as one signal point for 4 QPSK signals. Therefore, this enables the QPSK receiver to reproduce the first subchannel in the same manner as in the previous embodiment.
More specifically, the signal points are disposed on the lines of θ=π/4, θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offset by an amount ±θ in the angular direction of the polar coordinate system from the QPSK signals disposed on the lines of θ=π/4. Since all the signals are in the range of θ=π/4±π/8, they can be regarded as one of the QPSK signal points on the line of θ=π/4. Although the error rate is lowered a little bit in this case, the QPSK receiver 23 shown in
In case of the angular shift C-CDM, if signal points are disposed on the lines of π/n, the carrier wave reproduction circuit can reproduce the carrier wave by the use of an n-multiplier circuit in the same manner as in other embodiments. If the signal points are not disposed on the lines of π/n, the carrier wave can be reproduced by transmitting several pieces of carrier information within a predetermined period in the same manner as in other embodiment embodiments. Assuming that an angle between two signal points of the QPSK or 8-SP-APSK is 2θ0 in the polar coordinate system and a first angular shift factor is P1, two signal points (r0, θ0+P1θ0) and (r0, % θ0-P1θ0) are obtained by shifting the QPSK signal point in the angular θ direction by an amount ±P1θ0. Thus, the number of signal points are doubled. Thus, the 1-bit subchannel 3 can be added and is referred to as 8-SP-PSK of P=P1. If eight signal points are further added by shifting the 8-SP-PSK signals in the radius r direction by an amount S1r0, it will become possible to obtain 16-SP-APSK (P, S1 type) as shown in FIG. 142. The subchannels 1 and 2 can be reproduced by two 8PS-PSKs having the same phase. Returning to FIG. 25(b), as the C-CDM based on the angular shift in the polar coordinate system can be applied to the PSK as shown in
The system shown in FIG. 25(b) is compatible with both the rectangular and polar coordinate systems. As the signal points are disposed on the angular lines of the 16 PSK, they can be demodulated by the 16 PSK. Furthermore, as the signal points are divided into groups, the QPSK receiver can be used for demodulation. Still further, as the signal points are also allocated to suit the rectangular coordinate system, the demodulation will be performed by the 16-SRQAM. Consequently, the compatibility between the rectangular coordinate system C-CDM and the polar coordinate system C-CDM can be assured in any of the QPSK, 16PSK, and 16-SRQAM.
The demodulation controller 231 has a memory 231a for storing therein different threshold values (i.e., the shift factors, the number of signal points, the synchronization rules, etc.) which correspond to different TV broadcast channels. When receiving one of the channels again, the values corresponding to the receiving channel will be read out of the memory to thereby stabilize the reception quickly.
If the demodulation data is lost, the demodulation of the second data stream will hardly be executed. This will be explained referring to a flowchart shown in FIG. 24.
Even if the demodulation data is not available, demodulation of the 4 PSK at Step 313 and of the first data stream at Step 301 can be implemented. At Step 302, the demodulation data retrieved by the first data stream reproducing unit 232 is transferred to the demodulation controller 231. If m is 4 or 2 at Step 303, the demodulation controller 231 triggers demodulation of 4 PSK or 2 PSK at Step 313. If not, the procedure moves to Step 310. At Step 305, two threshold values TH8 and TH16 are calculated. The threshold value TH16 for AM demodulation is fed at Step 306 from the demodulation controller 231 to both the first 136 and the second discrimination/demodulation circuit 137. Hence, demodulation of the modified 16 QAM signal and reproduction of the second data stream can be carried out at Steps 307 and 315 respectively. At Step 308, the error rate is examined and if high, the procedure returns to Step 313 for repeating the 4 PSK demodulation.
As shown in
The transmitter 1 is arranged to transmit carrier timing signals at intervals of a given time with the first data stream for the purpose of compensation for no demodulation of the second data stream. The carrier timing signal enables one to identify the signal points 83 and 85 of the first data stream regardless of demodulation of the second data stream. Hence, the reproduction of carrier wave can be triggered by the transmitting of carrier data to the carrier reproducing circuit 131.
A determination then made at Step 304 of the flowchart of
TH64=(A1+A2/2)/(A1+A2)
This calculation is equivalent to that of TH16 but its resultant distance between signal points is smaller.
If the signal point distance in the first sub segment 181 is A3, the distance between the first 181 and the second sub segment 182 is expressed by (A2−2A3). Then, the average distance is (A2−2A3)/(A1+A2) which is designated as d64. when d64 is smaller than T2 which represents the signal point discrimination capability of the second receiver 33, any two signal points in the segment will hardly be distinguished from each other. This judgement is executed at Step 313. If d64 is out of a permissive range, the procedure moves back to Step 313 for 4 PSK mode demodulation. If d64 is within the range, the procedure advances to Step 305 for allowing the demodulation of 16 QAM at Step 307. If it is determined at Step 308 that the error rate is too high, the procedure goes back to Step 313 for 4 PSK mode demodulation.
When the transmitter 1 supplied a modified 8 QAM signal such as shown in FIG. 25(a) in which all the signal points are at angles of cos (2πf+π/4), the carrier waves of the signal are lengthened to the same phase and will thus be reproduced with much ease. At the time, two-bit data of the first data stream are demodulated by the 4-PSK receiver while one-bit data of the second data stream is demodulated by the second receiver 33 and the total of three-bit data can be reproduced.
The third receiver 43 will be described in more detail.
As shown in
TH164 =(A1+A3/2)/(A1+A2)
TH264=(A1+A2/2)/(A1+A2) and
TH364=(A1+A2−A3/2)/(A1+A2)
Through AM demodulation of a phase detected signal using the three threshold values, the third data stream can be reproduced like the first and second data stream explained with FIG. 21. The third data stream contains e.g. four signal points 201, 202, 203, and 204 at the first sub segment 181 shown in
The demodulation controller 231 detects the value m, A1, A2, and A3 from the demodulation data contained in the first data stream demodulated by the first data stream reproducing unit 232 and calculates the three threshold values TH164, TH264, and TH364 which are then fed to the first 136 and the second discrimination/demodulation circuit 137 so that the modified 64 QAM signal is demodulated with certainty. Also, if the demodulation data have been scrambled, the modified 64 QAM signal can be demodulated only with a specific or subscriber receiver.
The action of carrier wave reproduction needed for execution of a satisfactory demodulating procedure will now be described. The scope of the present invention includes reproduction of the first data stream of a modified 16 or 64 QAM signal using a 4 PSK receiver. However, a common 4 PSK receiver rarely reconstructs carrier waves, thus failing to perform a correct demodulation. For compensation, some arrangements are necessary at both the transmitter and receiver sides.
Two techniques for compensation are provided according to the present invention. A first technique relates to transmission of signal points aligned at angles of (2n−1)π/4 at intervals of a given time. A second technique offers transmission of signal points arranged at intervals of an angle of nπ/8.
According to the first technique, the eight signal points including 83 and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown in FIG. 38. In action, at least one of the eight signal points is transmitted during sync time slot periods 452, 453, 454, and 455 arranged at equal intervals of time in a time slot gap 451 shown in the time chart of FIG. 38. Any desired signal points are transmitted during the other time slots. The transmitter 1 is also arranged to assign a data for the time slot interval to the sync timing data region 499 of a sync data block, as shown in FIG. 41.
The content of a transmitting signal will be explained in more detail referring to FIG. 41. The time slot group 451 containing the sync time slots 452, 453, 454, and 455 represents a unit data stream or block 491 carrying a data of Dn.
The sync time slots in the signal are arranged at equal intervals of a given time determined by the time slot interval or sync timing data. Hence, when the arrangement of the sync time slots is detected, reproduction of carrier waves will be executed slot by slot through extracting the sync timing data from their respective time slots. Such a sync timing data S is contained in a sync block 493 at the front end of a data frame 492, which consists of a number of sync time slots denoted by the hatching in FIG. 41. Accordingly, the data to be extracted for carrier wave reproduction are increased, thus allowing the 4 PSK receiver to reproduce desired carrier waves at higher accuracy and efficiency.
The sync block 493 comprises sync data regions 496, 497, and 498, —containing sync data S1, S2, and S3, —respectively which include unique words and demodulation data. The phase sync signal assignment region 499 is at the end of the sync block 493, which holds a data of IT including information about interval arrangement and assignment of the sync time slots.
The signal point data in the phase sync time slot has a particular phase and can thus be reproduced by the 4 PSK receiver. Accordingly, IT in the phase sync signal assignment region 499 can be retrieved without error thus ensuring the reproduction of carrier waves at with accuracy.
As shown in
The assignment is distinguished from that of
The carrier wave reproduction of the first receiver 23 shown in
In this manner, the signal point data of the (2n−1)π/4 phase denoted by the shaded areas in
For transmission of a modified 64 QAM signal such as shown in
The foregoing carrier reproducing is of COSTAS type. A carrier reproducing circuit of the reverse modulation type will now be explained according to the embodiment.
The reproduction of a carrier wave by 16× frequency multiplication will be explained. The transmitter 1 shown in
The arrangement of the 16× multiplier circuit will be explained referring to
Similarly, a sin 8θ signal is produced from the two, sin 2θ and cos 2θ, signals by the combination of an adder circuit 667, a subtracter circuit 668, and a multiplier 670. Furthermore, a sin 16θ signal is produced by the combination of an adder circuit 671, a subtractor circuit 672, and a multiplier 673. Then, the 16× multiplication is completed.
Through the foregoing 16× multiplication, the carrier wave of all the signal points of the modified 16 QAM signal shown in
However, reproduction of the carrier wave of the modified 64 QAM signal shown in FIG. 47 can involve an increase in the error rate due to dislocation of some signal points from the sync areas 471.
Two techniques are known for compensation for the consequences. One is inhibiting transmission of the signal points dislocated from the sync areas. This causes the total amount of transmitted data to be reduced but allows the arrangement to be facilitated. The other is providing the sync time slots as described in FIG. 38. In more particular, the signal points in the nπ/8 sync phase areas, e.g. 471 and 471a, are transmitted during the period of the corresponding sync time slots in the time slot group 451. This triggers an accurate synchronizing action during the period thus minimizing phase error.
As now understood, the 16× multiplication allows the simple 4 PSK receiver to reproduce the carrier wave of a modified 16 or 64 QAM signal. Also, the insertion of the sync time slots causes the phasic accuracy to be increased during the reproduction of carrier waves from a modified 64 QAM signal.
As set forth above, the signal transmission system of the present invention is capable of transmitting a plurality of data on a single carrier wave simultaneously in the multiple signal level arrangement.
More specifically, three different level receivers which have discrete characteristics of signal intercepting sensitivity and demodulating capability are provided in relation to one single transmitter so that any one of them can be selected depending on a wanted data size to be demodulated which is proportional to the price. When the first receiver of low resolution quality and low price is acquired together with a small antenna, its owner can intercept and reproduce the first data stream of a transmission signal. When the second receiver of medium resolution quality and medium price is acquired together with a medium antenna, its owner can intercept and reproduce both the first and second data streams of the signal. When the third receiver of high resolution quality and high price is acquired with a large antenna, its owner can intercept and reproduce all the first, second, and third data streams of the signal.
If the first receiver is a home-use digital satellite broadcast receiver of low price, it will overwhelmingly be welcome by a majority of viewers. The second receiver accompanied with the medium antenna costs more and will be accepted by not common viewers but particular people who want to enjoy HDTV services. The third receiver accompanied with the large antenna at least before the satellite output is increased, is not appropriate for home use and will possibly be used in relevant industries. For example, the third data stream carrying super HDTV signals is transmitted via a satellite to subscriber cinemas which can thus play video tapes rather than traditional movie films and run movies at low cost.
When the present invention is applied to a TV signal transmission service, three different quality pictures are carried on one signal channel wave and will offer compatibility with each other. Although the first embodiment refers to a 4 PSK, a modified 8 QAM, a modified 16 QAM, and a modified 64 QAM signal, other signals will also be employed with equal success including a 32 QAM, a 256 QAM, an 8 PSK, and a 16 PSK, and a 32 PSK signal. It would be understood that the present invention is not limited to a satellite transmission system and can be applied to a terrestrial communications system or a cable transmission system.
The transmission method of the invention can also be applied to a 4-level or 8-level ASK signal as shown in FIG. 58 and FIGS. 68(a) and (b), respectively.
EMBODIMENT 2A second embodiment of the present invention is featured in which the physical multi-level arrangement of the first embodiment is divided into small levels through e.g. discrimination in error correction capability, thus forming a logic multi-level construction. In the first embodiment, each multi-level channel has different levels in the electrical electric signal amplitude or physical demodulating capability. The second embodiment offers different levels in the logic reproduction capability such as error correction. For example, the data D1 in a multi-level channel is divided into two, D1-1 and D1-2, components and D1-1 is more increased in the error correction capability than D1-2 for discrimination. Accordingly, as the error detection and correction capability is different between D1-1 and D1-2 at demodulation, D1-1 can successfully be reproduced within a given error rate when the C/N level of an original transmitting signal is as low as disenabling the reproduction of D1-2. This will be implemented using the logic multi-level arrangement.
More specifically, the logic multi-level arrangement consists of dividing data of a modulated multi-level channel and discriminating distances between error correction codes by mixing error correction codes with product codes for varying error correction capability. Hence, a more multi-level signal can be transmitted.
In fact, a D1-1 channel is divided into two sub channels D1 and D1-2 and a D2 channel is divided into two sub channels D2-1 and D2-2.
This will be explained in more detail referring to FIG. 87 85 in which D1-1 is reproduced from a lowest C/N signal. If the C/N rate is d at minimum, three components D1-2, D2-1 and D2-2 cannot be reproduced while D1-1 is reproduced. If C/N is not less than c, D1-2 can also be reproduced. Equally, when C/N is b, D2-1 is reproduced and when C/N is a, D2-2 is reproduced. As the C/N rate increases, the reproducible signal levels are increased in number. The lower the C/N, the fewer the reproducible signal levels. This will be explained in the form of relationship between transmitting distance and reproducible C/N value referring to FIG. 86. In common, the C/N value of a received signal is decreased in proportion to the distance of transmission as expressed by the real line 861 in FIG. 86. It is now assumed that the distance from a transmitter antenna to a receiver antenna is La when C/N=a, Lb when C/N=b, Lc when C/N=c, Ld when C/N=d, and Le when C/N=e. If the distance from the transmitter antenna is greater than Ld, D1-1 can be reproduced as shown in
The construction of the logic multi-level arrangement will be described in which there are provided two physical levels and two logic levels.
The main ECC encoder 872a has a higher error correction capability than that of the sub ECC encoder 873a. Hence, D1-1 can be reproduced at a lower rate of C/N than D1-2 as apparent from the CN-level diagram of FIG. 85. More particularly, the logic level of D1-1 is less affected by declination of the C/N than that of D1-2. After error correction code encoding, D1-1 and D2-2 D1-2 are summed by a summer 874a to a D1 signal which is then transferred to the modulator 4. The other two signals D2-1 and D2-2 of the divider circuit 3 are error correction encoded by two, main and sub, ECC encoders 872 b 872b and 873b of a second ECC encoder 871b respectively and then, summed by a summer 874b to a D2 signal which is transmitted to the modulator 4. The main ECC encoder 872b is higher in the error correction capability than the sub ECC encoder 873b. The modulator 4 in turn produces from the two, D1 and D2, input signals a multi-level modulated signal which is further transmitted from the transmitter unit 5. As understood, the output signal from the transmitter 1 has two physical levels D1 and D2 and also, four logic levels D1-1, D1-2, D2-1, and D2-2 based on the two physical levels for providing different error correction capabilities.
The reception of such a multi-level signal will be expelained explained.
As shown in
Since D1-1 and D2-2 are higher in the error correction capability than D1-2 and D2-2 respectively, the error rate remains less than a given value although C/N is fairly low as shown in FIG. 85 and thus, an original signal will be reproduced successfully.
The action of discriminating the error correction capability between the main ECC decoders 877a and 877b of high code gain and the sub ECC decoders 878a and 878b of low code gain will now be described in more detail. It is a good idea for having a difference in the error correction capability, i.e., in the code gain, to use in the sub ECC decoder a common coding technique, e.g. Reed-Solomon or BCH method, as shown in FIG. 165(b) for the ECC decoder, having a standard code distance and in %he the main ECC decoder, another encoding technique in which distance between correction codes is increased using Reed-Solomon codes, their product codes, or other long-length codes or a trellis decoder 744p, 744q, and 744r shown in FIGS. 128(d), 128(e), 128(f). A variety of known techniques for increasing the error correction code distance have been introduced and will not be explained in detail. The present invention can be associated with any known technique for having the logic multi-level arrangement.
Also, as shown in the block diagram of
The logic multi-level arrangement will be explained in conjuction conjunction with a diagram of
As the C/N rate of an input signal decreases, the error rate increases after error correction. If C/N is lower than a given value, the error rate exceeds a reference value Eth determined by the system design standards and no original data will normally be reconstructed. When C/N is lowered to less than e, the D1 signal fails to be reproduced as expressed by the line 881 of D1-1 in FIG. 89. When e≦C/N<d, D1-1 of the D1 signal exhibits a higher error rate than Eth and will not be reproduced.
When C/N is d at the point 885d, D1-1 having a higher error correction capability than D1-2 becomes not higher in the error rate than Eth and can be reproduced. At the time, the error rate of D1-2 remains higher than Eth after error correction and will no longer be reproduced.
When C/N is increased up to c at the point 885c, D1-2 becomes not higher in the error rate than Eth and can be reproduced. At the time, D2-1 and D2-2 remain in no demodulation state. After the C/N rate is increased further to b′, the D2 signal becomes ready to be demodulated.
When C/N is increased to b at the point 885b, D2-1 of the D2 signal becomes not higher in the error rate than Eth and can be reproduced. At the time, the error rate of D2-2 remains higher than Eth and will not be reproduced. When C/N is increased up to a at the point 885a, D2-2 becomes not higher than Eth and can be reproduced.
As described above, the four different signal logic levels divided from two, D1 and D2, physical levels through discrimination of the error correction capability between the levels, can be transmitted simultaneously.
Using the logic multi-level arrangement of the present invention with a multi-level construction in which at least a part of the original signal is reproduced even if data in a higher level is lost, digital signal transmission will successfully be executed without losing the advantageous effect of an analogue signal transmission in which transmitting data is gradually decreased as the C/N rate becomes low.
Thanking Thanks to up-to-date compression techniques, compressed image data can be transmitted in the logic multi-level arrangement for enabling a receiver station to reproduce a higher quality image than that of an analogue system and also, with not sharply but at steps declining the signal level for ensuring signal interception in a wider area. The present invention can provide an extra effect of the multi-layer arrangement which is hardly implemented by a known digital signal transmission system without deteriorating high quality image data.
In addition, the address data of the image segment data, the base image data for image compression, the scramble cancellation data shown in the descrambler (FIG. 66), and high priority (HP) data, i.e., the data (e.g., the frame synchronization signal and header) that is most essential to image expansion of the HDTV signal, is transmitted as D1-1 by the high code gain ECC encoder 743a (
This high priority data is protected because the error rate of priority data D1-1 does not increase noticeably. Fatal deterioration of the characteristic image quality of digital video transmissions is thus avoided, and a “graceful degradation” effect whereby image quality gradually deteriorates is obtained. The modulator 749 and demodulator 760 of
Furthermore, as shown in the block diagrams of
As a result, even if the C/N ratio of the transmission system deteriorates significantly, the high priority data can be received. Therefore, while the image quality deteriorates with the deterioration of the low priority data, the high priority data can also be reproduced in applications subject to severe C/N ratio deterioration, as found in the reception conditions encountered with mobile television receivers, and the pixel block positioning information is also reproduced. Because image block destruction is thus prevented, viewers are still able to receive and view broadcast programing under extremely poor reception conditions.
EMBODIMENT 3A third embodiment of the present invention will be described referring to the relevant drawings.
Equally, other three input video signals 406, 407, and 408 are fed to a second 409, a third 410, and a fourth video encoder 411 respectively which all are arranged identical in construction to the first video encoder 401 for data compression.
The four first data streams from their respective encoders 401, 409, 410, and 411 are transferred to a first multiplexer 413 of a multiplexer 412 where they are time multiplexed by a TDM process into a first data stream multiplex signal which is fed to a transmitter 1.
A part or all of the four second data streams from their respective encoders 401, 409, 410, and 411 are transferred to a second multiplexer 414 of the multiplexer 412 where they are time multiplexed to a second data stream multiplex signal which is then fed to transmitter 1. Also, a part or all of the four third data streams are transferred to a third multiplexer 415 where they are time multiplexed to a data stream multiplex signal which is then fed to the transmitter 1.
The transmitter 1 performs modulation of the three data stream signals with its modulator 4 by the same manner as described in the first embodiment. The modulated signals are sent from a transmitter unit 5 through an antenna 6 and an uplink 7 to a transponder 12 of a satellite 10 which in turn transmits it to three different receivers including a first receiver 23.
The modulated signal transmitted through a downlink 21 is intercepted by a small antenna 22 having a radius r1 and fed to a first data stream reproducing unit 232 of the first receiver 23 where its first data stream only is demodulated. The demodulated first data stream is then converted by a first video decoder 421 to a traditional 425 or wide-picture NTSC or video output signal 426 of low image resolution.
Also, the modulated signal transmitted through a downlink 31 is intercepted by a medium antenna 32 having a radius r2 and fed to a first 232 and a second data stream reproducing unit 233 of a second receiver 33 where its first and second data streams are demodulated respectively. The demodulated first and second data streams are then summed and converted by a second video decoder 422 to an HDTV or video output signal 427 of high image resolution and/or to the video output signals 425 and 426.
Also, the modulated signal transmitted through a downlink 41 is intercepted by a large antenna 42 having a radius r3 and fed to a first 232, a second 233, and a third data steam reproducing unit 234 of a third receiver 43 where its first, second, and third data streams are demodulated respectively. The demodulated first, second, and third data streams are then summed and converted by a third video decoder 423 to a super HDTV or video output signal 428 of super high image resolution for use in a video theater or cinema. The video output signals 425, 426, and 427 can also be reproduced if desired. A common digital TV signal is transmitted from a conventional digital transmitter 51 and when intercepted by the first-receiver 23, will be converted to the video output signal 426 such as a low resolution NTSC TV signal.
The first video encoder 401 will now be explained in more detail referring to the block diagram of FIG. 30. An input video signal of super high resolution is fed through the input unit 403 to the divider circuit 404 where it is divided into four components by sub-band coding process. In particular, the input video signal is separated by passing through a horizontal lowpass filter 451 and a horizontal highpass filter 452 of e.g. QAM mode to two, low and high, horizontal frequency components which are then subsampled into half of their quantities by two subsamplers 453 and 454 respectively. The low horizontal component is filtered by a vertical lowpass filter 455 and a vertical highpass filter 456 into a low horizontal low vertical component or HLVL signal and a low horizontal high vertical component for HLVH signal respectively. The two, HLVL and HLVH, signals are then subsampled into one half by two subsamblers 457 and 458 respectively and transferred to the compressing circuit 405.
The high horizontal component is filtered by a vertical lowpass filter 459 and a vertical highpass filter 460 into a high horizontal low vertical highpass component or HHVL signal and a high horizontal high vertical component or HHVH signal respectively. The two, HHVL and HHVH, signals are then subsampled into one half by two subsamplers 461 and 462 respectively and transferred to the compressing circuit 405.
The HLVL signal is preferably DCT compressed by a first compressor 471 of the compressing circuit 405 and fed to a first output circuit 472 as the first data stream.
Also, the HLVH signal is compressed by a second compressor 473 and fed to a second output circuit 464. The HHVL signal is compressed by a third compressor 463 and fed to the second output circuit 464.
The HHVH signal is divided by a divider 465 into two high resolution (HHVH1) and super high resolution (HHVH2) video signals which are then transferred to the second output circuit 464 and a third output circuit 468 respectively.
The first video decoder 421 will now be explained in more detail referring to FIG. 31. The first data stream or D1 signal of the first receiver 23 is fed through an input unit 501 to a descrambler 502 of the first video decoder 421 where it is descrambled. The descrambled D1 signal is expanded by an expander 503 to HLVL which is then fed to an aspect ratio changing circuit 504. Thus, the HLVL signal can be delivered through an output unit 505 as a standard 500, letterbox format 507, wide-screen 508, or sidepanel format NTSC signal 509. The scanning format may be of non-interlace or interlace type and its NTSC mode lines may be 525 or doubled to 1050 by double tracing. When the received signal from the digital transmitter 51 is a digital TV signal of 4 PSK mode, it can also be converted by the first receiver 23 and the first video decoder 421 to a TV picture. The second video decoder 422 will be explained in more detail referring to the block diagram of FIG. 32. The D1 signal of the second receiver 33 is fed through a first input 521 to a first expander 522 for data expansion and then, transferred to an oversampler 523 where it is sampled at 2x. The oversampled signal is filtered by a vertical lowpass filter 524 into HLVL. Also, the D2 signal of the second receiver 33 is fed through a second input 530 to a divider 531 where it is divided into three components which are then transferred to second, third, and fourth expanders 532-534 respectively for data expansion. The three expanded components are sampled at 2x by three oversamplers 535, 536, and 537 and filtered by a vertical highpass 538, a vertical lowpass 539, and a vertical highpass filter 540 respectively. Then, HLVL from the vertical lowpass filter 524 and HLVH from the vertical highpass filter 538 are summed by an adder 525, sampled by an oversampler 541, and filtered by a horizontal lowpass filter 542 into a low frequency horizontal video signal. HHVL from the vertical lowpass filter 539 and HHVH1 from the vertical highpass filter 540 are summed by an adder 526, sampled by an oversampler 544, and filtered by a horizontal highpass filter 545 to a high frequency horizontal video signal. The two, high and low frequency, horizontal video signals are then summed by an adder 543 into a high resolution video signal HD which is further transmitted through an output unit 546 as a video output 547 of e.g. HDTV format. If desired a traditional NTSC video output can be reconstructed with equal success.
The action of multiplexing in the multiplexer 412 shown in
The selection of the first channel TV signal will now be described. When intercepted by the first receiver 23 with a small antenna coupled to the first video decoder 421, the first channel signal is converted to a standard or widescreen NTSC TV signal as shown in FIG. 31. When intercepted by the second receiver 33 with a medium antenna coupled to the second video decoder 422, the signal is converted by summing L1 of the first data stream D1 assigned into the domain 601 and M1 of the second data stream D2 assigned to the domain 602 to an HDTV signal of the first channel equivalent in program to the NTSC signal.
When intercepted by the third receiver 43 with a large antenna coupled to the third video decoder 423, the signal is converted by summing L1 of D1 assigned to the domain 601, M1 of D2 assigned to the domain 602, and H1 of D3 assigned to the domain 603 into a super HDTV signal of the first channel equivalent in program to the NTSC signal. The other channel signals can be reproduced in an equal manner.
The foregoing data assignment makes the use of one of two, horizontal and vertical, polarization planes of a transmission wave. When both the horizontal and vertical polarization planes are used, the frequency utilization will be doubled. This will be explained below.
As set forth above, the compatible digital TV signal transmission system of the third embodiment can offer three, super HDTV, HDTV, and conventional NTSC, TV broadcast services simultaneously. In addition, a video signal intercepted by a commercial station or cinema can be electronized.
The modified QAM of the embodiments is now termed as SRQAM and its error rate will be examined.
First, the error rate in 16 SRQAM will be calculated.
The signal point 83a is spaced δ from both the I-axis and the Q-axis of the coordinate. It is now assumed that n is a shift value of the 16 SRQAM. In 16 SRQAM, the signal point 83a of 16 QAM is shifted to a signal point 83 where the distance from each axis is nδ. The shift value n is thus expressed as:
0<n<3.
The other signal points 84a and 86a are also shifted to two points 84 and 86 respectively.
If the error rate of the first data stream is Pe1, it is obtained from:
Also, the error rate Pe2 of the second data stream is obtained from:
The error rate of 36 or 32 SRQAM will be calculated.
The signal point 83a of 36 QAM is spaced δ from each axis of the coordinate. It is now assumed that n is a shift value of the 16 SRQAM. In 36 SRQAM, the signal point 83a is shifted to a signal point 83 where the distance from each axis is nδ. Similarly, the nine 36 QAM signal points in the first quadrant are shifted to points 83, 84, 85, 86, 97, 98, 99, 100, 101 respectively. If a signal point group 90 comprising the nine signal points is regarded as a single signal point, the error rate Pe1 in reproduction of only the first data stream D1 with a modified 4 PSK receiver and the error rate Pe2 in reproduction of the second data stream D2 after discriminating the nine signal points of the group 90 from each other, are obtained respectively from:
The curve 902a represents a D2 level SRQAM signal at n=1.5 which can be reproduced at the error rate of 10−1.5 only when its C/N rate is 2.5 dB higher than that of the conventional 32 QAM of the curve 900. Also, the curves 901b and 902b represent D1 and D2 SRQAM signals at n=2.0 respectively. The curves curve 902c represents a D2 SRQAM signal at n=2.5. It is apparent that the C/N rate of the SRQAM signal at the error data of 10−1.5 is 5 dB, 8 dB, and 10 dB higher at n=1.5, 2.0, and 2.5 respectively in the D1 level and 2.5 dB lower in the D2 level than that of a common 32 QAM signal.
Shown in
The C/N rate of the first and second data streams D1, D2 of a 16 SRQAM signal is shown in
One example of propagation of SRQAM signals of the present invention will now be described for use with a digital TV terrestrial broadcast service.
The C/N rate varies 5 dB under a worst case receiving condition such as bad weather. If a change in the relevant condition, e.g. weather, attenuates the C/N rate, the interception of an HDTV signal will hardly be ensured. Also, geographical conditions largely affect the propagation of signals and a decrease of about 10 dB at least will be unavoidable. Hence, successful signal interception within 60 miles will never be guaranteed and above all, a digital signal will be harder to propagate than an analogue signal. It would be understood that the service area of a conventional digital TV broadcast service is less dependable.
In case of the 32 SRQAM signal of the present invention or the 8-VSB shown in
For common 32 QAM signal, the 60-mile-radius service area can be established theoretically. The signal level will however be attenuated by geographical or weather conditions and particularly, considerably declined at near the limit of the service area.
If the low frequency band TV component of MPEG1 grade is carried on the 1-1 level D1-1 data and the medium frequency band TV component of NTSC grade on the 1-2 level D1-2 data and high frequency band TV component of HDTV on the second level D2 data, the service area of the 32 SRQAM signal of the present invention is increased by 10 miles in radius for reception of an EDTV signal of medium resolution grade and 18 miles for reception of an LDTV signal of low resolution grade although decreased by 5 miles for reception of an HDTV signal of high resolution grade, as shown in FIG. 106.
More particularly, the medium resolution component of a digital TV broadcast signal of the SRQAM mode of the preset invention can successfully be intercepted in an unfavorable service region or shadow area where a conventional medium frequency band TV signal is hardly propagated and attenuated due to obstacles. Within at least the predetermined service area, the NTSC TV signal of the SRQAM mode can be intercepted by any traditional TV receiver. As the shadow or signal attenuating area developed by building structures and other obstacles or by interference of a neighbor analogue TV signal or produced in a low land is decreased to a minimum, TV viewers or subscribers will be increased in number.
Also, the HDTV service can be appreciated by only a few viewers who afford to have a set of high cost HDTV receiver and display, according to the conventional system. The system of the present invention allows a traditional NTSC, PAL, or SECAM receiver to intercept a medium resolution component of the digital HDTV signal with the use of an additional digital tuner. A majority of TV viewers can hence enjoy the service at less cost and will be increased in number. This will encourage the TV broadcast business and create an extra social benefit.
Furthermore, the signal receivable area for medium resolution or NTSC TV service according to the present invention is increased about 36% at n=2.5, as compared with the conventional system, . As the service area thus the number of TV viewers is increased, the TV broadcast business enjoys an increasing profit. This reduces a risk in the development of a new digital TV business which will thus be encouraged to put into practice.
This advantage is given when:
n>1.0
Hence, if the 32 SRQAM signal is selected, the shift n is determined by:
1<n<5
Also, if the 16 SRQAM signal is employed, n is determined by:
1<n<3
In the SRQAM mode signal terrestrial broadcast service in which the first and second data levels are created by shifting corresponding signal points as shown in
In the above embodiments, the low and high frequency band components of a video signal are transmitted as the first and second data streams. However, the transmitted signal may be an audio signal. In this case, low frequency or low resolution components of an audio signal may be transmitted as the first data stream, and high frequency or high resolution components of the audio signal may be transmitted as the second data stream. Accordingly, it is possible to receive high C/N portion in high sound quality, and low C/N portion in low sound quality. This can be utilized in PCM broadcast, radio, portable telephone and the like. In this case, the broadcasting area or communication distance can be expanded as compared with the conventional systems.
Furthermore, the third embodiment can incorporate a time division multiplexing (TDM) system as shown in FIG. 133. Utilization of the TDM makes it possible to increase the number of subchannels. An ECC encoder 743a and ECC encoder 743b, provided in two subchannels, differentiate ECC code gains so as to make a difference between thresholds of these two subchannels, whereby an increase in the number of channels of the multi-level signal transmission can be realized. In this case, it is also possible to provide the ECC encoder, such as two Trellis encoders 743a and 743b for VSB-ASK signals of 4 VSB, 8 VSB, 16 VSB as shown in FIG. 137 and differentiate their code gains. The explanation of this block diagram is substantially identical to that of later described block diagram of
The up converter of the transmitter and the down converter of the receiver of the transmission apparatus can be substituted for the magnetic head recording signal amplifier circuit and the magnetic head reproducing signal amplifier circuit of the magnetic recording and reproducing apparatus, respectively, and there respective components are therefore identically constructed. The configuration and operation of the modulator and demodulator of the magnetic recording and reproducing apparatus are therefore also identical to those of the transmission apparatus. Similarly, the recording/reproducing/transmission system shown in
In a simulation of
An SRQAM is the system applying a C-CDM (Constellation-Code Division Multiplex) of the present invention to a rectangle-QAM. A C-CDM, which is a multiplexing method independent of TDM or FDM, can obtain subchannels by dividing a constellation-code corresponding to a code. An increase of the number of codes will bring an expansion of transmission capacity, which is not attained by TDM or FDM alone, while maintaining almost perfect compatibility with conventional communication apparatus. Thus C-CDM can bring excellent effects.
Although above embodiment combines the C-CDM and the TDM, it is also possible to combine the C-CDM with the FDM (Frequency Division Multiplex) to obtain similar modulation effect of threshold values. Such a system can be used for a TV broadcasting, and FIGS. 108(a)-108(e) show shows a frequency distribution of a TV signal. A spectrum 725 represents a frequency distribution of a conventional analogue, e.g. NTSC, broadcasting signal. The largest signal is a video carrier 722. A color carrier 723 and a sound carrier 724 are not so large. There is known a method of using an FDM for dividing a digital broadcasting signal into two frequencies. In this case, a carrier is divided into a first carrier 726 and a second carrier 727 to transmit a first 720 and a second signal 721 respectively. Interference can be lowered by placing first and second carriers 726 and 727 sufficiently far from the video carrier 722. The first signal 720 serves to transmit a low resolution TV signal at a large output level, while the second signal 721 serves to transmit a high resolution TV signal at a small output level. Consequently, the multi-level signal transmission making use of an FDM can be realized without being bothered by obstruction.
However, the present invention resolves this problem. According to the present invention, the first signal 720 is given by 32 SRQAM mode which is obtained through C-CDM modulation so that the subchannel A is divided into two subchannels 1 of A and 2 of A. The newly added subchannel 1 of A, having a lowest threshold value, carries a low resolution component. The second signal 721 is also given by 32 SRQAM mode, and a threshold value for the subchannel 1 of B is equalized with the threshold 2.
With this arrangement, the region in which a transmitted signal is not received when the signal level decreases below the threshold 2 is reduced to a shaded portion of the second signal 721a in FIG. 108. As the subchannel 1 of B and the subchannel A are both receivable, the transmission amount is not so much reduced in total. Accordingly, a better picture quality is reproduced even in the second level at the signal level of the threshold 2.
By transmitting a normal resolution component in one subchannel, it becomes possible to increase the number of multiple level and expand a low resolution service area. This low-threshold subchannel is utilized for transmitting important information such as sound information, sync information, headers of respective data, because these information carried on this low-threshold subchannel can be surely received. Thus stable reception is feasible. If a subchannel is newly added in the second signal 721 in the same manner, the number of levels of multi-level transmission can be increased in the service area. In the case where an HDTV signal has 1050 scanning lines, an a new service area equivalent to 775 lines can be provided in addition to 525 lines.
Accordingly, the combination of the FDM and the C-CDM realizes an increase of service area. Although above embodiment divides a subchannel into two, it is needless to say it is also possible to divide it into three or more parts.
Next, a method of avoiding obstruction by combining the TDM and the C-CDM will be explained. As shown in
For example, the error rate of the subchannel 1 of 8PS-APSK, explained in the embodiment 1 with reference to
The error rate of the subchannel 2 is expressed as follows:
Furthermore, the error rate of the subchannel 1 of 16-PS-APSK (PS type), explained with reference to
The error rate of the subchannel 2 is expressed as follows:
The error rate of the subchannel 3 is expressed as follows:
A fourth embodiment of the present invention will be described referring to the relevant drawings.
The signal transmission system of the present invention allows the first receiver 23 equipped with the antenna 22a, which is located at a far distance as shown in
Also, the second receiver 33 with the antenna 32a is located at a medium distance from the antenna 6a and can thus intercept and demodulate both first and second data streams or D1 and D2 components of the modified 16 or 64 QAM signal to an HDTV video signal which in turn produces an HDTV program picture.
The third receiver 43 with the antenna 42a is located at a near distance and can intercept and demodulate the first, second, and third data streams or D1, D2, and D3 components of the modified 16 or 64 QAM signal to a super HDTV video signal which in turn produces a super HDTV picture in quality to a common movie picture.
The assignment of frequencies is determined by the same manner as of the time division multiplexing shown in
As understood, the system allows three different but compatible digital TV signals to be carried on a single channel or using D2 and D3 regions of other channels. Also, the medium resolution TV picture data of each channel can be intercepted in a wider service area according to the present invention.
A variety of terrestrial digital TV broadcast systems employing a 16 QAM HDTV signal of 6 MHz bandwidth have been proposed. Those are however not compatible with the existing NTSC system and thus, have to be associated with a simulcast technique for transmitting NTSC signals of the same program on another channel. Also, such a common 16 QAM signal limits a service area. The terrestrial service system of the present invention allows a receiver located at a relatively far distance to intercept successfully a medium resolution TV signal with no use of an additional device nor an extra channel.
FIG. 52. shows an interference region of the service area 702 of a conventional terrestrial digital HDTV broadcast station 701. As shown, the service area 702 of the conventional HDTV station 701 intersects with the service area 712 of a neighboring analogue TV station 711. At the intersecting region 713, an HDTV signal is attenuated by signal interference from the analogue TV station 711 and will thus be intercepted with less consistency.
In the service area of the digital TV station 701, there are three interference regions developed by signal interference from the analogue TV station 711. Both HDTV and NTSC signals can hardly be intercepted in the first region 705. Although fairly interfered, an NTSC signal may be intercepted at an equal level in the second region 706 denoted by the left down hatching. The NTSC signal is carried on the first data stream which can be reproduced at a relatively low C/N rate and will thus be minimum minimally affected when the C/N rate is declined by signal interference from the analogue TV station 711.
At the third region 707 denoted by the right down hatching, an HDTV signal can also be intercepted when signal interference is absent while the NTSC signal can constantly be intercepted at a low level.
Accordingly, the overall signal receivable area of the system will be increased although the service area of HDTV signals becomes a little bit smaller than that of the conventional system. Also, at the signal attenuating regions produced by interference from a neighboring analogue TV station, NTSC level signals of an HDTV program can successfully be intercepted as compared with the conventional system where no HDTV program is viewed in the same area. The system of the present invention much reduces the size of signal attenuating area and when increases the energy of signal transmission at a transmitter or transponder station, can extend the HDTV signal service area to an equal size to the conventional system. Also, NTSC level signals of a TV program can be intercepted more or less in a far distance area where no service is given by the conventional system or a signal interference area caused by an adjacent analogue TV station.
Although the embodiment employs a two-level signal transmission method, a three-level method such as shown in
The description will now be continued, provided that no digital TV station should cause a signal interference to any neighboring analogue TV station. According to a novel system under consideration in U.S.A., no-use channels of the existing service channels are utilized for HDTV and thus, digital signals must not interfere with analogue signals. For the purpose, the transmitting level of a digital signal has to be decreased lower than that shown in FIG. 53. If the digital signal is of conventional 16 QAM or 4 PSK mode, its HDTV service area 708 becomes decreased as the signal interference region 713 is fairly large as shown in FIG. 54. This results in a less number of viewers and sponsors, whereby such a digital system will have much difficulty in operating as a profitable business.
When the level of signals is equal, the multi-level transmission system of the present invention provides a smaller HDTV service area and a greater NTSC service area for interception of an HDTV program at an NTSC signal level. Accordingly, the overall service area of each station is increased and more viewers can enjoy its TV broadcasting service. Furthermore, HDTV/NTSC compatible TV business can be operated with economical advantages and consistency. It is also intended that the level of a transmitting signal is increased when the control on averting signal interference to neighboring analogue TV stations is lessened corresponding to a sharp increase in the number of home-use digital receivers. Hence, the service area of HDTV signals will be increased and in this respect, the two different regions for interception of HDTV/NTSC and NTSC digital TV signal levels respectively, shown in
At the intersecting region 714 between two service areas of their respective stations, the received signal can be reproduced not to an HDTV level picture using a common directional antenna due to signal interference but to an NTSC level picture with a particular directional antenna directed towards a desired TV station. If a highly directional antenna is used, the received signal from a target station will be reproduced as an HDTV picture. The low resolution signal receivable area 704 is increased larger than the analogue TV system service area 702 and a pair of intersecting regions 715 and 716 developed by the two low resolution signal receivable areas 704 and 704a of their respective digital TV stations 701 and 701a permit the received signal from an antenna directed to one of the two stations to be reproduced as an NTSC level picture.
The HDTV service area of the multi-level signal transmission system of the present invention itself will be much increased when applicable signal restriction rules are withdrawn in a coming digital TV broadcast service maturity time.
At the time, the system of the present invention also provides as a wide HDTV signal receivable area as of the conventional system and particularly, allows its transmitting signal to be reproduced at an NTSC level in a further distance or intersecting areas where TV signals of the conventional system are hardly intercepted. Accordingly, signal attenuating or shadow regions in the service area will be minimized.
EMBODIMENT 5A fifth embodiment of the present invention resides in amplitude modulation or ASK procedure.
In
This is expressed as:
Lo>L
Hence, the multi-level signal transmission system of the embodiment is based on L0>L. The embodiment is however not limited to L0>L, and L0=L will be employed temporarily or permanently depending on the requirements of design, condition, and setting. In the case of VSB, the constellation shown in FIGS. 68(a) and (b) are taken.
The two signal point group groups are assigned one-bit patterns of the first data stream D1, as shown in FIG. 59(a). More particularly, a bit 0 of binary system is assigned to the first signal point group 725 and another bit 1 to the second signal point group 726. Then, a one-bit pattern of the second data stream D2 is assigned to each signal point. For example, the two signal points 721 and 723 are assigned D2=0 and the other two signal points 722 and 724 are assigned D2=1.
The multi-level signal transmission of the present invention can be implemented in an ASK mode using the foregoing signal point assignment. The system of the present invention works in the same manner as of a conventional equal signal point distance technique when the signal to noise ratio or C/N rate is high. If the C/N rate becomes low and no data can be reproduced by the conventional technique, the present system ensures reproduction of the first data stream D1 but not the second data stream D2. In more detail, the state at a low C/N is shown in
Accordingly, the two different level data D1 and D2 can be transmitted simultaneously. More particularly, both the first and second data streams D1 and D2 of a given signal transmitted through the multi-level transmission system can be reproduced at the area where the C/N rate is high and the first data stream D1 only can be reproduced in the area where the C/N rate is low.
The waveform of the ASK signal after filtering will now be examined. FIG. 62(a) shows a frequency spectrum of the ASK modulated signal in which two sidebands are provided on both sides of the carrier frequency band. One of the two sidebands is eliminated by the filter 474 to produce a signal 749 which contains a carrier component as shown in FIG. 62(b). The signal 749 is a VSB signal and if the modulation frequency band is f0, will be transmitted in a frequency band of about f0/2. Hence, the frequency utilization becomes high. Using VSB mode transmission, the ASK signal of two bit per symbol shown in
The transmission of a TV signal using such a transmitter and a receiver will be explained.
The input TV signal of the embodiment is an HDTV signal and HLVL becomes a wide-screen NTSC signal. If the aspect ratio of an available display is 16:9, HLVL is directly delivered through an output unit as a 16:9 video output 426. If the display has an aspect ratio of 4:3, HLVL is shifted by the aspect ratio changing circuit 779 to a letterbox or sidepanel format and then, delivered from the output unit 780 as a corresponding format video output 425.
The second data stream D2 fed from the second data stream output 759 to the summer 778 is summed with the output of the divider 777 to a sum signal which is then fed to the second input 531 of the second video decoder 422. The sum signal is further transferred to a divider circuit 531 while it is divided into three compressed forms of HLVH, HHVL, and HHVH. The three compressed signals are then fed to a second 535, a third 536, and a fourth expander 537 respectively for converting by expansion to HLVH, HHVL, and HHVH, of the original length. The three signals are summed with HLVL, by the video mixer 548 to a composite HDTV signal which is fed through an output 546 of the second video decoder to the output unit 780. Finally, the HDTV signal is delivered from the output unit 780 as an HDTV video signal 427.
The output unit 780 is arranged for detecting an error rate in the second data stream of the second data stream output 759 through an error rate detector 782 and if a condition in which the error rate is high continues for a predetermined time, HLVL of low resolution video data systematically are produced for a predetermined time.
Accordingly, the multi-level signal transmission system for digital TV signal transmission and reception becomes feasible. For example, if a TV signal transmitter station is near, both the first and second data streams of a received signal can successfully be reproduced to exhibit an HDTV quality picture. If the transmitter station is far, the first data stream can be reproduced to HLVL which is converted to a low resolution TV picture. Hence, any TV program will be intercepted in a wider area and displayed at a picture quality ranging from HDTV to NTSC level.
When a scrambled 4-level VSB or 8-level VSB is received as shown in
The TV receiver 781 may have a further arrangement shown in
According to the embodiment, a 4-level ASK signal is divided into two, D1 and D2, level components for execution of the one-bit mode multi-level signal transmission. If an 8-level ASk ASK signal as shown in FIGS. 68(a) and (b) illustrating the constellation of the 8-VSB signal, i.e., 8-level VSB signal is used, it can be transmitted in a one-bit mode three-level, D1, D2, and D3, arrangement, thus, three bits pre per symbol in total. A shown in FIG. 68(a), the first bit coding is done as follows. D3 is assigned to eight signal points 721a and 721b; 722a and 722b; 723a and 723b; and 724a and 724b, each pair, i.e., small group, representing a two-level pattern using one bit. Next, the second bit coding is done as follows. D2 is assigned to two signal point groups 721 and 722; and 723 and 724, two mid groups representing a two-level pattern using one bit. Next, the third bit coding is done as follows. D1, is assigned to two large signal point groups 725 and 726 representing a two-level pattern using one bit. More particularly, this is equivalent to a form in which each of the four signal points 721, 722, 723, and 724 shown in
As understood from the above, each of the eight signal points is assigned with three bit data (D1, D2, D3). For example, if left side and right side are defined as logic 0 and logic 1, respectively, the three bit data (D1, D2, D3) for the signal point 722a will be (0, 1, 0). This can be explained as follows. Since the signal point 722a is in the left side of the two large groups 725 and 726, logic 0 is given to D1. Also, the signal point 722a is in the right side of the two mid groups 721 and 722, so that logic 1 is given to D2. Further the signal point 722a is in the left side of the two small groups 722a and 722b, so that logic 0 is given to D3. In a similar manner, the three bit data (D1, D2, D3) for the signal point 723a will be (1, 0, 0).
The three-level signal transmission, such as for the digital HDTV, is identical to that described in the third and fourth embodiments and will no further be explained in detail.
The effects of television broadcasting using the 8-level VSB shown in FIGS. 68(a), (b), and (c) are described below.
While the transmitted data quantity is high with 8-level VSB, it also has a higher error rate than 4-level VSB for the same C/N value. However, in high image quality HDTV transmissions, the available transmission capacity makes it possible to apply more error correction coding, and the error rate can thus be reduced. This band capacity also enable enables multi-level (hierarchical) television broadcasts and other new features in the future.
The relative effects of 4-level, 8-level, and 16-level VSB are described below.
In ground station broadcasts using the NTSC or PAL frequency band, the usable transmission band is effectively limited to approximately 5 MHz because of the 6-MHz frequency limit of the NTSC format, for example, as shown in FIG. 136. With 4-level VSB, the effective data transmission quantity is 5 MHz×4=20 Mbps because the frequency utilization efficiency is 4 bits/Hz. A minimum of 15 Mbps-18 Mbps is required, however, for digital HDTV signal transmission. Because there is no spare capacity with 4-level VSB, the redundancy used for error correction is only 10-20% of the HDTV effective transmission quantity as shown in the comparison chart in FIG. 169.
With 8-level VSB, the effective data transmission quantity is 5 MHz×6=30 Mbps because the frequency utilization efficiency is 5 bits/Hz. While 15 Mbps-18 Mbps is required for digital HDTV signal transmission as described above, when using 8-level VSB modulation, more than 50% of the actual HDTV signal transmission quantity can be used for error correction coding as shown in FIG. 169. As shown by error rate curves 805 and 806 in FIG. 161 163, the error rate relative to the same C/N value in the transmission system is less with TCM 8-level VSB than with 4-level VSB, even through error correction code gain is greater with 8-level VSB than with 4-level VSB, because significantly more error correction coding can be added with 8-level VSB during ground station broadcasting of same-data-rate HDTV digital signals using the 6-MHz band. As a result, 8-level VSB with high code gain error correction coding also has the effect of enabling a larger service area for ground station HDTV broadcasts than does 4-level VSB. While the increased size of the error correction circuits required with 8-level VSB does increase the complexity of the receiver circuitry, the circuit scale of the equalizer in the receiver is significantly smaller than that of receivers using QAM modulation, which contains a phase component, because VSB and ASK are amplitude modulation methods. As a result, an 8-level VSB circuit board containing the error correction circuit is smaller than an equivalent 32-level QAM board with the same transmission capacity.
A digital HDTV receiver with an appropriate circuit scale and a large ground station broadcasting service area can therefore be achieved using 8-level VSB.
Note that the ECC 744a and trellis encoder 744b in the transmitter and receiver block diagrams of
As shown in FIGS. 160(a) and (b) described below with the sixth embodiment, the ECC encoder 744a uses a Reed-Solomon encoder 744j and interleaver 744k, and uses a deinterleaver 759k and Reed-Solomon decoder 759j for the ECC decoder 759a. Applying interleaving as described in the previous embodiment improves resistance to transmission system noise such as burst error.
Code gain can be further increased and the error rate decreased by using a trellis encoder as shown in FIGS. 128(a), (b), (c), (d), (e), and (f). A ratio 2/3 trellis encoder 744b 743c and decoder 759b as shown in
The embodiments have been described using primarily the example of a multilevel (hierarchical) digital television signal. While an ideal broadcasting format can be achieved using a multilevel signal, the image compression circuit and modulator/demodulator circuits become more complex, and are therefore not preferable due to cost for the start of new broadcasting services. As described at the beginning of the fifth embodiment, a broadcasting system with a simple television circuit can be achieved by using a signal-signal interval L=Lo, i.e., an equal interval, in the 4-level VSB and 8-level VSB signals and a non-multilevel television transmission, and by simplifying the circuit shown in
Four- and 8-level VSB have been described above, and 16- and 32-level VSB are described below with reference to FIGS. 159(a)-(b). FIG. 159(a) shows the 16-level VSB constellation. As shown in FIG. 159(b), the signal between two signal points is grouped into eight groups 722a-722h, which are treated as eight signal points and can be treated as 8-level VSB signals to enable a two-stage multilevel transmission. In this case, multilevel transmission can be achieved with time division multiplexing even when intermittently transmitting an 8-level VSB signal. The maximum data rate with this method is 2/3. In FIG. 157(c), the data is further grouped into four groups 723a-723d, which can be handled as 4-level VSB signals adding one more level to the hierarchy. While the maximum data rate drops with time division multiplex transmission of 4-level VSB signals, multilevel transmission is possible with 3-stage multilevel VSB transmission.
With this method, a multilevel transmission whereby 8-level VSB or 4-level VSB data can be reproduced when the C/N ratio of the 16-level VSB data deteriorates can be achieved. By doubling the signal points of the 16-level VSB format as shown in FIG. 159(d), 32-level VSB transmission is enabled. When 16-level VSB capacity is increased in the future, this method will maintain compatibility while making it possible to obtain a 6-bit/symbol data capacity.
By summarizing the above, the VSB receiver shown in the block diagram of FIG. 161 and the VSB transmitter shown in the block diagram of
While 4-level VSB and 8-level VSB are used by way of example above, 16-level VSB as shown in FIGS. 159(a)-(c) can also be used for transmission. With 16-level VSB, a 40-Mbps transmission capacity can be used with a 6-MHz band in ground station broadcasting. Because the data rate of the HDTV digital compression signal is 15-18 Mbps using the MPEG standard, there is excessive reserve in the transmission capacity. As shown in
In particular, the arrangement of the video encoder 401 of the third embodiment shown in
At the receiver side, such a video decoder as described in the third embodiment and shown in
Also, the third video decoder 423 may be modified in which the same action is executed with one single mixer 556 as shown in FIG. 72. At the first timing, five swatches switches 765, 765a, 765b, 765c, 765d remains turned to the position 1. Hence, HLVL, HLVH, HHVL, and HHVH are fed from a first 522, a second 522a, a third 522b and a fourth expander 522c to through their respective switches to the mixer 556 where they are mixed to a single video signal. The video signal which represents HLVL-H of an input high resolution video signal is then fed back through the terminal 1 of the switch 765d to the terminal 2 of the switch 765c. At the second timing, the four switches 765, 765a, 765b, 765c are turned to the point 2. Thus, HHVH-H, HHVL-H, HLVH-H, and HLVL-H are transferred to the mixer 556 where they are mixed to a single video signal which is then sent across the terminal 2 of the switch 765d to the output unit 554 for further delivery.
In this manner of time division processing of a three-level signal, two mixers can be replaced with one mixer.
More particularly, four components HLVL, HLVH, HHVL, and HHVH are fed to produce HLVL-H at the first timing. Then, HLVH-H, HHVL-H, and HHVH-H are fed at the second timing delayed from the first timing and mixed with HLVL-H to a target video signal. It is thus essential to perform the two actions at an interval of time.
If the four components are overlap each other or are supplied in a variable sequence, they have to be time-base adjusted to a given sequence through using memories accompanied with their respective switches 765, 765a, 765b, and 765c. In the foregoing manner, a signal is transmitted from the transmitter at two different timing periods as shown in
As shown in
The technique of reducing the number of the expanders in the decoder will now be explained. FIG. 74(b) shows a time-base assignment of four data components 810, 810a, 810b, and 810c of a signal. When other four data components 811, 811a, 811b, and 811c are inserted between the four data components 811, 811a, 811b, and 811c respectively, the latter can be transmitted at intervals of time. In operation, the second video decoder 422 shown in FIG. 74(a) receives the four components of the first data stream D1 at a first input 521 and transfers them through a switch 812 to an expander 503 one after another.
More particularly, the component 810 first fed is expanded during the feeding of the component 811 and after completion of processing the component 810, the succeeding component 810a is fed. Hence, the expander 503 can process a row of the components at time intervals by the same time division manner as of the mixer, thus substituting for the simultaneous operation of a number of expanders.
Similar effects will be ensured by assignment of the data components to other domains 821, 821a, 821b, and 821c as shown in FIG. 76. This becomes more effective in transmission and reception of a common 4 PSK or ASK signal having no different digital levels.
Here, as shown by the block diagrams of
In view of the error rate, the data 821 of D1-1 channel is less affected by signal interference than the other data 821a, 821b, and 821c of D1-2 channel, while being specifically discriminated and stayed in a different logic level, as described in the second embodiment. While D1 and D2 are divided into two physically different levels, the levels determined by discrimination of the distance between error correcting codes are arranged differently in the logic level.
The demodulation of D2 data requires a higher C/N rate than that for D1 data. In action, HLVL or low resolution NTSC signal can at least be reproduced in a distant or lower C/N service area. HLVH, HHVL, and HHVH can in addition be reproduced at a lower C/N area. Then, at a high C/N area, HLVH -H, HHVL-H, and HHVH-H components can also be reproduced to develop an HDTV signal. Accordingly, three different level broadcast signals can be played back. This method allows the signal receivable area shown in
FIGS. FIG. 78 is a block diagram of the third video decoder arranged for the time-base assignment of data shown in
In operation, both the D1 and D2 signals are fed through two input units 521 and 530 respectively to a switch 812 at the first timing. As their components including HLVL are time divided, they are transferred in a sequence by the switch 812 to an expander 503. This sequence will now be explained referring to the time-base assignment of
At the second timing, HLVH-H, HHVL-H, and HHVH-H of the D2 signal shown in
The operation of the modified decoder 423 is almost identical to that shown in FIG. 78 and associated with the time-base assignment shown in FIG. 77 and will not be explained in greater detail. It is also possible to multiplex data components on the D1 signal as shown in FIG. 81. However, two data components 821 and 822 have increased error correction capability than the other data components 821a, 812b, and 812c, thus staying at a higher signal level. More particularly, the data assignment for transmissions made in one physical level but two logic level relationship. Also, each data component of the second channel is inserted between two adjacent data components of the first channel so that serial processing can be executed at the receiver side and the same effects as of the time-base assignment shown in
The time-base assignment of data components shown in
As understood, the time-base assignment of data components shown in
Accordingly, the receiver can be minimized in the overall construction.
It would be understood that the fifth embodiment is not limited to ASK modulation and the other methods including PSK and QAM modulation, such as described in the first, second, and third embodiments, will be employed with equal success.
Also, FSK modulation will be eligible in any of the embodiments. For example, the signal points of a multiple-level FSK signal consisting of four frequency components F1 f1, f2, f3, and f4 are divided into groups as shown in FIG. 58 and when the distance between any two groups are spaced from each other for ease of discrimination, the multi-level transmission of the FSK signal can be implemented, as illustrated in FIG. 83.
More particularly, it is assumed that the frequency group 841 of f1 and f2 is assigned D1=0 and the group 842 of f3 and f4 is assigned D1=1. If f1 and f3 represent 0 at D2 and f2 and f4 represent 1 at D2, two-bit data transmission, one bit at D1 or D2, will be possible as shown in FIG. 83. When the C/N rate is high, a combination of D1=0 and D2=1 is reconstructed at t=t3 and a combination of D1=1 and D2=0 at t=t4. When the C/N rate is low, D1=0 only is reproduced at t=t3 and D1=1 at t=t4. In this manner, the FSK signal can be transmitted in the multi-level arrangement. This multi-state FSK signal transmission is applicable to each of the third, fourth, and fifth embodiments.
The fifth embodiment may also be implemented in the form of a magnetic record/playback apparatus of which block diagram shown in
The effects of television broadcasting using the 8-level VSB shown in FIGS. 68(a), (b), and (c) are described below.
While the transmitted data quantity is high with 8-level VSB, it also has a higher error rate than 4-level VSB for the same C/N value. However, in high image quality HDTV transmissions, the available transmission capacity makes it possible to apply more error correction coding, and the error rate can thus be reduced. This band capacity also enable enables multi-level (hierarchical) television broadcasts and other new features in the future.
The relative effects of 4-level, 8-level, and 16-level VSB are described below.
In ground station broadcasts using the NTSC or PAL frequency band, the usable transmission band is effectively limited to approximately 5 MHz because of the 6-MHz frequency limit of the NTSC format, for example, as shown in FIG. 136. With 4-level VSB, the effective data transmission quantity is 5 MHz×4=20 Mbps because the frequency utilization efficiency is 4 bits/Hz. A minimum of 15 Mbps-18 Mbps is required, however, for digital HDTV signal transmission. Because there is no spare capacity with 4-level VSB, the redundancy used for error correction is only 10-20% of the HDTV effective transmission quantity as shown in the comparison chart in FIG. 169.
With 8-level VSB, the effective data transmission quantity is 5 MHz×6=30 Mbps because the frequency utilization efficiency is 5 bits/Hz. While 15 Mbps-18 Mbps is required for digital HDTV signal transmission as described above, when using 8-level VSB modulation, more than 50% of the actual HDTV signal transmission quantity can be used for error correction coding as shown in FIG. 169. As shown by error rate curves 805 and 806 in FIG. 161 163, the error rate relative to the same C/N value in the transmission system is less with TCM 8-level VSB than with 4-level VSB, even through error correction code gain is greater with 8-level VSB than with 4-level VSB, because significantly more error correction coding can be added with 8-level VSB during ground station broadcasting of same-data-rate HDTV digital signals using the 6-MHz band. As a result, 8-level VSB with high code gain error correction coding also has the effect of enabling a larger service area for ground station HDTV broadcasts than does 4-level VSB. While the increased size of the error correction circuits required with 8-level VSB does increase the complexity of the receiver circuitry, the circuit scale of the equalizer in the receiver is significantly smaller than that of receivers using QAM modulation, which contains a phase component, because VSB and ASK are amplitude modulation methods. As a result, an 8-level VSB circuit board containing the error correction circuit is smaller than an equivalent 32-level QAM board with the same transmission capacity.
A digital HDTV receiver with an appropriate circuit scale and a large ground station broadcasting service area can therefore be achieved using 8-level VSB.
Note that the ECC 744a and trellis encoder 744b in the transmitter and receiver block diagrams of
As shown in FIGS. 160(a) and (b) described below with the sixth embodiment, the ECC encoder 744a uses a Reed-Solomon encoder 744j and interleaver 744k, and uses a deinterleaver 759k and Reed-Solomon decoder 759j for the ECC decoder 759a. Applying interleaving as described in the previous embodiment improves resistance to transmission system noise such as burst error.
Code gain can be further increased and the error rate decreased by using a trellis encoder as shown in FIGS. 128(a), (b), (c), (d), (e), and (f). A ratio 2/3 trellis encoder 744b 743c and decoder 759b as shown in
The embodiments have been described using primarily the example of a multilevel (hierarchical) digital television signal. While an ideal broadcasting format can be achieved using a multilevel signal, the image compression circuit and modulator/demodulator circuits become more complex, and are therefore not preferable due to cost for the start of new broadcasting services. As described at the beginning of the fifth embodiment, a broadcasting system with a simple television circuit can be achieved by using a signal-signal interval L=L0, i.e., an equal interval, in the 4-level VSB and 8-level VSB signals and a non-multilevel television transmission, and by simplifying the circuit shown in
Four- and 8-level VSB have been described above, and 16- and 32-level VSB are described below with reference to FIGS. 159(a)-(b). FIG. 159(a) shows the 16-level VSB constellation. As shown in FIG. 159(b), the signal between two signal points is grouped into eight groups 722a-722h, which are treated as eight signal points and can be treated as 8-level VSB signals to enable a two-stage multilevel transmission. In this case, multilevel transmission can be achieves achieved with time division multiplexing even when intermittently transmitting an 8-level VSB signal. The maximum data rate with this method is ⅔. In FIG. 157(c) 159(c), the data is further grouped into four groups 723a-723d, which can be handled as 4-level VSB signals adding one more level to the hierarchy. While the maximum data rate drops with time division multiplex transmission of 4-level VSB signals, multilevel transmission is possible with 3-stage multilevel VSB transmission.
With this method, a multilevel transmission whereby 8-level VSB or 4-level VSB data can be reproduced when the C/N ratio of the 16-level VSB data deteriorates can be achieved. By doubling the signal points of the 16-level VSB format as shown in FIG. 159(d), 32-level VSB transmission is enabled. When 16-level VSB capacity is increased in the future, this method will maintain compatibility while making it possible to obtain a 6-bit/symbol data capacity.
By summarizing the above, the VSB receiver shown in the block diagram of FIG. 161 and the VSB transmitter shown in the block diagram of
While 4-level VSB and 8-level VSB are used by way of example above, 16-level VSB as shown in FIGS. 159(a)-(c) can also be used for transmission. With 16-level VSB, a 40-Mbps transmission capacity can be used with a 6-MHz band in ground station broadcasting. Because the data rate of the HDTV digital compression signal is 15-18 Mbps using the MPEG standard, there is excessive reserve in the transmission capacity. As shown in
As explained, the foregoing compatible hardware arrangement is capable of producing both QAM and VSB modulated signals by selectively determining the coefficient values of the coefficient generator 1008.
In this arrangement, the filter circuit is simple as uniaxial although the filter processing speed has to be increased to two times that of the previous arrangement.
In the third modification, the carrier frequency of the analog modulated signal can be changed by varying the counting interval of the counter 1025.
The performance of the Trellis decoder depends on how a code path is determined through internal calculation and should be associated with some path memories.
The results of calculation with the Trellis decoder denoted by 1030 are stored in a given number of path memories, generally 1032, 1033, and 1034, merged in a path memory group 1031. The number of the results is preliminarily determined from the needs at the receiver side, depending on e.g. the transmission channel basis or the broadcasting system of a type. For the purpose, an address control signal is supplied to a memory address generator 1035 which in turn addresses a corresponding one of the path memories to select a code path.
This arrangement allows the single Trellis decoder 1030 associated with the plural memories to perform multiple characteristics as acting as a number of the Trellis decoders. Thus, the demodulator will be reduced in the hardware construction as compared with a conventional circuit arrangement where a plurality of the Trellis decoders are coupled in parallel for selective use.
For elimination of such an interference, notch filters have been used with optimum success. However, any fixed notch filter can hardly conduct a frequency offset signal used in the conventional analog broadcasting system. The receiver shown in
Intercepted broadcasting waveforms including QAM and VSB signals are fed to a tuner 1050. After tuning, an intermediate frequency signal of the selected waveform is band rejected by a filter (SAW) 1051 and converted by an analog converter 1052 to an A/D converter enable frequency. When the selected waveform is a VSB signal, it is subjected to uniaxial detection and AFC detection of a VSB detector 1054. An A/D converter 1053 digitizes a QAM modulated signal directly and a baseband signal extracted from the VSB signal.
When the digitized signal is a QAM signal, it is subjected to digital AFC action of an AFC 1055, I/Q data reconstruction of a QAM detector 1056, and waveform shaping of a roll-off filter 1057. The waveform shaped signal is then transmitted to a waveform equalizer 1059 for removal of unwanted ghost. The output signal of the equalizer 1059 is phase compensated by a carrier recovery 1060 and code reconstructed by a decision 1061. Meanwhile, an AGC detector 1064 produces an AGC signal and an AFC detector 1062 calculates a frequency change from the output of the carrier recovery 1060.
A code form signal of the decision 1061 is then transmitted through a deinterleaver 1065, a Trellis decoder 1066, a deinterleaver 1067, and a Reed-Solomon decoder 1068 and released after an error correction.
The VSB signal output of the A/D converter 1053 is directly fed to the equalizer 1059 and then processed in the same manner as for the QAM signal. It is also subjected to the same error correction as the QAM signal.
There is a clock recovery 1058 operable with the entire symbols for the QAM signal. For the VSB signal, a clock reference signal of a burst form is extracted by a gate signal generator from the VSB output signal of the VSB detector 1054 and fed to the clock recovery 1058 for actuation of its PLL.
EMBODIMENT 6A sixth embodiment of the present invention is a magnetic recording and playback apparatus in which the above transmission and recording method is employed. Although in the fifth embodiment a multiple-level ASK data transmission is described, but it is also feasible in the same manner to adopt this invention in a magnetic recording and playback apparatus of a multi-level ASK recording system as shown in the block diagram of
First of all, the method of realizing a multi-level recording in a 16 QAM or 32 QAM magnetic recording playback apparatus will be explained in compliance with the C-CDM method of the present invention.
AS As shown in
Since the 8 VSB transmission system described above in connection with the fifth embodiment takes 3 bits per symbol, the Trellis encoder 744g and the Trellis decoder 744q with the ratio 2/3 as shown in FIGS. 128(b), (e) can be used, and the entire block diagram will be as shown in FIG. 171.
It is, however, possible to constitute the first data stream input 743 not to include a Trellis encoder as shown in
A format of the recorded signal is shown in a recording signal frequency assignment of
A main signal of 16 SRQAM will have a signal point assignment shown in FIG. 10. Furthermore, a main signal of 36 SRQAM will have a signal point assignment shown in FIG. 100. When 4 ASK, 8 ASK are used, the constellation will be as shown in
As shown in a block diagram of
Above-described high C/N rate video tape 855 capable of recording a high bit-rate signal, e.g. HDTV signal, will be able to use be used in such a low bit-rate dedicated magnetic recording/playback apparatus but will reproduce the first data stream D1 only. That is, the wide NTSC signal is outputted, while the second data stream is not reproduced. In other words, one recording/playback apparatus having a complicated configuration can reproduce a an HDTV signal and the other recording/playback apparatus having a simple configuration can reproduce a wide NTSC signal if a given video tape 855 includes the same multi-level HDTV signal. Accordingly in case of two-level multiple state, four combinations will be realized with prefect perfect compatibility among two tapes having different C/N rates and two recording/playback apparatus having different recording/playback data rates. This will bring a remarkable effect. In this case, an NTSC dedicated apparatus will be simple in construction as compared with an HDTV dedicated apparatus. In more detail, a circuitry scale of ECTV EDTV decoder will be ⅙ of that of HDTV decoder. Therefore, a low function apparatus can be realized at fairly low cost. Realization of two, HDTV and EDTV, types recording/playback apparatus having different recording/reproducing capability of picture quality will provide various type products ranging in a wide price range. Users can freely select a tape among a plurality of tapes from an expensive high C/N rate tape to a cheaper low C/N rate tape, as occasion demands so as to satisfy required picture quality. Not only maintaining perfect compatibility but obtaining expandable capability will be attained and further compatibility with a future system will be ensured. Consequently, it will be possible to establish long-lasting standards for recording/playback apparatus. Other recording methods will be used in the same manner. For example, a multi-level recording will be realized by use of phase modulation explained in the first and third embodiments. A recording using ASK explained in the fifth embodiment will also be possible. A two or three multi-level state will be realized by converting present recording from two-level to four-level ASK or to eight-level ASK and dividing into two group as shown in FIGS. 59(c) and 59(d) or in FIGS. 68(a) and 68(b).
A circuit block diagram for ASK will be as is shown in
Compatibility with future standards will be described below. A setting of standards for recording/playback apparatus such as VCR is normally done by taking account of the most highest C/N rate tape available in practice. The recording characteristics of a tape progresses rapidly. For example, the C/N rate has been improved more than 10 dB compared with the tape used 10 years ago. If it is supposed that new standards will be established after 10 to 20 years due to an advancement of tape property, a conventional method will encounter with difficulty in maintaining compatibility with older standards. New and old standards, in fact, used to be one-way compatible or non-compatible with each other. On the contrary, in accordance with the present invention, the standards are first of all established for recording and/or reproducing the first data stream and/or second data stream on present day tapes. Subsequently, if the C/N rate is improved magnificently in future, an upper level data stream, e.g. a third data stream, will be added without any difficulty as long as the present inventions incorporated in the system. For example, a super HDTV VCR capable of recording or reproducing three-level 64 SRQAM or 8 ASK will be realized while maintaining perfect compatibility with the conventional standards. A magnetic tape, recording first to third data streams in compliance with new standards, will be able to use, of course, in the older two-level magnetic recording/playback apparatus capable of recording and/or reproducing only first and second data streams. In this case, first and second data streams can be reproduced perfectly although the third data stream is left non-reproduced. Therefore, an HDTV signal can be reproduced. For these reasons, the merit of expanding recording data amount while maintaining compatibility between new and old standards is expected.
Returning to the explanation of reproducing operation of
As shown in
In playback operation, a recording signal reproduced through the magnetic head 854 is demodulated into D1 and D2 by the C-CDM demodulator 760 in the same manner as in the explanation of FIG. 84. The first data stream D1 is demodulated into two, D1-1 and D1-2, subchannels through the TDM 758c provided in the first data stream output 758. D1-1 data is error corrected in an ECC decoder 758a having high code gain. Therefore, D1-1 data can be demodulated at a lower C/N rate as compared with D1-2 data. A 1-1 video decoder 402a decodes the D1-1 data and outputs an LDTV signal. On the other hand, D1-2 data is error corrected in an ECC decoder 758b 758b having normal code gain. Therefore, D1-2 data has a threshold value of high C/N rate compared with D1-1 data and thus will not be demodulated when a signal level is not large. D1-2 data is then demodulated in a 1-2 video decoder 402d 402d and summed with D1-1 data to output an EDTV signal of wide NTSC grade.
The second data stream D2 is Vitabi demodulated in a Trellis decoder 759b and error corrected at an ECC decoder 759a 759a. Thereafter, D2 data is converted into a high frequency band video signal through a second video decoder 402b and, then, summed with D1-1 and D1-2 data to output an HDTV signal. In this case, a threshold value of the C/N rate of D2 data is set larger than that of C/N rate of D1-2 2 data. Accordingly, D1-1 data, i.e. an LDTV signal, will be reproduced from a tape 855 having a smaller C/N rate. D1-1 and D1-2 data, i.e. an EDTV signal, will be reproduced from a tape 855 having a normal C/N rate. And, D1-1, D1-2, and D2, i.e. an HDTV signal, will be reproduced from a tape 855 having a high C/N rate.
Three-level magnetic recording/playback apparatus can be realized in this manner. As described in the foregoing description, the tape 855 has an interrelation between C/N rate and cost. The present invention allows users to select a grade of tape in accordance with a content of TV program they want to record because video signals having picture qualities of three grades are recorded and/or reproduced in accordance with tape cost.
Next, an effect of multi-level recording will be described with respect to fast feed playback. As shown in a recording track diagram of
A fast feed reproduction in a reverse direction does not allow a magnetic head trace 855f having an azimuth angle A to coincide with the magnetic track as shown in the drawing. As the present invention provides the D1-1 recording region 855c 855c at a central narrow region of the magnetic tape as shown in
A head trace 855g corresponds to a head trace in the reverse playback operation, from which it is understood only a part of the magnetic track is traced in the reverse playback operation. The recording/playback format shown in
Accordingly, the present invention makes it possible to record a picture image of LDTV grade within a narrow region on the recording track, which results in intermittent reproduction of almost perfect still pictures with picture quality of LDTV grade during normal and reverse fast feed playback operations. Thus, the users can easily confirm picture images even in high-speed seaching.
Next, another method will be described to respond a higher speed fast feed playback operation. A D1-1 recording region 855c is provided as shown at lower right of
In case of a two-level VCR, this method is useless in reproducing the D2 recording region 855j and therefore this region will not be reproduced in a high-speed fast feed playback operation. On the other hand, a three-level high performance VCR will allow users to confirm a picture image even if a fast feed playback operation is executed at a faster, 3 to 5 times as fast as two-level VCR, speed. In other words, not only excellent picture quality is obtained in accordance with the cost but a maximum fast feed speed capable of reproducing picture images can be increased in accordance with the cost.
Although this embodiment utilizes a multi-level modulation system, it is needless to say that a normal, e.g. 16 QAM, modulation system can also be adopted to realize the fast feed playback operation in accordance with the present invention as long as an encoding of picture images is of multiple type.
A recording method of a conventional non-multiple digital VCR, in which picture images are highly compressed, disperses video data uniformly. Therefore, it was not possible in a fast feed playback operation to reproduce all the picture images on a picture plane of the same time. The picture reproduced was the one consisting of a plurality of picture image blocks having non-coincident time bases. The present invention, however, provides a multi-level HDTV VCR which can reproduce picture image blocks having coincided time bases on a picture plane during a fast feed playback operation although its picture quality is of LDTV grade.
The three-level recording in accordance with the present invention will be able to reproduce a high resolution TV signal such as HDTV signal when the recording/playback system has a high C/N rate. Meanwhile, a TV signal of EDTV grade, e.g. a wide NTSC signal, or a TV signal of LDTV grade, e.g. a low resolution NTSC signal, will be outputted when the recording/playback system has a low C/N rate or poor function.
As it is described in the foregoing description, the magnetic recording/playback apparatus in accordance with the present invention can reproduce picture images consisting of the same content even if C/N rate is low or error rate is high, although the resolution or the picture quality is relatively low.
EMBODIMENT 7A seventh embodiment of the present invention will be described for execution of four-level video signal transmission. A combination of the four-level signal transmission and the four-level video data construction will create a four-level signal service area as shown in FIG. 91. The four-level service area is consisted of, from innermost, a first 890a, a second 890b, a third 890c, and a fourth signal receiving area 890d. The method of developing such a four-level service area will be explained in more detail.
The four-level arrangement can be implemented by using four physically different levels determined through modulation or four logic levels defined by data discrimination in the error correction capability. The former provides a large difference in the C/N rate between two adjacent levels and the C/N rate has to be increased to discriminate all the four levels from each other. The latter is based on the action of demodulation and a difference in the C/N rate between two adjacent levels should stay at minimum. Hence, the four-level arrangement is best constructed using a combination of two physical levels and two logic levels. The division of a video signal into four signal levels will be explained.
The input video signal is first divided by the divider 404a 404a into two, high and low, frequency band components, each component being divided into two, horizontal and vertical, segments. The intermediate between the high and low frequency ranges is a dividing point according to the embodiment. Hence, if the input video signal is an HDTV signal of 1000 line vertical resolution, HLVL has a vertical resolution of 500 lines and a horizontal resolution of a half value.
Each of two, horizontal and vertical, data of the low frequency component HLVL is further divided by the divider 404c into two frequency band segments. Hence, an HLVL segment output is 250 lines in the vertical resolution and ¼ of the original horizontal resolution. This output of the divider 404c which is termed as an LL signal is then compressed by the compressor 405a to a D1-1 signal.
The other three higher frequency segments of HLVL are mixed by a mixer 772c to an LH signal which is then compressed by the compressor 405b to a D1-2 signal. The compressor 405b may be replaced by three compressors provided between the divider 404c and the mixer 772c.
HLVH, HHVL, and HHVH form from the divider 404a are mixed by a mixer 772a to an HHVH-H signal. If the input signal is as high as 1000 lines in both horizontal and vertical resolution, HHVH-H has 500 to 1000 lines of a horizontal and a vertical resolution. HHVH-H is fed to the divider 404b where it is divided again into four components.
Similarly, HLVL from the divider 404b has 500 to 750 lines of a horizontal and a vertical resolution and transferred as an HL signal to the compressor 405c. The other three components, HLVH, HHVL, and HHVH, from the divider 404b have 750 to 1000 lines of a horizontal and a vertical resolution and are mixed by a mixer 772b to an HH signal which is then compressed by the compressor 405d and delivered as a D202 D2-2 signal. After compression, the HL signal is delivered as a D2-1 signal. As the result, LL or D1-1 carries a frequency data of 0 to 250 lines, LH or D1-2 carries a frequency data from more than 250 lines up to 500 lines, HL or D2-1 carries a frequency data of more than 500 lines up to 750 lines, and HH or D2-2 carries a frequency data of more than 750 lines to 1000 lines so that the divider circuit 3 can provide a four-level signal. Accordingly, when the divider circuit 3 of the transmitter 1 shown in
The combination of multi-level data and multi-level transmission allows a video signal to be at steps declined in the picture quality in proportion to the C/N rate during transmission, thus contributing to the enlargement of the TV broadcast service area. At the receiving side, the action of demodulation and reconstruction is identical to that of the second receiver of the second embodiment shown in FIG. 88 and will not be explained in greater detail. In particular, the mixer 37 is modified for video signal transmission rather than data communications and will now be explained in more detail.
As described in the second embodiment, a received signal after being demodulated and error corrected, is fed as a set of four components D1-1, D1-2, D2-1, and D2-2 to the mixer 37 of the second receiver 33 of FIG. 88.
The picture quality of the four different components will be described in more detail. The illustration of
Also, as shown in
Hereinafter, an eighth embodiment of the present invention will be explained referring to the drawings. The eighth embodiment employs a multi-level signal transmission system of the present invention for a transmitter/receiver in a cellular telephone system.
On the contrary, the antenna 22 receives transmission radio waves from other base stations as communication signals from other telephones. A received signal is demodulated in a multiple-level, e.g. SRQAM, type demodulator 45 into D1, D2, and D3 data. A timing circuit 767 detects timing signals on the basis of demodulated signals. These timing signals are fed into the time division circuit 765. Demodulated signals D1, D2, and D3 are fed into an expander 503 and expanded into a sound signal, which are transmitted to a speaker 763 and converted into sound.
It is ideal for the improvement of an overall system efficiency to increase the frequency efficiency of the region having a larger traffic amount and decrease the frequency efficiency of the region having a smaller traffic amount. A multi-level signal transmission system in accordance with the present invention realizes this ideal modification. This will be explained with reference to
More specifically,
The receiving cells 768, 769, and 770 employ the multi-level, e.g. SRQAM, signal transmission system. Therefore, it is possible to obtain a frequency utilization efficiency of 6 bit/Hz, three times as large as 2 bit/Hz of QPSK, in the vicinity of the base stations as denoted by data 776a, 776b, and 776c. Meanwhile, the frequency utilization efficiency decreases at steps from 6 bit/Hz to 4 bit/Hz, and 4 bit/Hz to 2 bit/Hz, as it goes to suburban area. if If the transmission power is insufficient, 2 bit/Hz areas become narrower than the receiving cells, denoted by dotted lines 777a, 777b, and 777c, of QPSK. However, an equivalent receiving cell will be easily obtained by slightly increasing the transmission power of the base stations.
Transmitting/receiving operation of a mobile station capable of responding to a 64 SRQAM signal is carried out by use of modified QPSK, which is obtained by set setting a shift amount of SRQAM to S=1, at the place far from the base station, by use of 16 SRQAM at a place not so far from the same, and 64 SRQAM at the near place. Accordingly, the maximum transmission power does not increase as compared with QPSK.
Furthermore, 4 SRQAM type transmitter/receiver, whose circuit configuration is simplified as shown in a block diagram of
As is explained above, the transmission system having a distribution like d=A+B of
Next, data assignment of each time slot will be explained referring to
The present invention, which is characterized by a multi-level, e.g. 64 SRQAM, signal transmission system, allows to have three-level data consisting of D1, D2, and D3 of 2 bit/Hz as shown in FIG. 119(b). As both the A1 and A2 data are transmitted by 16 SRQAM, their time slots have two times the data rate as shown by slots 782b and 782c and 783b and 783c. It means the same quality sound can be transmitted half the time. Accordingly, a time width of respective time slots 782 782b and 782c becomes halved. In this manner, two times the transmission capacity can be acquired at the two-level region 776c shown in
In the same way, time slots 782g and 783g carry out the transmission/reception of E1 data by use of a 64 SRQAM signal. As the transmission capacity is three times, one time slot can be used for three channels of E1, E2, and E3. This would be used for an area further close to the base station. Thus, up to three times the communication capacity can be obtained at the same frequency band. An actual transmission efficiency, however, would be reduced to 90%. It is desirable for enhancing the effect of the present invention to coincide the transmission amount distribution according to the present invention with the regional distributution of the actual traffic amount as perfect as possible.
In fact, an actual urban area consists of a crowded building district and a greenbelt zone surrounding this building area. Even an actual suburb area consists of a residential district and fields or a forest surrounding this residential district. These urban and suburb areas resemble the distribution of the TF diagram. Thus, the application of the present invention will be effective.
On the contrary, 16 SRQAM mode of the present invention uses a time slot 788a for reception of A1 channel and a time slot 788c for transmission to A1 channels as shown in FIG. 120(b) 120(b). A width of the time slot becomes approximately ½. In case of 64 SRQAM mode, a time slot 788i is used for reception of D1 channel and a time slot 7881 is used for transmission to D1 channel. A width of the time slot becomes approximately ⅓.
In order to save electric power, a transmission of E1 channel is executed by use of a normal 4 SRQAM time slot 788 r while reception of E1 channel is executed by use of a 16 SRQAM time slot 788p being a ½ time slot. Transmission power is surely suppressed, although communication cost may increase due to a long occupation time. This will be effective for a small and light portable telephone equipped with a small battery or when the battery is almost worn out. As is described in the foregoing description, the present invention makes it possible to determine the distribution of transmission capacity so as to coincide with an actual traffic distribution, thereby increasing substantial transmission capacity. Furthermore, the present invention allows base stations or terminal stations to freely select one among two or three transmission capacities. If the frequency utilization efficiency is lowered, power consumption will be decreased. If the frequency utilization efficiency is selected higher, communication cost will be saved. Moreover, adoption of a 4 SRQAM having smaller capacity will simplify the circuitry and reduce the size and cost of the telephone. As explained in the previous embodiments, one characteristics of the present invention is that compatibility is maintained among all of associated stations. In this manner, the present invention not only increases transmission capacity but allows to provide customers a wide variety of series from a super mini telephone to a high performance telephone.
EMBODIMENT 9Hereinafter, a ninth embodiment of the present invention will be described referring to the drawings. The ninth embodiment employs this invention in an OFDM transmission system.
As shown in the principle diagram of
A transmitting/receiving action of HDTV signal in accordance with this ninth embodiment will be explained referring to the block diagram of
In a first data stream input 743, D1-1 signal is ECC encoded with high code gain and D1-2 signal is ECC coded with a normal code gain. A TDM 743 performs time division multiplexing of D1-1 and D1-2 signals to produce a D1 signal, which is then fed to a D1 serial to parallel converter 791d in a modulator 852a. D1 signal consists of n pieces of parallel data, which are inputted into first inputs of n pieces of C-CDM modulator 4a, 4b,—respectively.
On the other hand, the high frequency band signal D2 is fed into a second data stream input 744 of the input section 742, in which D2 signal is ECC (Error Correction Code) encoded in an ECC 744a and then Trellis encoded in a Trellis encoder 744b. Thereafter, the D2 signal is supplied to a D2 serial to parallel converter 791b of the modulator 852a and converted into n pieces of parallel data, which are inputted into second inputs of the n pieces of C-CDM modulator 4a, 4b,—respectively.
The C-CDM modulators 4a, 4b, 4c—respectively produces produce 16 SRQAM signal on the basis of D1 data of the first data stream input and D2 data of the second data stream input. These n pieces of C-CDM modulator respectively has have a carrier different from each other. As shown in
Next, an action of a receiver 43 will be described. A received signal, shown as time-base symbol signal 796e of
Since the signal transmission system is of C-CDM multi-level shown in 125(b), both D1 and D2 signals will be demodulated under better receiving condition but only D1 signal will be demodulated under worse, e.g. low C/N rate, receiving condition. Demodulated D1 signal is demodulated in an output section 757. As D1-1 signal has higher ECC code gain as compared with the D1-2 signal, an error signal of the D1-1 signal is reproduced even under worse receiving condition.
The D1-1 signal is converted by a 1-1 video decoder 402c into a low frequency band signal and outputted as an LDTV, and the D1-2 signal is converted by a 1-2 video decoder 402d into a medium frequency band signal and outputted as EDTV.
The D2 signal is Trellis decoded by a Trellis decoder 759b 759b and converted by a second video decoder 402b into a high frequency band signal and outputted as an HDTV signal. Namely, an LDTV signal is outputted in case of the low frequency band signal only. An EDTV signal of a wide NTSC grade is outputted if the medium frequency band signal is added to the low frequency band signal, and an HDTV signal is produced by adding low, medium, and high frequency band signals. As well as the previous embodiment, a TV signal having a picture quality depending on a receiving C/N rate can be received. Thus, the ninth embodiment realizes a novel multi-level signal transmission system by combining an OFDM and a C-CDM, which was not obtained by the OFDM alone.
An OFDM is certainly strong against multipath such as TV ghost because the guard time Tg can absorb an interference signal of multipath. Accordingly, the OFDM is applicable to the digital TV broadcasting for automotive vehicle TV receivers. Meanwhile, no OFDM signal is received when the C/N rate is less than a predetermined value because its signal transmission pattern is non not of a multi-level type.
However the present invention can solve this disadvantage by combining the OFDM with the C-CDM, thus realizing a graditional gradational degradation depending on the C/N rate in a video signal reception without being disturbed by multipath.
When a TV signal is received in a compartment of a vehicle, not only the reception is disturbed by multipath but the C/N rate is deteriorated. Therefore, the broadcast service area of a TV broadcast station will not be expanded as expected if the countermeasure is only for multipath.
On the other hand, a reception of TV signal of at least LDTV grade will be ensured by the combination with the multi-level transmission C-CDM even if the C/N rate is fairly deteriorated. As a picture plane size of an automotive vehicle TV is normally less than 10 inches, a TV signal of an LDTV grade will provide a satisfactory picture quality. Thus, the LDTV grade service area of automotive vehicle TV will be largely expanded. If an OFDM is used in an entire frequency band of HDTV signal, present semiconductor technologies cannot prevent circuitry scale from increasing so far.
Now, an OFDM method of transmitting only D1-1 of low frequency band TV signal will be explained below. As shown in a block diagram in
On the other hand, a signal received by a receiver 43 is first of all frequency separated by an FDM 40e and, then, demodulated by a C-CDM demodulator 4b of the present invention. Thereafter, thus C-CDM demodulated signal is reproduced into medium and high frequency components of HDTV in the same way as in FIG. 123. An operation of a video decoder 402 is identical to that of embodiments 1, 2, and 3 and will not be explained in greater detail.
Meanwhile, the D1-1 signal, a low frequency band signal of MPEG 1 grade of HDTV, is converted by a serial to parallel converter 791 into a parallel signal and fed to an OFDM modulator 852c, which executes QPSK or 16 QAM modulation. Subsequently, the D1-1 signal is converted by an inverse FFT 40 into a time-base signal and transmitted at a frequency band B through a FDM 40d.
On the other hand, a signal received by the receiver 43 is frequency separated in the FDM 40e and then converted into a number of frequency-base signals in an FFT 40a of an OFDM modulator demodulator 852d. Thereafter, frequency-base signals are demodulated in respective demodulators 4a, 4b,—and are fed into a parallel to serial converter 882a, wherein a D1-1 signal is demodulated. Thus, a D1-1 signal of LDTV grade is outputted from the receiver 43.
In this manner, only an LDTV signal is OFDM modulated in the multi-level signal transmission. The method of
An OFDM signal transmission system is strong against multipath and will soon be applied to a moving station, such as a portable TV, an automotive vehicle TV, or a digital music broadcast receiver, which is exposed under strong and variable multipath obstruction. For such usages a small picture size of less than 10 inches, 4 to 8 inches, is the mainstream. It will be thus guessed that the OFDM modulation of a high resolution TV signal such as HDTV or EDTV will bring less effect. In other words, the reception of a TV signal of LDTV grade would be sufficient for an automotive vehicle TV.
On the contrary, multipath is constant at a fixed station such as a home TV. Therefore, a countermeasure against multipath is relatively easy. Less effect will be brought to such a fixed station by OFDM unless it is in a ghost area. Using OFDM for medium and high frequency band components of HDTV is not advantageous in view of present circuit scale of OFDM which is still large.
Accordingly, the method of the present invention, in which OFDM is used only for a low frequency band TV signal as sown shown in
Although the OFDM modulation of
The multi-level signal transmission according to the present invention is feasible in this manner and produces various effects as previously described. Furthermore, if the multi-level signal transmission of the present invention is incorporated with an OFDM, it will become possible to provide a system strong against multipath and to alter data transmission grade in accordance with receivable signal level change.
FIG. 126(a) shows another method of realizing the multi-level signal transmission system, wherein the subchannels 794a-794c of the OFDM are assigned to a first layer 801a and the subchannels 794d-794f are assigned to a second layer 801b. There is provided a frequency guard zone 802a of fg between these two, first and second, layers. FIG. 126(b) shows an electric power difference 802b of Pg which is provided to differentiate the transmission power of the first and second layers 801a and 801b.
Utilization of this differentiation makes it possible to increase electric power of the first layer 801a in the range not obstructing the analogue TV broadcast service as shown in FIG. 108(d) previously described. In this case, a threshold value of the C/N ratio capable of receiving the first layer 801a becomes lower than that for the second layer 801b as shown in FIG. 108(e). Accordingly, the first layer 801a can be received even in a low signal-level area or in a large-noise area. Thus, a two-layer signal transmission is realized as shown in FIG. 147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) in this specification. If this PW-OFDM system is combined with the C-CDM system previously explained, three layers will be realized as shown in FIG. 108(e) and, accordingly, the signal receivable area will be correspondingly expanded.
A signal received by the receiver 43 is separated into several signals having carriers of f1-fn through the FFT 40a. The carriers f1-f3 are demodulated by the demodulators 45a-45c to reproduce the first data stream D1, i.e. the first layer 801a. On the other hand, the carriers f6-f8 are demodulated by the demodulators 45d-45f to reproduce the second data stream D2, i.e. the second layer 801b.
The first layer 801a has so large electrical power that it can be received even in a weak-signal area. In this manner, the PW-OFDM system realizes the two-layer multi-level signal transmission. If this PW-OFDM is combined with the C-CDM, it will become possible to provide 3-4 layers. As the circuit of
Next, a method of realizing a multi-level signal transmission in Time-Weighted-OFDM (i.e. TW-OFDM) in accordance with the present invention will be explained. Although the OFDM System is accompanied with the guard time zone tg as previously described, adverse affects of ghosts will be eliminated if the delay time tM of the ghost, i.e. multipath, signal satisfies the requirement of tM<tg. The delay time tM will be relatively small, for example in the range of several microsounds microseconds, in a fixed station such as a TV receiver used for home use. Furthermore, as its value is constant, cancellation of ghosts will be relatively easily done. On the contrary, reflected waves will increase in case of a mobile station such as a vehicle TV receiver. Therefore, the delay time tM becomes relatively large, for example in the range of several tens microsound microsecond. Furthermore, the magnitude of tM varies in response to the running movement of the vehicle. Thus, cancellation of ghosts tends to be difficult. Hence, the multi-level signal transmission is key or essential for such a mobile station TV receiver in order to eliminate adverse affection of multipath.
The multi-level signal transmission in accordance with the present invention will be explained below. A symbol contained in the subcannel subchannel layer A can be intensified against the ghosts by setting a guard time tga of the layer A to be larger than a guard time tgb of the layer B as shown in FIG. 146. In this manner, the multi-layer signal transmission can be realized against multipath by use of weighting of guard time. This system is referred to as Guard-Time-Weighted-OFDM (i.e. QTW-OFDM).
If the symbol number of the symbol time Ts is not different in the layer A and in the layer B, a symbol time tsa of the layer A is set to be larger smaller than a symbol time tsb of the layer B. With this differentiation, a carrier width Δfa of the carrier A becomes larger than a carrier width Δfb of the carrier B. (Δfa>Δfb) (Δfa<Δfb) Therefore, the error rate becomes lower in the demodulation of the symbol of the layer A compared with the demodulation of the symbol of the layer B. Thus, the differentiation of the layer A and B in the weighting of the symbol time Ts can realize a two-layer signal transmission against multipath. This system is referred to as Carrier-Spacing-Weighted-OFDM (i.e. CSW-OFDM).
By realizing the two-layer signal transmission based on the GTW-OFDM, wherein a low-resolution TV signal is transmitted by the layer A and a high-frequency component is transmitted by the layer B, the vehicle TV receiver can stably receive the low-resolution TV signal regardless of tough ghost. Furthermore, the multi-level signal transmission with respect to the C/N ratio can be realized by differentiating the symbol time tS based on the CSW-OFDM between the layers A and B. If this CSW-OFDM is combined with the GTW-OFDM, the signal reception in the vehicle TV receiver can be further stabilized. High resolution is not normally required to the vehicle TV or the portable TV.
As the time ratio of the symbol time including a low-resolution TV signal is small, an overall transmission efficiency will not decrease so much even if the guard time is enlarged. Accordingly, using the GTW-OFDM of the present invention for suppressing multipath by laying emphasis on the low-resolution TV signal will realize the multi-layer type TV broadcast service wherein the mobile station such as the portable or vehicle TV receiver can be compatible with the stationary station such as the home TV without substantially lowering the transmission efficiency. If combined with the CSW-OFDM or the C-CDM as described previously, the multi-layer to the C/N ratio can be also realized. Thus, the signal reception in the mobile station will be further stabilized.
An affection of the multipath will be explained in more detail. In case of multipath 810a, 810b, 810c, and 810d having shorter delay time as shown in FIG. 145(a), the signals of both the first and second layers can be received and therefore the HDTV signal can be demodulated. On the contrary, in case of multipath 811a, 811b, 811c, and 811d having longer delay time as shown in FIG. 145(b), the B signal of the second layer cannot be received since its guard time tgb is not sufficiently long. However, the A signal of the first layer can be received without being bothered by the multipath since its guard time tga is sufficiently long. As described above, the B signal includes the high-frequency component of TV signal. The A signal includes the low-frequency component of TV signal. Accordingly, the vehicle TV can reproduce the LDTV signal. Furthermore, as the symbol time Tsa is set larger than symbol time Tsb, the first layer is strong against deterioration of C/N ratio.
Such a discrimination of the guard time and the symbol time is effective to realize two-dimensional multi-layer signal transmission of the OFDM in a simple manner. If the discrimination of guard time is combined with the C-CDM in the circuit shown in
Next, a specific example will be described below.
The smaller the D/U ratio of the receiving signal becomes, the larger the multipath delay time TM becomes. Because, the reflected wave increases compared with the direct wave. For example, as shown in
Accordingly, as shown in detail in FIGS. 149(a) and 149(b), three groups of first 801a, second 801b, and third 801 c 801c layers are assigned in a 2 ms period of 1 sec TV signal. The guard times 797a, 797b, and 797c, i.e. Tga, Tgb, and Tgc, of these three groups are weighted to be, for example, 50 microsounds microseconds, 5 microsounds microseconds, and 1 microsound microsecond, respectively, as shown in FIG. 149(c). Thus, three-layer signal transmission effective to the multipath will be realized as shown in
If the GTW-OFDM is applied to ass the picture quality, it is doubtless
At the same time, the multi-layer signal transmission effective to C/N ratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, a two-dimensional multi-layer signal transmission is realized with respect to the multipath and the C/N ratio as shown in FIG. 151. As described previously, it is possible to combine the CSW-OFDM and the C-CDM of the present invention for preventing the overall transmission efficiency from being lowered. In the first, 1-2, and 1-3 layers 801a, 851a, and 851az, the LDTV grade signal can be stably received by, for example, the vehicle TV receiver subjected to the large multipath TM and low C/N ratio. In the second and 2-3 layers 801b and 851b, the standard-resolution SDTV grade signal can be received by the fixed or stationary station located, for example, in the fringe of the service area which is generally subjected to the lower C/N ratio and ghost. In the third layer 801c which occupies more than half of the service area, the HDTV grade signal can be received since the C/N ratio is high and the ghost is less because of large direct wave. In this manner, a two-dimensional multi-layer broadcast service effective to both the C/N ratio and the multipath can be realized by the combination of the GTW-OFDM and the C-CDM or the combination of the GTW-OFDM and the CSW-C-CDM in accordance with the present invention. thus Thus, the present invention realizes a two-dimensional, matrix type, multi-layer signal transmission system effective to both the C/N ratio and the 194 multipath, which has not ever been realized by the prior art technologies.
A timing chart of a three level (HDTV, SDTV, LDTV) television signal in a two-dimensional multilevel broadcast of three C/N levels and three multipath levels is shown in FIG. 152. As shown in the figure, the LDTV signal is positioned in slot 796a1 of the first level of level layer A, the level with the greatest resistance to multipath interference; the SDTV synchronization signal, address signal, and other important high priority signals are positioned in slot 796a2, which has the next greatest resistance to multipath interference, and slot 796b1, which has strong resistance to C/N deterioration. The SDTV common signal, i.e., low priority signals, and HDTV high priority signals are positioned in levels 2 and 3 of level B. SDTV, EDTV, HDTV, and other high frequency component television signals are positioned in levels 1, 2, and 3 of level C.
As the resistance to C/N deterioration and multipath interference increases, the transmission rate drops, causing the TV signal resolution to drop, and achieving the three-dimensional graceful degradation effect shown in FIG. 153 and unobtainable with conventional methods. As shown in
The present embodiment has been described using the example of a two-dimensional multilevel broadcast structure obtained by combining GTW-OFDM of the invention with C-CDM of the invention as previously described, or combining GTW-OFDM, CSW-C-CDM, but other two-dimensional multilevel broadcast structures can be obtained by combining GTW-OFDM and power-weighted OFDM, or GTW-OFDM with other C/N ratio multilevel transmission methods.
By using a time positioning varying the time width of guard times 797a, 797b, and 797c for each symbol 796a, 796b, and 796c as shown in
Block diagrams of a specific ECC encoder 744j and a specific ECC decoder 749j 759j are shown in FIG. 160a and
Burst noise interference can be reduced by interleaving the data in this way. By using a 4-level VSB, 8-level VSB, or 16-level VSB transmission apparatus as described in embodiments 4, 5, and 6, respectively, and shown in the VSB receiver block diagram (
By using 3-level broadcasting by means of the method shown in
The multi-level signal transmission method of the present invention is intended to increase the utilization of frequencies but may be suited for not all the transmission systems since causing some type receivers to be declined in the energy utilization. It is a good idea for use with a satellite communications system for selected subscribers to employ most advanced transmitters and receivers designed for best utilization of applicable frequencies and energy. Such a specific purpose signal transmission system will not be bound by the present invention.
The present invention will be advantageous for use with a satellite or terrestrial broadcast service which is essential to run in the same standards for as long as 50 years. During the service period, the broadcast standards must not be altered but improvements will be provided time to time corresponding to up-to-date technological achievements. Particularly, the energy for signal transmission will surely be increased on any satellite. Each TV station should provide a compatible service for guaranteeing TV program signal reception to any type receivers ranging from today's common ones to future advanced ones. The signal transmission system of the present invention can provide a compatible broadcast service of both the existing NTSC and HDTV systems and also, ensure a future extension to match mass date data transmission.
The present invention concerns much on the frequency utilization than the energy utilization. The signal receiving sensitivity of each receiver is arranged different differently depending on a signal state level to be received so that the transmitting power of a transmitter needs not be increased largely. Hence, existing satellites which offer a small energy for reception and transmission of a signal can best be used with the system of the present invention. The system is also arranged for performing the same standards corresponding to an increase in the transmission energy in the future and offering the compatibility between old and new type receivers. In addition, the present invention will be more advantageous for use with the satellite broadcast standards.
The multi-level signal transmission method of the present invention is more preferably employed for terrestrial TV broadcast service in which the energy utilization is not crucial, as compared with satellite broadcast service. The results are such that the signal attenuating regions in a service area which are attributed to a conventional digital HDTV broadcast system are considerably reduced in extension and also, the compatibility of an HDTV receiver or display with the existing NTSC system is obtained. Furthermore, the service area is substantially increased so that program suppliers and sponsors can appreciate more viewers. Although the embodiments of the present invention refer to 16 and 32 QAM procedures, other modulation techniques including 64, 128, and 256 QAM will be employed with equal success. Also, multiple PSK, ASK, and FSK techniques will be applicable as described with the embodiments.
A combination of the TDM with the SRQAM of the present invention has been described in the above. However, the SRQAM of the present invention can be combined also with any of the FDM, CDMA and frequency dispersal communications systems.
Claims
1. A signal transmission and reception apparatus for transmitting and receiving an n-level VSB signal, the apparatus comprising a transmitter and a receiver;
- said transmitter comprising: a compression means for compressing an input video signal to a digital video compression signal; an error correction encoding means for adding an error correction code to the digital video compression signal to produce an error correction coded signal; a modulation means for modulating the error correction coded signal to an n-level VSB modulation signal, said modulation means comprising a means for allocating code points along a uniaxial modulation coordinate system, and a filter means having a plurality of coefficients which are a series of impulse responses defined by plotting timebase responses to the VSB modulation signal along the in-phase axis and its orthogonal axis for filtering a series of said code points allocated along the uniaxial modulation coordinate system; and a transmission means for transmitting the modulation signal, and said receiver comprising: a means for receiving a transmitted n-level VSB modulation signal; a demodulation means for demodulating the received n-level VSB modulation signal into a digital reception signal; an error correction means for error correcting the digital reception signal to obtain an error-corrected digital signal; and an expanding means for expanding the error-corrected digital signal to obtain a video output signal.
2. A transmission and reception apparatus according to claim 1, wherein the error correction means comprises a trellis decoder.
3. A transmission and reception apparatus according to claim 2, wherein the trellis decoder is associated with a plurality of memories which each holds a number of selectable correct codes.
4. A transmission and reception apparatus according to claim 1, wherein the digital reception signal is divided into a high priority signal and a low priority signal, and wherein said error correction means comprises a high code gain first error correction means and a low code gain second error correction means, said first error correction means correcting the high priority signal.
5. A transmission and reception apparatus according to claim 4, wherein the high priority signal carries the address data for all data.
6. A transmission and reception apparatus according to claim 4, wherein the first error correction means comprises a trellis decoder.
7. A signal transmission and reception apparatus according to claim 1, further comprising a band path filtering means for filtering the n-level VSB modulation signal before being transmitted.
8. A signal transmission and reception apparatus for transmitting an n-level VSB signal, comprising:
- a compression means for compressing an input video signal into a digital video compression signal;
- an error correction encoding means for adding an error correction code to the digital video compression signal to produce an error correction coded signal;
- a modulation means for modulating the error correction coded signal to an n-level VSB modulation signal, said modulation means comprising a means for allocating code points along a uniaxial modulation coordinate system, and a filter means having a plurality of coefficients which are a series of impulse responses defined by plotting timebase responses to the VSB modulation signal along the in-phase axis and its orthogonal axis for filtering a series of said code points allocated along the uniaxial modulation coordinate system; and
- a transmission means for transmitting the modulation signal.
9. A signal transmission apparatus according to claim 8, further comprising a band path filtering means for filtering the n-level VSB modulation signal before being transmitted.
10. A signal receiving apparatus comprising:
- a tuner for receiving a transmission signal containing a digital modulation signal and an analog modulation signal and for selecting the digital modulation signal using a local oscillation signal;
- an interference detecting means for detecting interference caused by the analog modulation signal from the digital modulation signal selected by the tuner;
- a notch filter means responsive to the interference detected by the interference detecting means for removing a carrier of the analog modulation signal in a same frequency band as a frequency band of the digital modulation signal;
- an error ratio calculating means for calculating a bit error ratio of an output of the notch filter means; and
- an automatic frequency correcting means for changing a frequency of the local oscillation signal of the tuner according to a level of the interference detected by the interference detecting means and the bit error ratio calculated by the error ratio calculating means to compensate for a frequency offset of the carrier of the analog modulated signal.
11. A signal receiving apparatus according to claim 10, wherein the digital modulation signal is an n-level VSB modulation signal.
12. A signal receiving apparatus comprising:
- a tuner for receiving a transmission signal containing at least one of a VSB modulated signal and a QAM modulated signal and for selecting one of the VSB modulated signal and the QAM modulated signal to obtain a selected signal;
- an analog-to-digital converter for converting the selected signal into a series of digital codes;
- a transversal filter provided on an orthogonal axis for suppressing a transmission distortion of the series of digital codes with respect to both orthogonal axes to obtain a series of filtered digital codes allocated on the orthogonal axes;
- a carrier recovery means for phase-compensating a carrier of the filtered digital codes allocated on the orthogonal axis outputted from the transversal filter; and
- a control means for producing a control signal to extract detected codes at equal time intervals from the VSB modulated signal;
- a clock reproducing means for phase synchronizing entire codes of the QAM modulated signal when the selected signal is the QAM modulated signal and for phase synchronizing codes of the VSB modulated signal intermittently at predetermined intervals when the selected signal is the VSB modulated signal; and
- a decoding means for decoding an output of the carrier recovery means.
13. A signal transmission system comprising a transmission apparatus and a receiving apparatus,
- said transmission apparatus comprising: a multiplexer operable to time division multiplex a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data included in each frame in this order, to produce a data stream including a plurality of frames, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame; a modulator operable to modulate the data stream to produce a digital modulated signal; and a transmitter operable to transmit the digital modulated signal;
- said receiving apparatus comprising: a receiver operable to receive the digital modulated signal; and a demodulator operable to demodulate the received signal modulated signal to produce a demodulated signal according to the frame sync data, the first block sync data and second block sync data.
14. A signal transmission apparatus comprising:
- a multiplexer operable to time division multiplex a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data included in each frame in this order, to produce a data stream including a plurality of frames, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame;
- a modulator operable to modulate the data stream to produce a digital modulated signal; and
- a transmitter operable to transmit the digital modulated signal.
15. A receiving apparatus comprising:
- a receiver operable to receive the digital modulated signal,
- wherein said received digital modulated signal includes a data stream, data stream has a plurality of frames, each frame of the data stream has a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data, and is time-division-multiplexed in this order, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame; and
- a demodulator operable to demodulate the received digital modulated signal to produce a demodulated signal according to the frame sync data, the first block sync data and second block sync data.
16. A signal receiving apparatus according to claim 15, further comprising:
- a decoder operable to decode the demodulated signal to a video signal.
17. A signal receiving apparatus according to claim 16, further comprising:
- an output operable to output the video signal.
18. A signal receiving apparatus according to claim 16, further comprising:
- a display operable to display the video signal.
19. A signal transmission and receiving method comprising a transmission method and a receiving method,
- said transmission method comprising: time-division-multiplexing a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data included in each frame in this order, to produce a data stream including a plurality of frames, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame; modulating the data stream to produce a digital modulated signal; and transmitting the digital modulated signal;
- said receiving method comprising: receiving the digital modulated signal; and demodulating the received digital modulated signal to produce a demodulated signal according to the frame sync data, the first block sync data and second block sync data.
20. A signal transmission method comprising:
- time-division-multiplexing a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data included in each frame in this order, to produce a data stream including a plurality of frames, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame;
- modulating the data stream to produce a digital modulated signal; and
- transmitting the digital modulated signal.
21. A receiving method comprising:
- receiving a digital modulated signal,
- wherein said received digital modulated signal includes a data stream, data stream has a plurality of frames, each frame of the data stream has a frame sync data, a first block sync data, a first block data, a second block sync data and a second block data, and is time-division-multiplexed in this order, wherein the first block sync data is located following the frame sync data, the first block data is located following the first block sync data, the second block sync data is located following the first block data, and the second block data is located following the second block sync data, and wherein the frame sync data is used for both synchronization of the frame and identification of the head of the frame; and
- demodulating the received digital modulated signal to produce a demodulated signal according to the frame sync data, the first block sync data and second block sync data.
| 3824548 | July 1974 | Sullivan et al. |
| 3909721 | September 1975 | Bussgang et al. |
| 3959726 | May 25, 1976 | Hinoshita et al. |
| 4003002 | January 11, 1977 | Snijders et al. |
| 4048572 | September 13, 1977 | Dogliotti et al. |
| 4053713 | October 11, 1977 | Nitadori |
| 4077021 | February 28, 1978 | Csajka et al. |
| 4227152 | October 7, 1980 | Godard et al. |
| 4267592 | May 12, 1981 | Craiglow |
| 4271527 | June 2, 1981 | Armstrong |
| 4303939 | December 1, 1981 | Stephens et al. |
| 4303941 | December 1, 1981 | Marti et al. |
| 4309726 | January 5, 1982 | Tanaka et al. |
| 4525846 | June 25, 1985 | Bremer et al. |
| 4535352 | August 13, 1985 | Haskell |
| 4564858 | January 14, 1986 | Resch |
| 4581639 | April 8, 1986 | Judge |
| 4597090 | June 24, 1986 | Forney, Jr. |
| 4601045 | July 15, 1986 | Lubarsky |
| 4630287 | December 16, 1986 | Armstrong |
| 4636876 | January 13, 1987 | Schwartz |
| 4751478 | June 14, 1988 | Yoshida |
| 4769819 | September 6, 1988 | Matsutani et al. |
| 4794447 | December 27, 1988 | Tsinberg |
| 4800426 | January 24, 1989 | Glenn |
| 4817192 | March 28, 1989 | Phillips et al. |
| 4855692 | August 8, 1989 | Kennard et al. |
| 4873701 | October 10, 1989 | Tretter |
| 4882725 | November 21, 1989 | Noda et al. |
| 4891806 | January 2, 1990 | Farias et al. |
| 4903125 | February 20, 1990 | Parker |
| 4912706 | March 27, 1990 | Eisenberg et al. |
| 4914655 | April 3, 1990 | Johannes et al. |
| 4918437 | April 17, 1990 | Jasinski et al. |
| 4937844 | June 26, 1990 | Kao |
| 4940954 | July 10, 1990 | Aubert et al. |
| 4958360 | September 18, 1990 | Sari |
| 4985769 | January 15, 1991 | Yasumoto et al. |
| 5007088 | April 9, 1991 | Ooi et al. |
| 5018198 | May 21, 1991 | Takahashi |
| 5023889 | June 11, 1991 | Divsalar et al. |
| 5029003 | July 2, 1991 | Jonnalagadda |
| 5038219 | August 6, 1991 | Yamashita et al. |
| 5038402 | August 6, 1991 | Robbins |
| 5050188 | September 17, 1991 | Dirr |
| 5063445 | November 5, 1991 | Nishizawa et al. |
| 5063452 | November 5, 1991 | Higurashi |
| 5086340 | February 4, 1992 | Citta et al. |
| 5087975 | February 11, 1992 | Citta et al. |
| 5105442 | April 14, 1992 | Wei |
| 5115453 | May 19, 1992 | Calderbank et al. |
| 5124852 | June 23, 1992 | Suwa et al. |
| 5128758 | July 7, 1992 | Azadegan et al. |
| 5134464 | July 28, 1992 | Basile et al. |
| 5136380 | August 4, 1992 | Cho |
| 5142353 | August 25, 1992 | Kageyama et al. |
| 5146335 | September 8, 1992 | Kim et al. |
| 5148422 | September 15, 1992 | Sako et al. |
| 5164963 | November 17, 1992 | Lawrence et al. |
| 5166924 | November 24, 1992 | Moose |
| 5168509 | December 1, 1992 | Nakamura et al. |
| 5170413 | December 8, 1992 | Hess et al. |
| 5181112 | January 19, 1993 | Citta et al. |
| 5197061 | March 23, 1993 | Halbert-Lassalle et al. |
| 5210770 | May 11, 1993 | Rice |
| 5214656 | May 25, 1993 | Chung et al. |
| 5218436 | June 8, 1993 | Sugiyama et al. |
| 5218622 | June 8, 1993 | Fazel et al. |
| 5243629 | September 7, 1993 | Wei |
| 5267021 | November 30, 1993 | Ramchandran et al. |
| 5276507 | January 4, 1994 | Uwabata et al. |
| 5282019 | January 25, 1994 | Basile et al. |
| 5291289 | March 1, 1994 | Hulyalkar et al. |
| 5311547 | May 10, 1994 | Wei |
| 5311550 | May 10, 1994 | Fouche et al. |
| 5377051 | December 27, 1994 | Lane et al. |
| 5398073 | March 14, 1995 | Wei |
| 5400084 | March 21, 1995 | Scarpa |
| 5452015 | September 19, 1995 | Hulyalkar |
| 5481412 | January 2, 1996 | Bannai et al. |
| 5519730 | May 21, 1996 | Jasper et al. |
| 5555275 | September 10, 1996 | Oshima |
| 5561468 | October 1, 1996 | Bryan et al. |
| 5565926 | October 15, 1996 | Bryan et al. |
| 5565932 | October 15, 1996 | Citta et al. |
| 5576902 | November 19, 1996 | Lane et al. |
| 5598220 | January 28, 1997 | Citta et al. |
| 5638112 | June 10, 1997 | Bestler et al. |
| 5706428 | January 6, 1998 | Boer et al. |
| 5802241 | September 1, 1998 | Oshima |
| 5819000 | October 6, 1998 | Oshima |
| 5892879 | April 6, 1999 | Oshima |
| 5950124 | September 7, 1999 | Trompower et al. |
| 5999569 | December 7, 1999 | Oshima |
| 6049651 | April 11, 2000 | Oshima |
| 6256357 | July 3, 2001 | Oshima |
| 6526264 | February 25, 2003 | Sugar et al. |
| 6977944 | December 20, 2005 | Brockmann et al. |
| 7158577 | January 2, 2007 | Oshima |
| 7352822 | April 1, 2008 | Oshima |
| 7362813 | April 22, 2008 | Oshima |
| 20050058633 | March 17, 2005 | Epstein et al. |
| 20050084750 | April 21, 2005 | Wang et al. |
| 2095435 | May 1992 | CA |
| 1499725 | May 2004 | CN |
| 2627574 | July 2004 | CN |
| 1569334 | January 2005 | CN |
| 0031193 | July 1981 | EP |
| 0122805 | October 1984 | EP |
| 0282298 | September 1988 | EP |
| 0311188 | April 1989 | EP |
| 0329158 | August 1989 | EP |
| 0365431 | April 1990 | EP |
| 0392538 | October 1990 | EP |
| 0448492 | September 1991 | EP |
| 0485105 | May 1992 | EP |
| 0485108 | May 1992 | EP |
| 0490552 | June 1992 | EP |
| 0506400 | September 1992 | EP |
| 0525641 | February 1993 | EP |
| 0531046 | March 1993 | EP |
| 0540231 | May 1993 | EP |
| 1528854 | May 2005 | EP |
| 1529838 | May 2005 | EP |
| 2 187 611 | September 1987 | GB |
| 53-108215 | September 1978 | JP |
| 53-137657 | December 1978 | JP |
| 57-39629 | March 1982 | JP |
| 58-107740 | June 1983 | JP |
| 58-161427 | September 1983 | JP |
| 58-161547 | September 1983 | JP |
| 60-24753 | February 1985 | JP |
| 61-70861 | April 1986 | JP |
| 62-133842 | June 1987 | JP |
| 62-502932 | November 1987 | JP |
| 63-28145 | February 1988 | JP |
| 63-180222 | July 1988 | JP |
| 63-180280 | July 1988 | JP |
| 63-253738 | October 1988 | JP |
| 64-5135 | January 1989 | JP |
| 64-68144 | March 1989 | JP |
| 64-74836 | March 1989 | JP |
| 2-94814 | April 1990 | JP |
| 2-141049 | May 1990 | JP |
| 2-154583 | June 1990 | JP |
| 2-166979 | June 1990 | JP |
| 2-195732 | August 1990 | JP |
| 2-218279 | August 1990 | JP |
| 2-260726 | October 1990 | JP |
| 2-278940 | November 1990 | JP |
| 3-13145 | January 1991 | JP |
| 3-48587 | March 1991 | JP |
| 3-69295 | March 1991 | JP |
| 04-196822 | July 1992 | JP |
| 4-322592 | November 1992 | JP |
| 5-75568 | March 1993 | JP |
| 5-167633 | July 1993 | JP |
| 5-218978 | August 1993 | JP |
| 7-79415 | March 1995 | JP |
| 7-99522 | April 1995 | JP |
| 2004-159207 | June 2004 | JP |
| 52118 | December 2002 | UA |
| 85/04541 | October 1985 | WO |
| 86-07223 | December 1986 | WO |
| 89/08366 | September 1989 | WO |
| 91/20137 | December 1991 | WO |
| 92/14343 | August 1992 | WO |
| 92/22162 | December 1992 | WO |
- Kazuhiko Nitadori, Synthesis of Multichannel Orthogonal VSB Signals by Quadrature Method, 1976 International Conference on Communications, Jun. 14-16, 1976, vol. 1, pp. 3-25-3-29.
- Thomas M. Cover, Broadcast Channels, IEEE Transactions on Information Theory, Jan. 1972, pp. 2-14.
- “Optimum Weighted PCM for Speech Signals”, Sundberg, IEEE Transactions on Communications, vol. COM-26, No. 6, Jun. 1978, pp. 872-881.
- Martin Vetterli et al., Multiresolution Coding Techniques for Digital Television: A Review, Multidimensional Systems and Signal Processing, vol. 3, May 1992, pp. 161-187.
- Ezio Biglieri et al., Introduction to Trellis-Coded Modulation with Applications, Apr. 6, 1992, pp. 173-207.
- English Language Abstract of European Patent No. 93 30 7575.
- K. M. Uz et al., Multiresolution Source and Channel Coding for Digital Broadcast of HDTV, 1992, pp. 61-69.
- Nambi Seshadri et al., Multi-Level Block Coded Modulations with Unequal Error Protection for the Rayleigh Fading Channel, vol. 4, No. 3, May-Jun. 1993, pp. 325-334.
- William F. Schreiber, Spread-Spectrum Television Broadcasting, SMPTE Journal, Aug. 1992, pp. 538-549.
- Khaled Fazel et al., Combined multilevel coding and multiresolution modulation, Feb. 8, 1993, pp. 1081-1085.
- P. Hoeher et al., Performance of an RCPC-Coded OFDM-based Digital Audio Broadcasting (DAB) System, IEEE Global Telecommunications Conference “Globecom '91”, Dec. 2-5, 1991, vol. 1 of 3, pp. 2.1.1-2.1.7.
- K. M. Uz et al., Combined multiresolution source coding and modulation for digital broadcast of HDTV*, 1992, pp. 283-292.
- Tricia Hill et al., “A Performance Study of NLA 64-State QAM”, IEEE Transactions on Communications, vol. COM-31, Jun. 1983, No. 6, pp. 821-826.
- Hideki Ishio et al., “A Proposal of a Carrier Digital Transmission System Using Multi-Level APSK”, pp. 1-20.
- Shanmugam, “Digital and Analog Communication Systems” 1979, p. 272.
- Shinji Matsumoto et al., “200 Mb/s 16 QAM Digital Radio-Relay System Operating in 4 and 5 GHz Bands”, Japan Telecommunications Review, Jan. 1982, vol. 24, No. 1, pp. 65-73.
- M. Pecot et al., “Compatible Coding of Television Images, Part 2. Compatible System”, Signal Processing Image Communication, Oct. 2, 1990, No. 3, pp. 259-268.
- Mitsuaki Oshima, “Constellation-Code Division Multiplex for Digital HDTV”, IEEE, 1992, pp. 1086-1092.
- Shigeki Moriyama et al., “Delay Propagation Characteristics at VHF and UHF bands in Urban Area”, 1991 Spring National Convention Record, The Institute of Electronics, Information and Communication Engineers, Part 2, p. 406, Mar. 15, 1991.
- Yasuhiro ITO et al., “Adaptive Weighted Code Division Multiplexing (AW-CDM) Transmission System for Terrestrial Digital Television Broadcasting”, ITE Technical Report vol. 17, No. 13, pp. 27-32, Feb. 25, 1993.
- J. A. C. Bingham, “Multicarrier Modulation for Data Transmission: An Idea Whose Time Has Come”, IEEE Communications Magazine, vol. 28, May 1990, pp. 5-8 and 11-14.
- B. Hirosaki, “An Orthogonally Multiplexed QAM System Using the Discrete Fourier Transform”, IEEE Transactions on Communications, vol. COM-29, No. 7, Jul. 1981, pp. 982-989.
- L. J. Cimini, Jr., “Analysis and Simulation of a Digital Mobile Channel Using Orthogonal Frequency Division Multiplexing”, IEEE Transactions on Communications, vol. Com-33, No. 7, Jul. 1985, pp. 665-675 and Annex.
- Masafumi Saito et al., “Bit Error Rate Characteristics of OFDM in Multipath Environment”, 1991.
- Ryutaro Ohmoto et al., “π/4-shift QPSK Subcarrier Transmission”, 1991.
- U.S. Office Action issued May 5, 2008 in U.S. Appl. No. 11/980,510.
- U.S. Office Action issued Jun. 28, 2008 in U.S. Appl. No. 11/905,978.
Type: Grant
Filed: Oct 27, 2003
Date of Patent: Apr 14, 2009
Assignee: Panasonic Corporation (Osaka)
Inventors: Mitsuaki Oshima (Kyoto), Seiji Sakashita (Osaka)
Primary Examiner: Dac V. Ha
Attorney: Wenderoth, Lind & Ponack, L.L.P.
Application Number: 10/693,526
International Classification: H04L 5/16 (20060101);
